01707921

IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 54, NO. 10, OCTOBER 2006 2827

Dual-Polarization Dual-Coverage Reflectarray forSpace Applications

Jose A. Encinar, Member, IEEE, Leri Sh. Datashvili, J. Agustn Zornoza, Manuel Arrebola,Manuel Sierra-Castaer, Member, IEEE, Jose Luis Besada-Sanmartn, Horst Baier, and Herv Legay

AbstractA breadboard of a three-layer printed reflectarrayfor dual polarization with a different coverage in each polarizationhas been designed, manufactured, and tested. The reflectarrayconsists of three layers of rectangular patch arrays separated bya honeycomb and backed by a ground plane. The beam shapingfor each polarization is achieved by adjusting the phase of thereflection coefficient at each reflective element independently foreach linear polarization. The phase shift for each polarization iscontrolled by varying either the x or y patch dimensions. Thedimensions of the rectangular patches are optimized to achievethe required phase shift for each beam at central and extremefrequencies in the working band. The reflectarray has been de-signed to produce a contoured beam for a European coveragein H-polarization in a 10% bandwidth, and a pencil beam toilluminate the East Coast in North America in V-polarization.The measured radiation patterns show that gain requirements arepractically fulfilled in a 10% bandwidth for both coverages, andthe electrical performances of the breadboard are close to those ofa classical dual gridded reflector.

Index TermsContoured beam, dual gridded reflector, dualpolarization, multilayer, printed arrays, reflectarray.

I. INTRODUCTION

PRINTED reflectarrays with contoured beams can be an al-ternative to the onboard shaped reflectors in space appli-cations, because of their lower cost and shorter manufacturingtime, since custom moulds are eliminated. A contoured beam re-flectarray was demonstrated in [1] for Direct Broadcast Satellite(DBS) applications using a single-layer printed reflectarray withpatches of variable size [2]. However, this breadboard sufferedfrom the bandwidth limitations inherent in single-layer reflec-tarrays, as shown in [3] and [4]. The bandwidth in large reflec-tarrays can be improved by stacking three layers of rectangularpatches and optimizing the patch dimensions to compensate thespatial phase delay in the working frequency band, as proposedin [5]. Using this technique, a 1-m reflectarray was designedfor a focused beam with 10% bandwidth at -band. The sametechnique was applied in [6] to design an 80-cm contoured beamreflectarray in -band for a South American coverage with10% bandwidth.

Manuscript received February 1, 2006; revised May 10, 2006. This work haswas supported by ESA ESTEC under Contract ESTEC/16919/02/NL/JA.

J. A. Encinar, M. Arrebola, M. Sierra-Castaer, and J. L. Besada-Sanmartnare with Universidad Politecnica de Madrid, 28040 Madrid, Spain (e-mail:[email protected]).

L. S. Datashvili and H. Baier are with the Institute of Lightweight Structures,TU Munich, D-85747 Garching, Germany.

J. A. Zornoza is with Antenna CoC, EADS Astrium Ltd., Stevenage SG12AS, U.K.

H. Legay is with Alcatel Alenia Space, Toulouse, France.Digital Object Identifier 10.1109/TAP.2006.882172

Reflectarrays can be easily designed for dual polarizationwith low levels of cross polarization. A reflectarray made upof two arrays of orthogonal dipoles of variable lengths wasproposed in [7] for dual polarization and frequency reuse,in which the array of vertical dipoles acts as a reflector forthe vertical polarization and that of horizontal dipoles for theother polarization. Reflectarrays using single or multiple layersof rectangular patches can also be designed for dual linearpolarization, by an independent adjustment of the orthogonaldimensions of the patches, as described in [3] and [4]. Fur-thermore, the reflectarray can be designed to work at differentfrequencies for each linear polarization. A reflectarray wasdesigned for operation at 24 and 60 GHz in orthogonal linearpolarizations, by choosing different period and by independentadjustment of the patch dimensions in the orthogonal directionsof the reflectarray [8].

Dual polarization antennas in space applications require veryhigh isolation between polarizations, which cannot be achievedwith parabolic or shaped offset reflectors. To obtain this isola-tion between polarizations, dual-gridded reflectors with two su-perimposed grid reflectors and a separate feed for each polariza-tion are used. The dual-gridded antenna is a mature concept interms of technological process and simulation tools, but suffersfrom high cost and large volume and mass. Provided that the re-quired bandwidth for space applications can be achieved with athree-layer configuration, and taking advantage of the low levelof cross polarization of the reflectarrays, dual polarization re-flectarrays can be an alternative to dual-gridded reflectors. Thephase adjustment in the reflectarray is carried out independentlyfor each polarization, allowing the use of two separate feeds oflinear polarization, as shown in Fig. 1. If two feeds located atdifferent focal points are used, one for each polarization, the di-mensions of the conductive patches in each element can be ad-justed to compensate the position of each feed. The dimensionscan also be adjusted in order to generate two collimated beamsin different directions, one for each polarization.

In the present work, a three-layer reflectarray has been de-signed to replace a dual-gridded reflector in a Telecom satellitefor dual polarization with different coverages in each polariza-tion. A contoured beam for an European coverage is requiredfor H-polarization in the frequency band 11.4512.75 GHz, anda pencil beam to illuminate the East Coast in North America(Washington DC, New York, and Montreal, QC, Canada) withV-polarization in the frequency band 11.4511.7 GHz. In re-flector technology, these requirements can only be met witha dual-gridded reflector, with a different reflecting surface for

0018-926X/$20.00 2006 IEEE

2828 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 54, NO. 10, OCTOBER 2006

Fig. 1. Dual-polarization dual-coverage three-layer reflectarray. (a) Periodiccell. (b) Reflectarray configuration.

each polarization. The novelty of this work is the demonstrationthat two independent beams, one for each polarization, can begenerated with a single reflectarray surface, and consequently, asignificant reduction in mass and volume is achieved. A bread-board has been designed, manufactured, and tested, includingmechanical and thermo-elastic aspects for space applications.The antenna performances are close to those of the referencedual-gridded antenna.

II. ELECTRICAL DESIGNThe coverage requirements for H- and V-polarization are

shown in Fig. 2 for a satellite at 5 West orbital position. ForH-polarization, the requirements consist of two gain contours of28.5 dBi (solid line) and 25.5 dBi (dashed line). For V-polariza-tion, the minimum gain requirement is 37 dBi in the coverageregion. The outer contours represent the specifications takinginto account typical pointing errors of the satellite (0.1 in roll,0.1 in pitch, and 0.5 in yaw). A cross-polar discriminationof 30 dB is required for both coverages. These requirementsare fulfilled by a conventional dual-gridded reflector antennawith 1-m diameter reflecting surfaces, which has been de-signed, manufactured, and tested by Alcatel Alenia Space, andit is considered as reference for comparison of the electricalperformances.

Fig. 2. Contoured requirements for Europe (H-polarization) and North Amer-ican (V-polarization) coverages.

A. Subsystem DefinitionTo fulfill the previous requirements in beam shaping, gain,

bandwidth, and cross polarization, a three-layer printed reflec-tarray with rectangular patches of variable size [6] is proposed.The periodic cell and the reflectarray configuration are shownin Fig. 1. The reflectarray consists of an elliptical flat panel withaxes 1036 980 mm, which is the same aperture surface as inthe dual-gridded antenna, and with the reflective surface placedon the plane. The reflectarray is made up of three stackedarrays of rectangular patches on a ground plane, with the ele-ments uniformly distributed in a square grid. The total numberof elements is 4068 distributed in 74 columns and 70 rows. Thetwo feedhorns are placed with the phase center on the plane,so that the projection of the field radiated by each feed on the

plane is parallel to one side of the rectangular patches, andthen the phase shift for each polarization can be controlled in-dependently by each patch dimension.

A corrugated horn is used for both feeds. It has been checkedthat the radiation patterns can be simulated as a func-tion. The power has been chosen at each frequency to fit the

ENCINAR et al.: DUAL-POLARIZATION DUAL-COVERAGE REFLECTARRAY FOR SPACE APPLICATIONS 2829

TABLE IVALUES USED FOR qqq FACTOR TO SIMULATE THE RADIATION

PATTERNS OF FEED HORN

horn patterns and the results are given in Table I. The coor-dinates (in millimeters) of the phase center for each feed arechosen as andfor V- and H-polarization, respectively. These feed positionsprovide an illumination at the reflectarray edges 18.6 dB forH-polarization and 16 dB for V-polarization at central fre-quencies, which are typical values in shaped reflector antennas.The period for the reflective elements is defined as 14 14 mm( at 12.75 GHz and at 11.45 GHz). Both feed loca-tions and period have been chosen to avoid the appearance ofgrating lobes at any reflectarray element, considering the angleof incidence of the field coming from the feed.

For the contoured European beam, a phase-only synthesistechnique based on the intersection approach has been appliedto obtain the phase distribution on the reflectarray surface forH-polarization. For V-polarization, a pencil beam is consid-ered because it provides the maximum gain and fulfills the gainrequirements.

B. Pattern Synthesis for European Coverage in H-PolarizationThe contoured beam requirements are specified by a mask

with minimum and maximum values of gain in a region of the plane for H-polarization. Due to the very large number

of elements in a reflectarray for space applications, a directsynthesis method, in which the patch dimensions are optimizedsimultaneously to synthesize a required radiation pattern, isimpractical and the procedure in two steps described in [6] isimplemented. In the first step, assuming a fixed amplitude dis-tribution on the reflectarray surface given by the feed radiationpattern, a phase-only synthesis is applied to compute the phaseof the reflected field at each reflectarray element that providesthe required contoured pattern. In the second step, the patchdimensions are adjusted element by element to achieve theprevious phase distribution and its frequency variations in agiven frequency band.

In the first step, a several-stage phase-only synthesis methodbased on the intersection approach technique [9] is applied toachieve the phase distribution for a required pattern at a givenfrequency as described in [10]. This synthesis method presentsan effective converge to a desired solution and is very efficientfor reflectarrays with a large number of elements as a result ofusing two-dimensional fast Fourier transform (FFT) algorithmsin its implementation. Besides, it allows to easily include theamplitude constraints imposed by the feed in the reflectarray,becoming then the phases the only variables to be optimized.The method is applied first at central frequency, and the requiredphase distribution on the reflectarray surface is obtained.

For the design of the reflectarray in a frequency band by com-pensating the phase delay as described in [5], the appropriatephase delay distribution for the contoured beam is required notonly at central, but also at extreme frequencies. Assuming thesame variation with frequency of the required phase shift on the

Fig. 3. Required phase shift at central frequency (12.1 GHz) obtained by thephase-only synthesis method.

reflectarray as that corresponding to a pencil beam, the phasedistribution is computed at any frequency from the one ob-tained at . This phase distribution is already a good approxi-mation for the required pattern at frequency as demonstratedin [6], and it is used as starting point for a new pattern synthesisat extreme frequencies, as described in [11]. For each extremefrequency, the intersection approach is applied in only one stepusing the real illumination taper and limiting the maximum vari-ation in phase to a small value (typically around 30 ). Sincethe initial phase distribution is close to the final solution, the al-gorithm rapidly converges to a solution that fulfils the require-ments. This process ensures a smooth variation of phase distri-bution with frequency, in the same range as the correspondingto a pencil beam, which is important to facilitate the design ofthe reflectarray patches.

The difference of phase delay at each extreme and central fre-quency is defined as , where

is the phase delay at element and frequency , thecentral frequency, and the frequency at one band extreme.The differences of phase delay are computed for the extremefrequencies, and the phase delay at other frequencies is obtainedby assuming a linear variation. For the reflectarray designed ina frequency band as described in [5], only phase shift (limitedto a 360 range) at central frequency , and differencesat the band extremes, and , are used.

The starting point for the first synthesis stage at central fre-quency (12.1 GHz) is the phase distribution corresponding toa pencil beam with an edge illumination on the reflectarray of

42.7-dB respect to the center, obtained by modeling the feed asa , with . After applying the intersection approachin several stages increasing the edge illumination (decreasingthe factor), the phase distribution on the reflectarray surface isreached for a 18.6-dB edge illumination, which correspondsto the specified horn with . The radiation patterns prac-tically fulfill the mask of requirements with very low sidelobes.The resulting phase shift at central frequency (12.1 GHz), isshown in Fig. 3. The phase is assumed constant in each cell,


Fig. 4. Difference of phase delay at extreme frequencies for H-polarization. (a)At 11.45 GHz. (b) At 12.75 GHz.

which is defined by the position (Nx, Ny) in the reflectarray.After applying the pattern synthesis at 11.45 and 12.75 GHz,the phase differences shown in Fig. 4 are obtained. The requiredphase distributions at extreme frequencies are obtained by di-rectly adding the phase differences of Fig. 4 and the phase-shiftat central frequency shown in Fig. 3. The radiation patterns cor-responding to these phase distributions are shown in Fig. 5 andpractically fulfill the contour requirements.

C. Reflectarray DesignOnce the phase distribution is obtained, the design of the re-

flectarray consists of determining the dimensions of the patchesto achieve the phase-shift distribution. First, the design is carriedout independently for each polarization at central frequency. Forthe dual-polarization dual-coverage reflectarray, the rectangularpatch dimensions are obtained by adjusting the -dimensionsto achieve the required phase shift at 12.1 GHz for H-polariza-tion shown in Fig. 3, and the -dimensions to produce the phase

Fig. 5. Radiation patterns for H-polarization corresponding to the phase distri-bution of Figs. 3 and 4. (a) At 11.45 GHz. (b) At 12.75 GHz.

distribution at 11.575 GHz corresponding to a pencil beam inV-polarization, shown in Fig. 6. In this stage, a fixed relativesize of the stacked patches is maintained, and the dimensionsare adjusted in each cell to match the objective phase for eachpolarization by using a zero finding routine that calls the anal-ysis routine iteratively.

The analysis technique is a full-wave Method of Moments inspectral domain [12], assuming each element in a periodic arrayenvironment. The real angles of incidence at each element andthe polarization of the incident field, obtained from the posi-tion of the feed and the element, are taken into account for theanalysis of each element, assumed in an infinite array. Underthis local periodicity approach, the analysis method is used tocompute the amplitude and phase of the field components of thereflected field at each reflectarray cell. The radiation pattern iscomputed from these field components on the reflectarray. Thisapproach is very efficient and it is accurate when the variation inpatch dimensions is smooth from one cell to the next, because it


Fig. 6. Required phase shift at central frequency (11.575 GHz) forV-polarization.

Fig. 7. Configuration of reflectarray panel with electrical and stiffening layers.(a) Lay-up. (b) Section of manufactured sandwich.

takes into account all mutual coupling between patches. All thedielectric layers defined in the mechanical design, see Fig. 7 andTables II and III, are accurately modeled in the analysis routine.

To overcome the frequency-band limitation in reflectarrays,the phase delay must be compensated on each element withthe phase of the reflection coefficient within a frequency band.At each element, the phase of the reflection coefficient at cen-tral frequency and the difference of phases at extremefrequencies , , and central frequency ,

, are com-

puted by the analysis routine for each polarization.Starting fromthe patch dimensions obtained from the design at central fre-quency, the next design stage is performed element by element,using an optimization routine based on FletcherPowell algo-rithm that adjusts all the dimensions of the stacked patches in el-ement simultaneously to match both, objective phaseand phase delay differences and forboth polarizations. Note that for each polarization, objectivephase, difference in phase delay, position of the feed (and there-fore angle of incidence) and phase of the reflection coefficientare different. Then, it is more convenient to perform the opti-mization in the frequency band sequentially, first for one po-larization and then for the other. For H-polarization, with the

TABLE IIMATERIALS AND LAY-UP OF THE REFLECTARRAY PANEL

TABLE IIIDIELECTRIC PROPERTIES AND THICKNESS OF MATERIALS

electric field on the X-direction, the optimization is carried outby minimizing the following error function:

(1)

where and are weighting coefficients and a superindexto indicate each extreme frequency in the band. After the opti-mization, the -dimensions of all the patches on each layer areobtained, then the process is repeated to adjust the -dimensionsby minimizing the error function

(2)

Note that for each polarization, the orthogonal dimensions ofthe patches are maintained unchanged.

The process is repeated several times, alternating optimiza-tion of - and -dimensions, to take into account the slight in-fluence of the orthogonal dimensions of the patches. After sev-eral alternating optimizations for - (Europe) and - (NorthAmerica) polarization, the final patch dimensions are obtained.For V-polarization the reflectarray is designed in the frequencyband 11.0512.1 GHz, which is larger than the requirement.


The radiation patterns computed by the analysis routinefor the resulting three-layer reflectarray practically fulfillthe coverage requirements in the whole frequency band11.4512.75 GHz for H-polarization, but with a small reduc-tion in gain (0.4 dB) produced by the dissipative losses. A slightdistortion of the contours is observed at extreme frequenciesdue to small phase errors that remain after optimizations. Theradiation patterns are computed in gain, since the radiatedfield is divided by the total power radiated by the feed. Themaximum gain including dissipative losses in the materials is30.1 dBi at 11.45 GHz and 30.0 dBi at 12.75 GHz. Gain hasalso been computed assuming lossless materials, and maximumgain is increased in 0.4 dB, therefore, this is the estimated valuefor dissipative losses in the reflectarray. For V-polarization, thecomputed radiation patterns fulfill the gain requirements in thewhole designed band 11.0512.1 GHz including dissipativelosses, which are also evaluated as 0.4 dB.

III. MECHANICAL DESIGN MANUFACTURE AND TESTSThe reflectarray panel consists of three electrical layers: B,

C, and D in Fig. 7(a), which include copper patches printedon Kapton foils with a Kevlar/resin stiffening layer on one sideonly. Electrical layers are separated using Nomex honeycomb.The electrical layers together with the ground plane (layer A)form the electrical sandwich shown in Fig. 7(a). A backsidestiffening sandwich with carbon fiber reinforced plastic (CFRP)face-sheets is used to support the reflecting layers of the elec-trical sandwich, see Fig. 7. It was demonstrated that the conduc-tive metal ground plane of the reflectarray can be substituted bya CFRP face-sheet, with an increase of dissipative losses around0.1 dB, which is a typical value in CFRP reflectors. Therefore,layer A of the stiffening sandwich serves as a ground plane atthe same time. The materials and lay up of the reflectarray panelare given in Table II.

A. Selection of MaterialsThe selection of materials to form the complete sandwich

panel is made according to two major requirements: to reducedissipative losses to the minimum, and to achieve a goodthermo-elastic stability of the sandwich.

Copper patches in the electrical layers produce high thermaldeformations due to their high coefficient of thermal expansion(CTE) and Yongs modulus. To compensate these deformations,a composite material with close to zero or even negative CTE isrequired at each electrical layer. Kevlar 49 plane-weave fabricwith a negative CTE of the fibers was selected for the thermaldeformation compensation at electrical layer level. It was iden-tified by electromagnetic simulations using the Method of Mo-ments that dissipative losses are mainly produced by the losstangent of the Kevlar/resin layers. The lowest radio-frequency(RF) loss factor compared to the other resin systems was foundin Cyanate Esters and is around 0.005. Cyanate Ester LTM 123(from ACG Ltd.) was selected as the resin system for the sand-wich manufacturing.

Based on previous considerations, a Kevlar fabric rein-forced Cyanate Ester LTM123 composite is used to stiffen theKaptoncopper reflective layers against thermal loads. The hon-eycomb separators are bonded to the electrical layers by a thinfilm of LTM 123 resin. For the backside sandwich of the panelthe same resin and T300 carbon-fiber fabric have been used.

Fig. 8. Reflectarray breadboard.

The electrical properties and the thickness of each dielectriclayer are given in Table III. Nominal values are considered forKapton and honeycomb layers. However, the dielectric constant(DK), loss tangent (LT), and thickness of the Kevlar/TM123layer, which are the most critical parameters in the electricaldesign, have been measured. The resin used to bond the honey-comb to the electric layers is modeled for the electrical designas a uniform film of 60- m thickness, computed from its weight

70 g/m , although in reality it is distributed in hexagonalcells as the honeycomb. The data shown in Table III are used inthe electrical design.

B. Manufacture, Mechanical Design, and Analysis of theReflectarray Panel

The reflectarray panel has been manufactured by a multistepcuring process in order to achieve maximum accuracy and re-peatability in the thickness and composition of each electricallayer. The process is based on independent pre- and post-curingof each layer, previous to bonding of the honeycomb separatorsto the Kevlar face-sheets.

The breadboard is assembled by using a support structure,which ensures an accurate positioning of the feedhorn and thereflectarray panel. The same feed is used for measurements in H-and V-polarization by displacement of the arm and by changingthe adaptor used to fix the feed, see Fig. 8.

A parametric finite-element (FE) model of the reflectarraysandwich panel was established using layered solid elementswith all layers of the reflectarray panel modeled as homoge-nous layers. Variation of all parameters have been performed forstudying the influence of each of them on the panel surface de-viation RMS. The most important parameters are the core thick-ness of backside support sandwich, thickness of Kevlar/epoxylayers, and fiber volume fraction. These parameters allow goodadjustment and tuning of the resulting RMS of the surfaces outof plane deformations.

The thickness of copper patches influences very much theRMS, and must be kept as low as possible. A thickness of17 m has been chosen for the breadboard, because it wasthe thinnest commercially available. Suppression of the patchthermal deformations using Kevlar/resin plies at both sidesof the Kaptoncopper layer gives better results for surfaceRMS. However, a design optimization, performed using an


TABLE IVEXPERIMENTAL AND SIMULATED RMS COMPARISON

TABLE VMEASURED GAIN FOR V-POLARIZATION

evolutionary algorithm, has shown the advantages of the use ofthe single Kevlar/resin ply. The option with one Kevlar/resinlayer on one side of the Kapton film is satisfying the RFrequirements having lower losses and less RMS within theweight region less than 3 kg/m . Based on these results, thesandwich configuration defined in Table II, with a backsidecore thickness of 30 mm, is chosen for the reflectarray panel.

The established FE model with optimized parameters wasused to carry out reflectarray surface deviation RMS analyses.As already mentioned, all layers of the laminate were con-sidered as homogeneous layers including the layers of copperpatches. For the models effective material properties prac-tically all values were experimentally measured. Two worstthermal load cases for the calculations were defined with thecorresponding through thickness gradient. The results show thatthe maximum RMS for the panel when exposed to the extremelow temperatures (temperature difference of 230 ) is equal to0.425 mm. It has been checked by electromagnetic simulationsthat this RMS value does not produce any appreciable effect onthe radiation pattern of the reflectarray.

C. Experimental Thermo-Mechanical VerificationMeasurement of thermo-elastic deformations were carried

out for the reflectarray panel at uniform temperatures. Surfaceshape deviations were measured using photogrammetry withmeasurement accuracy in the range of 20 m. An average ofmaximum out of plane deformations and the RMS of surfacethermal deformations for both simulation and experiment arecalculated and show a good correlation, as shown in Table IV.

Experimental modal analysis and corresponding finite-ele-ment method (FEM) analysis show good agreement as well. Thecalculated and measured values of eigenfrequencies are equal to156 and 146 Hz, respectively, for the unconstrained reflectarraypanel.

IV. ELECTRICAL PERFORMANCESCopolar and cross-polar radiation patterns and gain for both

polarizations have been measured in a planar near-field system.

Fig. 9. Measured copolar gain contour patterns for V-polarization at extremefrequencies. (a) At 11.05 GHz. (b) At 12.1 GHz.

The measured patterns in gain are represented as contoured linesand are superimposed to the mask with the requirements, in thereflectarray coordinate system shown in Fig. 1.

A. Measured Patterns for V-PolarizationThe measured copolar contour gain patterns at extreme fre-

quencies are shown in Fig. 9 for V-polarization. The gain re-quirements of 37 dBi are fulfilled in the whole frequency bandfrom 11.05 to 12.1 GHz, which is wider than required. Themaximum gain measured at the coaxial input connector (SMA)input, including all the losses, is given in Table V for several fre-quencies. Also, the maximum cross-polar radiation levels in dBi(outside of the coverage region) are given in the table. The cross-polarization levels in the USA coverage are always lower than3 dBi in the whole frequency band (11.0512.1 GHz), whichcorresponds to a cross-polar isolation better than 34 dB. Thecross-polar patterns are shown in Fig. 10 for central frequency.Note that the cross-polarization levels on the European cov-erage, where the interference with H-polarization will happenare very low ( 5 dBi).


Fig. 10. Measured cross-polar patterns in dBi for V-polarization at central fre-quency (11.575 GHz).

TABLE VIMEASURED GAIN FOR H-POLARIZATION

B. Measured Patterns for H-PolarizationFig. 11 shows the measured co- and cross-polar contour

gain patterns for H-polarization at central frequency. The gainpatterns practically fulfill the mask of requirements in closeagreement with the simulated radiation patterns. The contourof 28.5 dBi is however obtained with 28 dBi of gain in the 99%of the enlarged coverage including satellite pointing errors.Since the gain requirements of 28.5 dBi was achieved in wholefrequency band for the ideal phase distribution, the reductionof 0.5 dB in gain is the result of dissipative losses (around0.4 dB) and the small errors in phase after the optimizations ofthe patch dimensions. This small reduction in gain was alreadypredicted in the theoretical radiation patterns obtained byelectromagnetic simulations, using the analysis routine basedon Method of Moments. The contour lines for three gain levels(28, 25, and 20 dBi) at 12.1 GHz obtained from measurementsand simulations are compared in Fig. 12. Very good agreementis observed between measured and simulated gain, except forvery slight deviations in the shape of the contoured pattern. Thesmall distortions in the measured patterns can be produced bysmall variations in the thickness and composition (nonhomo-geneity) of the Kevlar-resin layers, which are the more criticalfactors. Fig. 13 shows the gain patterns at 11.7 and 13 GHz, andTable VI shows the maximum values of gain and cross-polarlevels at different frequencies. The cross-polar levels are verylow in the USA coverage ( 8 dBi) where the interferencewith V-polarization occurs.

Fig. 11. Measured gain contour patterns for H-polarization at 12.1 GHz.(a) Copolar pattern. (b) Cross-polar pattern.

The gain patterns practically fulfill the mask of requirementsfrom 11.7 to 13 GHz. However, a small shift in the frequencyband to higher frequencies is observed, because the breadboardwas designed in the 11.4512.75 GHz band. It must be noticedthat the frequency shift was observed only in H-polarization,and the reason for that must be the anisotropy of the honey-comb produced by its hexagonal structure. It was reported thatDK is different for each polarization [13]: DK is higher alongthe ribbon, with electric field in direction of L, see Fig. 14, andlower across the ribbon, with electric field in the direction ofW, with typical values from DK to DK . Also, theresin used for bonding to the honeycomb is distributed along thehexagonal cells and will also exhibit a similar anisotropy. In thebreadboard, electric field is across the ribbon for H-polarizationand along the ribbon for V-polarization. Since honeycomb andresin anisotropy was not considered in the design and DKwas assumed for both polarizations, the agreement is good forV-polarization, while a frequency shift is produced in H-polar-ization as a result of the lower DK. In addition, the small error inthe DK for H-polarization can also be the reason for the small


Fig. 12. Measured and simulated gain contours at central frequency.

deviations in the contour patterns, as a result of the nonlinearbehavior of the phase.

The radiation patterns were also measured at 12.1 GHz in thewhole angular range in a spherical near-field system, in orderto check possible spurious radiation out of the coverage region,and to measure the directivity and dissipative losses. The lossesare obtained as the difference between measurements of gainand directivity, and they are equal to 0.35 with an accuracy inthe measurements of 0.18 dB, which is in close agreementwith the predicted value of 0.4 dB. For other frequencies in theband, the dissipative losses have not been measured, but theymust be in a similar value, since measured gain is maintained invery similar levels and in close agreement with gain predictions.Also for V-polarization, the measured gain coincides with the-oretical predictions, and therefore the dissipative losses shouldbe similar. Fig. 15 shows the cuts of the co- and cross-polar ra-diation patterns at the different phi planes in the whole angularrange ( , ). This figure shows thatthere are no spurious sidelobes out of the coverage region. Themaximum levels of cross polarization referred to the maximumgain are also shown in the whole angular range, and are betterthan 28 dB below the maximum.

V. COMPARISON WITH DUAL-GRIDDED REFLECTORAND CONCLUSION

A 1-m reflectarray made of three stacked layers of rectangularpatches has been designed for dual polarization with a differentcoverage in each polarization. The patch dimensions have beenoptimized to match the required phase shift for both polariza-tions in the required frequency bands. Appropriate mechanicaldesign and analysis of the reflectarray panel has allowed a sig-nificant reduction of dissipative losses to 0.35 dB. A breadboardhas been manufactured and tested, including thermoelastic char-acterization, and RF tests. The measured radiation patterns prac-tically fulfill the requirements for both coverages.

The electrical performances of reflectarray breadboard havebeen compared with those of the reference dual-gridded an-tenna. For the directive beam in V-polarization, the gain mea-sured in the North American coverage is comparable with that

Fig. 13. Measured copolar gain patterns for H-polarization at extreme frequen-cies. (a) At 11.7 GHz. (b) At 13.0 GHz.

Fig. 14. Hexagonal configuration of honeycomb.

measured in the dual-gridded reflector. For the contoured beamin H-polarization, the gain is slightly lower than for the dual-gridded reflector and small distortions are observed in the con-tour pattern, produced by honeycomb anisotropy and manufac-turing tolerances. The reflectarray breadboard has demonstratedthe capacity to achieve satisfactory cross polarization isolation,with cross-polar isolation better than 30 dB in the other cov-erage. Therefore, the reflectarray antenna can be an alternative


Fig. 15. Measured radiation patterns at 12.1 GHz for different phi planes in thewhole angular range (0 90 , 0 ' 360 ). (a) Copolar patterns.(b) Cross-polar patterns.

to dual gridded reflectors, which are characterized by a very-lowcross polarization. The losses in the reflectarray (0.35 dB) areslightly higher than in the dual gridded reflector (0.10 dB forthe front shell and 0.20 dB for the rear shell).

The total mass of the elliptical reflectarray panel, includingelectrical and structural sandwiches, is 2.250 kg, i.e., 2.7 kg/m .This value represents a significant reduction in mass with re-spect to the dual gridded antenna, with a mass of 4.3 kg/m forthe two-shell configuration, excluding attachments fittings andthe blades.

In conclusion, reflectarray RF performances are slightly infe-rior than those associated to a traditional dual gridded reflectorantenna. On the other hand, reflectarray technology allows sig-nificant reductions in volume, weight, cost, and manufacturingtime, mainly because the electrical requirements are met with asingle reflecting surface.

The reflectarray antenna has demonstrated its ability torealize two linearly polarized independent beams, with a sat-isfactory cross-polarization level. For both polarizations, the

performances remain stable over a 10% frequency band. Theseresults show that the well-known bandwidth limitation in largereflectarrays can be partially overcome by the proposed designtechnique. The potential interest of the reflectarray technologyfor dual polarization antennas in space applications has beendemonstrated.

ACKNOWLEDGMENTThe authors would like to thank G. Toso and C. Mangenot

from ESA-ESTEC, for providing valuable technical comments.

REFERENCES[1] D. M. Pozar, S. D. Targonski, and R. Pokuls, A shaped-beam mi-

crostrip patch reflectarray, IEEE Trans. Antennas Propag., vol. 47, no.7, pp. 11671173, Jul. 1999.

[2] D. M. Pozar and T. A. Metzler, Analysis of a reflectarray antennausing microstrip patches of variable size, Electron. Lett., vol. 29, no.8, pp. 657658, Apr. 1993.

[3] J. A. Encinar, Printed Circuit Technology Multi-Layer Planar Re-flector and Method for the Design Thereof, European Patent EP 1 120856 A1, Jun. 1999.

[4] , Design of two-layer printed reflectarrays using patches ofvariable size, IEEE Trans. Antennas Propag., vol. 49, no. 10, pp.14031410, Oct. 2001.

[5] J. A. Encinar and J. A. Zornoza, Broadband design of three-layerprinted reflectarrays, IEEE Trans. Antennas Propag., vol. 51, no. 7,pp. 16621664, Jul. 2003.

[6] , Three-layer printed reflectarrays for contoured beam spaceapplications, IEEE Trans. Antennas Propag., vol. 52, no. 5, pp.11381148, May 2004.

[7] J. R. Profera and E. Charles, Reflectarray Antenna for CommunicationSatellite Frequency Re-Use Applications, Patent US5543809, Aug.1996.

[8] D. Pilz and W. Menzel, Printed millimeter-wave reflectarrays, Ann.Telecommun., vol. 56, no. 12, pp. 5160, 2001.

[9] O. Bucci, G. Franceschetti, G. Mazzarella, and G. Panariello, Inter-section approach to array pattern synthesis, Proc. Inst. Elec. Eng., vol.137, no. 6, pt. H, pp. 349357, Dec. 1990.

[10] J. A. Zornoza and J. A. Encinar, Efficient phase-only synthesis of con-toured beam patterns for very large reflectarrays, Int. J. RF and Mi-crowave Computer-Aided Eng., pp. 415423, Sep. 2004.

[11] J. A. Zornoza, M. Arrebola, and J. A. Encinar, Multi-frequency pat-tern synthesis for contoured beam reflectarrays, in Proc. 26th AntennaWorkshop on Satellite Antenna Modeling and Design Tools, Noord-wijk, The Netherlands, Nov. 2003, pp. 337342, ESTEC.

[12] C. Wan and J. A. Encinar, Efficient computation of generalizedscattering matrix for analyzing multilayered periodic structures,IEEE Trans. Antennas Propag., vol. 43, pp. 12331242, Nov. 1995.

[13] F. C. Smith, Effective permittivity of dielectric honeycomb, Proc.Inst. Elec. Eng.: Microw. Antennas Propag., vol. 146, no. 1, pp. 5559,Feb. 1999.

Jos A. Encinar (S81M86) was born in Madrid,Spain. He received the Electrical Engineer andPh.D. degrees, both from Universidad Politcnica deMadrid (UPM), in 1979 and 1985, respectively.

Since January 1980, has been with the AppliedElectromagnetism and Microwaves Group at UPM,as a Teaching and Research Assistant from 1980to 1982, as an Assistant Professor from 1983 to1986, and as Associate Professor from 1986 to1991. From February to October of 1987, he waswith the Polytechnic University, Brooklyn, NY, as a

Postdoctoral Fellow of the NATO Science Program. Since 1991, he has beena Professor of the Electromagnetism and Circuit Theory Department at UPM.During 1996, he was with the Laboratory of Electromagnetics and Acousticsat Ecole Polytechnique Fdrale de Lausanne (EPFL), Lausanne, Switzerland,as Visiting Professor. His research interests include numerical techniques forthe analysis of multilayer periodic structures, design of frequency selectivesurfaces, printed arrays and reflectarrays. He has published more than onehundred journal and conference papers, and is holder of three patents on arrayand reflectarray antennas.

Prof. Encinar was a corecipient of the 2005 H. A. Wheeler Applications PrizePaper Award given by IEEE Antennas and Propagation Society.


Leri Sh. Datashvili was born in Aspindza, Republicof Georgia. He received both the Civil Engineer andPh.D. degrees, from Georgian Technical University(GTU), Tbilisi, Georgia, in 1985 and 1997, respec-tively.

From January 1986 till April 2002, he hasbeen working at the Georgian Institute for SpaceConstructions (GISC), Tbilisi, as Junior ResearchAssistant, Research Assistant, Senior ResearchAssistant, and Vice Director General, and Vice ChiefDesigner. From 1987 to September 2001, he has

been at the Institute of Building and Special Constructions of GTU as anTeaching Assistant, Lecturer, Assistant Professor. From June to October 1996,he was at Daimler-Benz Aerospace Dornier Sattelitensisteme GmbH, Mechan-ical Systems department as a visiting scientist. From January to July 1999, haswas a Manager of the large deployable reflector (LDR) development projectfor space orbital technical experiment reflector held by RSC Energia andGISC on a MIR space station on 2328 July 1999. He is a coauthor of theflown reflector fully developed and created in Georgia. From October 1999 toJune 2000, he has been ESA Consultant on LDRs. From October to December2000German Academic Exchange Service (DAAD) funded visiting scientistat the Institute of Lightweight Structures of the Technical University of Munich(LLB, TUM), Munich, Germany. Since April 2002, he has been a ResearchAssociate and Group Leader at LLB, TUM. His research areas of interestinclude developments of LDRs for space applications, membrane structures,high-precision structures, composite materials and structures, multilayersandwich structures, all with design, numerical analyses and experimentalinvestigations. He has published over 60 journal and conference papers and isan author of 16 inventions.

In 1999 and 2001, respectively, Dr. Datashvili was awarded the Georgia StateOrder of Vakhtang Gorgasalli III and the Georgia State Order of Honor forachievements in his research.

J. Agustn Zornoza received both the ElectricalEngineering and Doctor Engineering degrees fromUniversidad Politcnica de Madrid (UPM), Madrid,Spain, in 1999 and 2004, respectively.

From 1998 to 2004, he worked at the Departmentof Electromagnetism and Circuit Theory at UPM, ini-tially as a collaborator and afterwards as researcher.As part of his Ph.D. training, he spent six monthsat Universit Federico II di Napoli, Naples, Italy, in2001, visited the University of Queensland, Brisbane,Australia, from August to December 2002, and en-

joyed a three-month stay at the University of Ulm, Ulm, Germany, in 2003.During that period, his research interest encompassed analysis and design tech-niques for multilayer printed antennas, numerical techniques applied to arraypower pattern synthesis, multibeam and shaped-beam microstrip reflectarrays,and analysis of finite arrays. In June 2004, he joined EADS Astrium Ltd., wherehis current areas of interest include design and synthesis of reflectors antennasfor satellite telecommunications, both onboard and ground tracking systems.

Dr. Zornoza is the corecipient of the 2005 H. A. Wheeler Applications PrizePaper Award given by the IEEE Antennas and Propagation Society.

Manuel Arrebola was born in Lucena, Crdoba,Spain, in 1978. He received the Ingeniero de Tele-comunicacin degree from Universidad de Mlaga(UMA), Mlaga, Spain, in 2002, and he is currentlyworking toward the Ph.D. degree at UniversidadPolitcnica de Madrid (UPM), Madrid, Spain.

Since 2003, he has been with the Electromag-netism and Circuit Theory Department, UPM. FromAugust to December 2005, he was with the Mi-crowave Techniques Department at Universitt Ulm,Ulm, Germany. His current research interests include

multibeam and multifeed reflectarrays and reflectarrays for DBS applications.

Manuel Sierra-Castaer (M01) was born in 1970in Zaragoza (Spain). He received the degree of En-gineer of Telecommunication in 1994 and the Ph.D.degree in 2000, both from the Technical Universityof Madrid (UPM), Madrid, Spain.

Since 1997, he has been in the University AlfonsoX as Teaching Assistant, and since 1998 at the Poly-technic University of Madrid as a Research Assis-tant, Assistant, and Associate Professor. His currentresearch interests are in planar antennas and antennameasurement systems.

Jos Luis Besada-Sanmartn was born in Pon-tevedra, Spain, in 1949. He received the degreeof Engineer of Telecommunication in 1971 andthe Ph.D. degree in 1979, both from the TechnicalUniversity of Madrid (UPM), Madrid, Spain.

Since 1971, he has been with the Technical Univer-sity of Madrid, and since 1987 has been a Full Pro-fessor in the Signals, Systems and Radio Communi-cation Department of UPM. His current research in-terests are in reflector antennas design and manufac-turing and antenna measurement systems.

Horst Baier graduated in mechanical engineeringand received the Ph.D. degree in 1977 from theTechnical University of Darmstadt (TU Darmstadt),Darmstadt, Germany.

From 1972 to 1977, has was a Research As-sistant at TU Darmstadt (working on methods,composites, optimization). From 1977 to 1997,he was with Dornier Satellitensysteme (AstriumFriedrichshafen), later was Head of StructuralMechanics and Technology, Chief Engineer Me-chanical Systems, with involvement in different

space hardware projects and technology studies. He held a university teachingposition and honorary professorship at TU Darmstadt during 19811997Since 1997, he has been a Professor and Head of the Institute of LightweightStructures at Aerospace Department of the Technical University of Munich. Hisresearch activities include composite and precision (including smart) structures,space structures and satellite reflectors, cryogenic structures, multidisciplinaryanalysis, and optimization. He is an author of two books and has publishedover 150 journal and conference papers.

Dr, Baier is a member of different national and international engineering sci-ence organizations.

Herv Legay was born in 1965. He received the elec-trical engineering degree and the Ph.D. degree fromthe National Institute of Applied Sciences (INSA),Rennes, France, in 1988 and 1991, respectively.

For two years, he was a Postdoctoral Fellow atthe University of Manitoba, Winnipeg, Canada,where he developed innovating planar antennas. Hejoined Alcatel Space, Toulouse, France, in 1994. Heinitially conducted studies in the areas of militarytelecommunication advanced antennas and antennaprocessing. He currently leads research projects in

integrated Front Ends and Reflectarray antennas and coordinates the collab-orations with academic and research partners in the area of antennas. He ismember of the Alcatel Technical Academy.

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