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IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, VOL. 25 , NO. 2, MARCHIAPRIL 1989 257

High-Performance Direct Torque Control of anInduction Motor

Abstract-A new direct torque control metho d for an induction moto ris presented which is quite different from field-oriented contr ol. Improv-ing the torque response of a large-capacity induction moto r using two setsof three-phase inverters and an open-delta induction motor is of specialconcern. Instantaneous voltage vectors applied by an inverter haveredundancy characteristics which provide som e flexibility fo r selecting theinverter switching modes. By using this switching freedom, control isachieved according to the following priorities; 1) high-speed torquecontrol, 2) regulation of the primary flux, 3) decreasing the zero phasesequence current, 4) minimization of the inverter switching frequency.Simula tions and experiments have been carried out to verify the feasibilityof this priority control, accompanied by comparisons with anothercontrol scheme. Torque frequency-response corner frequencies above2000 Hz have been experimentally measured, and time constan ts of 4 mshave been achieved for rotor speed step responses from - 500 to 50 0 r/min. The peak transient torque during the step change is about 20 timesthe rated torque. The proposed method is very promising fo r rapid torquecontrol.

I. INTRODUCTIONIGH DYNAMIC performance of servo motor drives isH ndispensable in many applications of todaysautomatically controlled machines. AC servo motor controlhas attracted much attention recently in the power electronicsfield. Field-oriented control has been developed, enabling anac motor to attain dynamic responses as rapid as for a dcmotor. The principle of field-oriented control is based onFlemings law, which describes the interaction force betweenfluxes and currents.Fig. l(a) shows a typical system configuration employing

field-oriented control. The system usually employs a positionsensor to drive a rotating reference frame transformation,which generates the phase current commands for the current-controlled inverter. The primary current reference i* iscalculated from the flux command $2* and the torquecommand T* by using an estimator. The control equation forthe estimator contains motor parameters which vary with thewinding temperatures and the flux saturation level of the ironcore. Many papers have reported the problems associated withcompensating these parameters. The current-controlled invert-ers typically used in the field-oriented drive system developPaper IPCSD 88-2, approved by the Industrial Drives Committee of theIEEE Industry Applications Society for presentation at the 1987 IndustryApplications Society Annual Meeting, Atlanta, G A, October 19-23. Manu-

script released for publication May 13, 1988.I. Takahashi is with the Department of Electrical and Electronic SystemEngineering, Nagaoka University of Technology, 1603-1 Kamitomioka,Nagaoka, Niigata, Japan 940-2 1.Y. Ohmori is with the Technical Development Center, Toyo ElectricCompany, Ltd., 388-1 Kamikusayanagi, Yamato, Japan 242.IEEE Log Number 8825305.

Fig. 1

iA(b )

and flux control.Torque control schem es. (a) Field-oriented control. (b) Direct torque

output waveforms which do not compare favorably with thoseof the voltage-controlled inverter. The current-controlledinverter often causes increased motor harmonic losses andacoustic noise during steady-state operation [11431.

This paper proposes new control schemes based on theprinciple of Aragos disk, which can be considered a basiclaw of torque generation in the induction motor. It makespossible both fast torque response and high-efficiency controlat the same time.Fig. l(b) shows a schematic diagram of the proposedcontrol scheme. In the system, instantaneous values of the fluxand the torque are calculated from primary variables andcontrolled independently by using an optimum switching table.Therefore, it can achieve not only the fastest torque responsebut also the lowest harmonic losses and acoustic noise [4].An object of the paper is how to apply the theory to a large-capacity servo system using a gate turn-off (GTO) inverter.The switching frequency of a GTO inverter is restricted tovalues no higher than several hundred hertz. To increase thedrive capacity while decreasing the switching frequency, twosets of three-phase GTO inverters are employed in theexperiment.

11. INDUCTIONOTOR DRIVEY A DOUBLEHREE-PHASEINVERTERFig. 2 shows a schematic diagram of the power circuit,which is composed of two sets of three-phase inverters, areactor L for suppressing the zero phase sequence current, andan open-delta connection induction motor ( IM). Since neutral

points of the induction motor are not isolated, zero phasesequence current, expressed as io = i, + ib + ic, exists. Thecurrent io increases the copper losses as well as the requiredinverter capacity. These effects can be limited somewhat bythe primary leakage impedances of the motor, but the effect isOO93-9994/89/03OO-0257$01 OO 0 989 IEEE

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2 5 8 IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, VOL. 25 , NO. 2 , MARCHIAPRIL 1989-Iig. 2. Schematic d iagram of system.so small that the zero phase sequence reactor with isolatedthree windings must be employed.The instantaneous voltage vector u I is represented asfollows:

U ]= J2/3 (Uu - , / 2 - 4 2 ) jJ2/3(u, - ,)}. (1)Assuming a dc link voltage of E, the line-to-line terminalvoltages uu, U,, U, applied by the inverters have three levels: E,

0, or - E . As shown in Fig. 3, there are 19 distinct vectorsets, including one zero voltage vector V O , ix intermediateamplitude vectors VI, V2, * * , v,, nd 12 large-amplitudevectors V ,, V8, * * . VIS. Since the total number of inverterswitch combinations is 2 , = 64, some of the switching modesoverlap, delivering the same voltage vector. This switchingredundancy can be employed to decrease the zero phasesequence current.

111. DIRECTONTROLF BOTHTORQUEN D PRIMARYLUXThe primary voltage vector u1 for the symmetrical inductionmotor is expressed by following equation:

u1= R I l+ d$ , d t (2)where RI is the stator resistance, is the flux linkage of thestator winding, and il is the input current vector expressed as

il =J2/3{(iu-i,/2-i,/2)+jJ2/3(i,-i,)}. (3 )The vector u I changes discretely at the switching instants butstays constant between the switching intervals. Therefore,from (2), during each interval is

G l = V l t - ! R l i l d t + $ l o (4)where ql0 s the initial value of at the beginning of theswitching interval. Assuming the voltage drop of R I s small,the trajectory of moves with constant speed approaching thesame orientation as uI . The speed is almost proportional to theamplitude of U,.Fig. 4shows the relationship between the locus of andselected voltage vectors to follow a circular reference I *.The selection of V(S , , S , , S ,) is made to retain the fluxamplitude error within the specified limits set by a hysteresisband of width A l $ l l , i.e.,

l ~ l l * - ~ l ~ l l ~ ~ ~ l ~ ~ l ~ l ~ l l * + ~ l $ l l ~ ~ .The voltage vector selection algorithm depends not only onthe amplitude error but also on the direction of G I . As shownin Fig. 4, the angle between the voltage vectors at switching

d l

Fig. 4 . Locus of $, and voltage vectors

instants is always a16 rad, which eliminates unnecessaryinverter switching.The Fig. 4 also shows that the flux vector locus changesdirection periodically by d6-rad steps. Considering theinherent symmetry, the switching plane pattern is divided into12 segments as follows:(d6)(n - 1)

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259TAKAHASHI AND OHMORI: DIRECT TORQUE CONTROL OF INDUCTION MOTOR

T ws=2n*25

time (ms )Fig. 5. Torque response for step change of U ,.

tirre(m5 )Fig. 6. Relations between torque and voltage vector groups.

Assuming that the speed response is very slow in compari-son with the torque response, the fastest response can beobtained by changing the slip frequency as much as possible.This means that the maximum rate of increase for thetorque, i.e., the minimum torque response time, is achievedby controlling the inverter maximum frequency under thecondition of constant 1 .Fig. 6 shows the torque control scheme based on thisprinciple. When the torque T increases and reaches the upperhysteresis limit T * + T l / 2 , t is better to decrease T as slowlyas possible to reduce the inverter switching frequency. Theslowest decay rate for the torque can be obtained by using theintermediate amplitude vector group VI , - , V, at ratherhigh-speed operation and the zero vector V, at low-speedoperation. These vectors are selected until the torque Treaches the lower hysteresis limit T * - A T l / 2 . Using thistechnique, the torque error is limited within A T .Fig. 7 shows the method for choosing the desired voltagevector of the large-amplitude group (LG) and the middle-amplitude group (IG), or the middle and the zero-amplitudevoltage group (ZG).Fig. 7 is composed for four hysteresis loops. Loop 1,employing LG vectors and IG vectors sequenced to produceclockwise rotation, is used during clockwise high-speedoperation or at times when large clockwise impulse torque isrequired.

c l o c kw is e

c o u n t ec l o c kwise r:g.l

Fig. 7 . Four-loop hysteresis comparator.

Fig. 8. Voltage vector to generate desired flux.

The IG and ZG vector loops are mainly used at low speedsor steady-state operation. Switching between the hysteresisloops occurs automatically by detecting the saturation of theloops. When loop 2 saturates at its upper limits, the compara-tor detects a saturation level A T 2 / 2 which is slightly largerthan A T l / 2 and causes the control to switch to loop 1.Similarly, when loop 1 is undersaturated, the loop is switchedover to loop 2 .The selection of the inverter switching elements is designedto achieve the following performance objectives, arranged inorder of control priority (1 is highest):1) fast dynamic torque control,2 ) regulation of the primary flux linkage,3) reduction of the zero phase sequence current, and4) minimization of the inverter switching frequency.

IV. SWITCHINGF TH E PWM INVERTERIn this section, the algorithm for selecting the voltage vectorwhich controls both torque and flux simultaneously is de-scribed. Fig. 8 shows the relation of the desired vector u I togenerate flux $, and several voltage vectors V 2, V8, V3,and

VI, which are neighbors of vector VI. From (2), since thedesired voltage vector is advanced by 90, $I can be rotatedclockwisely by using one of the vectors V,, V3, V8,or v15.Vectors V 3 and VI5 can decrease the amplitude of IG1l withtime, whereas vectors V2 and V8 ncrease 1 . The rotatingspeed 0 of the flux is minimized by selecting Vo, hasintermediate values using v2 or V3, nd is maximized by v8 orVIS.Therefore, in the case of small flux rotating speeds e whendecreasing torque is required, the optimum voltage vectorwould be Vo. In the case when both torque and flux arerequired to increase, the optimum voltage vector is V2.Whenincreasing torque and decreasing flux are required, the

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260 IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, VOL. 2 5 , NO . 2, MARCHIAPRIL 1989t v-

O2 02 11 01 10 00 1

-1 0-1 1-2 0-2 1

7

-I~@-2

Fig. 9. Optimum switching table and controller.optimum voltage vector is V ,. As mentioned before, in thecase of large e , the inverter switching is done between the Vz,V3 group and the V , , V I ,group. On the other hand, for smalle , the inverter switching is done between the Vz, V3group andthe VO roup. Thus the optimum switching can be uniquelyselected by using the methods already described.Fig. 9shows the optimum voltage vector table. The torqueerror T* - T and the flux error I I are digitized byfour-loop and one-loop hysteresis comparators, respectively.A 3-bit signal T, a 1-bit signal 4, and a 4-bit signal 8, aredeveloped. 0, is the angle signal for which divides theswitching plane into 12 areas.The switching mode redundancy provided by the invertercan be used as previously mentioned for suppressing the zerophase sequence current permitted by the open-delta connec-tion. That is, some of the voltage vectors can be produced byseveral different switch mode combinations, leaving the volt-age vector unchanged. The zero phase sequence voltage uodeveloped by the inverters is given by

uO= U,+ ub+ U,. ( 6 )

I * - I

Then the zero phase sequence current io s represented asuo = (Ro+Rl)io+ (1 + 3L0)di0/dt (7)

where Ro and Lo are the resistance and the reactance of thezero phase reactor. Equations ( 6 ) and (7) indicate that theincrease or decrease of io is determined by the polarity of uo.Fig. 10 shows a switching table for suppressing the zerophase sequence current. By using a three-level comparator, iois compared with the limits % Aio setting the comparatoroutput Io o 1 0, or - 1. In the case of Io = 1, a negative valueof Vo is desired to reduce io, but the minimum number ofswitch mode changes is also desired to minimize the averageswitching frequency.For example, under the condition of VO, odd and Io = - 1

It Io

Fig. 10. Switching table for decreasing io.in Fig. 10, iowould be increased most rapidly by selecting thetable (E,E, E ) , i.e., uo = 3E rather than (0 ,0, 0), i.e., uo =0. However, due to the high voltage of uo , the rate of the iocurrent increase is very large in this case. Therefore, to reducethe switching frequency, it is better to use the vector (0 ,0, 0),i.e., uo = 0 rather than the vector ( E , E , E ) . n this case, io iskept almost constant until the next switching mode. Thesuppression of io within the desired limits is not alwaysrealized during normal operation due to its low controlpriority.As shown in Fig. 3, the switching redundancies of Voandthe VI, * , V6 group are three and two, respectively. Ifswitching to decrease the zero phase sequence current isrequired, the inverter must be changed to another switchingpattern while retaining the same voltage vector. Since voltagevectors in the VI3, , VI, group have one switchingredundancy, switching to decrease io requires that the voltagevector be switched to the neighboring group V ,, * * , VIZ,which has uo = 0, detecting the angle of 8,. The prioritycontrols for generating optimum torque, flux, and zero phasesequence current uniquely select one voltage vector at everytime instant.The next task is to select the switching condition of theinverters for a given voltage vector. It can be considered thattwo sets of three-phase inverters in Fig. 2 are equivalent tothree sets of single-phase bridge inverters. For example, whenthe line-to-line voltage U, = E, or -E , the switchingcondition (SI, ib) in Fig. 2 is uniquely determined as (1, 0)and (0, l), respectively. When U, = 0, however, twoswitching modes (0, 0) and (1, 1) can be selected. Forequalizing the switching frequency of each device, it isdesirable to select modes (0 ,0), and (1, 1) alternately.Fig. 11 shows the switching frequency equalization circuitalready described. A separate equalization circuit is necessaryfor each individual phase. In this figure, the input signal u,*(x= a, b, c) represents phase voltage reference for the bridgeinverter. To get the desired control, the previous switching

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TAKAHASHI AN D OHMORI: DIRECT TORQUE CONTROL OF INDUCTION MOTOR 2 6 1r

s2xlatchsigMl

++D *dont care ( x - a , b , c )mFig. 11. Switching frequency equalizing circuit.

signal (SIX,Sk) nd the signal Soxare necessary. Sox ndicateswhether the changed signal during the previous cycle was SIXor S,. When the changed signal was SIX, Sox= 1, and whenthe changed signal was S,, Sox= 0.Experimental Systemsmotor can be written asUsing vector notation, the torque of the two-pole induction

T = $ , * ( - j i l ) . (8)This equation can be rewritten using d-q axis components as

T =$I& - l q i l d (9)where

$Id= j ( U l d - R l i l d ) dt$ l q = j ( u I ~ - R I ~ I ~ )t. (10)

A flux estimator using simple op-amp integrators tocalculate and $ l q has high drift levels for low-frequencyoperation. In [4], a more precise flux observer was developed.1 and torque Terrors aredigitized using a hysteresis comparator and a five-levelhysteresis comparator, respectively. The angle 8, of is alsodigitized into 12 sectors as shown in Table I.Fig. 12 represents the total configuration of the proposedcontrol system. The three-phase line-to-line voltages u I and theline current i l are transformed into d-q components. Using(lo), the primary flux linkages are estimated usingintegrators. The amplitude 1 $ 1 1 and its sector angle 8, aredetermined from the following equation and Table I, respec-tively:

The measured flux amplitudeThree bits of 7, ne bit of 4, 4 bits of 8, and 2 bits of IOareused to address a I-kbit RO M table. In this figure, the signal 7

is determined by the four-stage hysteresis loop shown in Fig.7.The output of the RO M is not the switching condition of theinverter but the line-to-line voltage references U;, U:, and U,*.The switching condition of the individual inverter is deter-mined by using the circuit shown in Fig. 11.

V. FREQUENCYESPONSE OF TORQUEFig. 13 shows the experimental results for a torque stepresponse. A response time of 30 0 ps for a step change of -

to 9 N*m (rated torque) is observed. From ( 5 ) , the rate oftorque increase is approximately expressed asdT/dt= (PM2 $ I I 2 ~ s ) / L I I(LIlL22-M2) / 2 . (12)

Fig. 12. Configuration of proposed system.

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2 6 2

-w---rnCI

v

tM 0-.0 -r(N

v

P

Fig. 13 . Experimental result of torque step response

- -_ _--~. r a t e d torque

- ---..

I; fc-ZkkhI

H

TABLE I1INDUCTION MOTORPARAMETERS

IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, VOL. 25 , NO . 2, MARCHIAPRIL 1989

Rated power 1.5 kWL , , = L22 = 87.5 mH M = 83.8 mHE = 273 V = 0. 5 Wb

Number of pole P = 4

In the case of locked rotor conditions, the slip frequency W, isidentical to the rotating speed of the primary flux.Under these conditions U, is given byws= J 2 / 3 E / I ~1 I. (13)

Experiments were carried out to verify the feasibility of theproposed control using a 1.5-kW three-phase induction motor.The maximum forcing voltage applied to the motor is 2.9times the rated voltage. The machine constants are listed inTable 11. Substituting these parameters into (12) and (13), acalculated value of dT/dt = 5.6 x lo4N.m/s is obtained forthe locked rotor condition. This value corresponds to 322 ps,which is almost the same as the step response measured fromthe experimental results.Fig. 14 shows the sinusoidal response of the torque withfrequency (fr). Fig. 14(a) is the case forfr = 200 Hz, and itslag time is almost zero. Fig. 14(b) is for fr = 1.5 kHz. Theamplitude of the output torque is slightly larger than that of thereference because of the hysteresis torque control.Fig. 15 shows the frequency characteristics of the torque forthe torque reference amplitudes of 13 .5, 8.9, and 4.3 Nem.The cutoff frequency of the system at the rated torque (8.9N*m) s about 2 kHz. These results may represent a worldrecord for such a large induction motor drive system.The mean switching frequency for operation with a torquereference frequency of fr = 5 kHz is slightly higher than 1kHz. Since the inverter has six switching arms, the idealizedswitching pattern for this same operating point would yield asomewhat lower mean switching frequency of 5000/6 = 83 3Hz.Fig. 16 shows a velocity step response obtained from theexperimental drive system. A proportional-integral (P-I)speed controller is used to get the torque reference signal T* ,and the flux reference 1 1* is set at its maximum value, 0.5Wb. The maximum impact torque is 180 N*m, whichcorresponds to 20 times the rated torque. The current in thisinstant is 125 A , which is 12 times the rated current. The zerophase sequence current io is limited within 2.0 A .The - 500 to + 500-r/min velocity step response time of4.0 ms is obtained under no-load conditions. These resultsmay represent the world record fo r induction motors rated at 1kW or more.

(b )liHZ

Fig. 14. Frequency characteristic of torque. (a)fr = 200 H z. ( b ) f r = 1.5

Fig. 16. Step response of speed

Both the harmonic torque and flux are restricted within thehysteresis bands with minimized switching frequency. There-fore, the acoustic noise and the temperature rise of the motor isvery small in comparison with that of a conventional inverterdrive system. The quality of the motor noise is quite differentfrom the case of the conventional PW M inverter drive-motor.It sounds broad-band in nature rather than a miscellaneousmonotone.

VI. COMPARISONITH ANOTHERONTROLThe following control schemes are compared:Scheme 1 (Fig. I7(a)): field-oriented control employing aninstantaneous current-controlled inverter using the same con-

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2 6 3AKAHASHI AND OHMORI: DIRECT TORQUE CONTROL OF INDUCTION MOTOR

-0.8 1 -4 0 1 48 1(C)

Fig. 17 . Comparison with field-oriented control and proposed control. (a)Field-oriented control. (b) Direct torque control by single inverter. (c)Direct torque control by double in verters.figuration inverter as shown in Fig. 2;Scheme 2 (Fig. 7(b)): proposed control employing one setof three-phase PW M inverter having a dc link voltage of E ;Scheme 3 (Fig. 17(c)): proposed control employing theconfiguration as shown in Fig. 2 .

Schemes 1 and 3 are compared under the same switchingfrequency of 600 Hz in steady state. The simulation resultsshow that these speed responses are almost the same. How-ever, the torque ripple and the amplitude of the current of theproposed system is decreased to about 70 percent of that of thefield-oriented control.Schemes 2 and 3 display comparable values of torqueripple, and the speed responses are almost the same for bothcases. However, the switching frequency of scheme 2 is 1.5kHz, which is about 2.5 times the value for scheme 3 .Fig. 18shows the mean switching frequency of switchingdevices for schemes 2 and 3 . These results are compared forthe same torque ripple amplitude and peak line-to-line volt-ages.The switching frequency is almost constant for scheme 3and limited within 600 Hz. This permits the use of large-capacity GTO switching elements without increasing torqueripples and harmonic currents.It has been said that a field-oriented controller can achieveinstantaneous torque response, but this is true only when the

3^'&0. 0 10 20 30 4 0 50

fundamental frequency (Hz)Fig. 18. Mean switch ing frequency under steady state and no loa d. A T , =2. 0 N . m , = 0.55 Wb, Al$ , l = 0.02 Wb, i = 2.0 A.inverter can supply the required current instantaneously togenerate the torque. It is very easy to calculate the optimumvoltage vector using the proposed scheme since the systemuses a voltage source control.

VII. CONCLUSIONA newly developed direct torque control for an inductionmotor using two sets of three-phase inverters has beenpresented. The control scheme is quite different from field-oriented control, because it depends on the concept of the

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264 IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, VOL. 25 , NO. 2, MARCHlAPRIL 1989instantaneous slip frequency control in spite of the magneticforce. This paper especially proposes switching methods for alarge-capacity inverter control system using a priority controlscheme. Through experimental and simulation techniques, thevalidity of the proposed system has been examined. The mainresults obtained in this paper are as follows.1) The adopted dual three-phase inverter can deliver 19different voltage vectors (approximately 2.5 times as many asfor a single bridge) using 64 switch position combinations.The redundancy of the switching combinations is employed bythe priority control.2) The measured frequency response of the torque exceeds 2kHz for the rated torque amplitude. This experimental resultmay represent a world record.

3) Impact torque values of about 20 times the rated torquehave been achieved. These high torques make possible the fastvelocity loop response time of 4 ms for a - 0 0 to + 5 0 0 rlminstep change.4) The switching frequency of the proposed system is 1/2.5of the frequency for the single-bridge inverter under the sameconditions of torque ripple amplitude. It makes use of a GTOinverter drive.These results show that the new control schemes are more

suitable for high-speed servo systems than field-orientedcontrol in all respects.ACKNOWLEDGMENT

The authors would like to express their appreciation to M r.I. Miyashita of Toyo Electric Co. Ltd and the PowerElectronics Laboratory members of the Nagaoka University ofTechnology.REFERENCES

[ I] A. Nabae, K. Otuka, H. Uchino, and R. Kurosawa, An approach toflux control of induction motors operated with variable-frequencypower supply, ZEEE Trans. Znd. A p p l . , vol. IA-16, pp. 342-350,1980.L. J. Garces, Parameter adaption for the speed controlled static ac2]

drive with a squirrel-cage induction motor, ZEEE Trans. Ind. A p p l . ,vol. IA-16, pp. 173-178, MarJApr. 1980.K. B. Nordin, D. W. Novotny, and D. S . Zinger, The influence ofmotor parameter deviations in feedforward field orientation drivesystems, in Conf. Rec. 1984 Ann. Meet. ZEEE Znd. A p p l . Soc.,I. Takahashi and T. Noguchi, A new quick response and highefficiency control strategy of an induction motor, ZEEE Trans. Znd.

[5 ] I . Racz, Dynamic behavior of inverter controlled induction motors,Conf.Rec. 1965 ZFAC, pp. 4B.1-4B.7.161 K. R. Jarda n, S. B . Dewan, and G. R. Slemon, General analysis ofthree-phase inverters, ZEEE Trans. Znd. Gen. A p p l . , vol. IGA-5,

[3]

pp. 525-531.141

A p p l . , vol. IA-22, pp. 820-827, 1986.

pp. 672-679, 1969.

Isao Takahashi (M86) was born in Japan onMarch 10, 1942. He received the B.S. degree in1966 and the Ph.D . degree in 1971, both inelectrical engineering, from the Tokyo Institute ofTechnology, Tokyo, Japan.He was an Assistant Professor at the TokyoInstitute of Technology from 1971 to 1975 and anAssociate Professor at Utunoniva Universitv from1975 to 1978. He w as a Visiting Associate Profes-sor at the University of Wisconsin, Madison, In1982. He is now a Professor in the Department ofElectrical and Electronics System Engineering at Nagaoka University ofTechnology, Japan. His curren t research interests are in high-power optimumcontrol, especially motor drives, pow er active filters, PWM inverter control,high-frequency power systems, flywheel energy storage systems, and powercontrol of an atomic fusion reactor.

Youichi Ohmori was born in Japan on December6 , 19 62. He received the M.S. degree in electricaland electronics system engineering from NagaokaUniversity of Tech nology, Niigata, Japan, in 1986.In 1986, he joined the Toyo Electric Mfg. Co. ,Ltd., Kanagawa Prefecture, Japan, and his currentworks are on the static power converter and motortorque control.