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DESCRIPTION
5v to 3.3v voltage regulator
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_______________General DescriptionThe MAX1626/MAX1627 step-down DC-DC switchingcontrollers provide high efficiency over loads rangingfrom 1mA to more than 2A. A unique current-limited,pulse-frequency-modulated (PFM) control schemeoperates with up to a 100% duty cycle, resulting in verylow dropout voltages. This control scheme eliminatesminimum load requirements and reduces the supplycurrent under light loads to 90µA (versus 2mA to 10mAfor common pulse-width modulation controllers).
These step-down controllers drive an external P-chan-nel MOSFET, allowing design flexibility for applicationsto 12W or higher. Soft-start reduces current surges dur-ing start-up. A high switching frequency (up to 300kHz)and operation in continuous-conduction mode allow theuse of tiny surface-mount inductors. Output capacitorrequirements are also reduced, minimizing PC boardarea and system costs.
The output voltage is preset at 5V or 3.3V for theMAX1626 and adjustable for the MAX1627. Input volt-ages can be up to 16.5V. The MAX1626/MAX1627 arefunctional upgrades for the MAX1649/MAX1651.
________________________ApplicationsPCMCIA Power Supplies
Personal Digital Assistants
Hand-Held Computers
Portable Terminals
Low-Cost Notebook Computer Supplies
5V to 3.3V Green PC Applications
High-Efficiency Step-Down Regulation
Minimum-Component DC-DC Converters
Battery-Powered Applications
____________________________Features♦ Low Dropout Voltage
♦ 100% Maximum Duty Cycle
♦ Soft-Start Limits Start-Up Current
♦ Efficiency >90% (3mA to 2A Loads)
♦ Output Power >12.5W
♦ 90µA Max Quiescent Current
♦ 1µA Max Shutdown Current
♦ Up to 300kHz Switching Frequency
♦ 16.5V Max Input Voltage
♦ Output Voltage: 5V/3.3V (MAX1626)Adjustable (MAX1627)
♦ Current-Limited Control Scheme
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5V/3.3V or Adjustable, 100% Duty-Cycle, High-Efficiency, Step-Down DC-DC Controllers
________________________________________________________________ Maxim Integrated Products 1
MAX1626
V+
CSSHDN
GND
3/5
ON/OFF
PEXT
REF OUT
OUTPUT 3.3V
INPUT 3.3V to 16.5V
__________________Pin Configuration
1
2
3
4
8
7
6
5
GND
EXT
CS
V+REF
SHDN
3/5 (FB)
OUT
( ) ARE FOR MAX1627
SO
TOP VIEW
MAX1626 MAX1627
__________Typical Operating Circuit
19-1075; Rev 0; 6/96
PART
MAX1626C/D
MAX1626ESA
MAX1627C/D 0°C to +70°C
-40°C to +85°C
0°C to +70°C
TEMP. RANGE PIN-PACKAGE
Dice*
8 SO
Dice*
EVALUATION KIT MANUAL
FOLLOWS DATA SHEET
______________Ordering Information
* Dice are tested at TA = +25°C.
For free samples & the latest literature: http://www.maxim-ic.com, or phone 1-800-998-8800
MAX1627ESA -40°C to +85°C 8 SO
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5V/3.3V or Adjustable, 100% Duty-Cycle, High-Efficiency, Step-Down DC-DC Controllers
2 _______________________________________________________________________________________
ABSOLUTE MAXIMUM RATINGS
ELECTRICAL CHARACTERISTICS(V+ = +3V to +16.5V, SHDN = 3/5 = 0V, TA = 0°C to +85°C, unless otherwise noted.)
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functionaloperation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure toabsolute maximum rating conditions for extended periods may affect device reliability.
Supply Voltage, V+ to GND.......................................-0.3V, +17VOUT, FB, 3/5, SHDN, REF, CS, EXT to GND ...-0.3V, (V+ + 0.3V)Maximum Current at REF (IREF) ..........................................15mAMaximum Current at EXT (IEXT) ..........................................50mAContinuous Power Dissipation (TA = +70°C)
SO (derate 5.88mW/°C above +70°C)..........................471mW
Operating Temperature RangeMAX1626ESA/MAX1627ESA ............................-40°C to +85°C
Storage Temperature Range .............................-65°C to +160°CLead Temperature (soldering, 10sec) .............................+300°C
0µA ≤ IREF ≤ 100µA
ILOAD = 0µA
30mA < ILOAD < 2.0A, V+ = 8V
V+ = SHDN = 16.5V (shutdown)
6.0V < V+ < 12.0V, ILOAD = 1A
Operating, no load
Output in regulation
Output forced to 0V
V+ = 5V
3/5 = 0V or V+
MAX1627
Circuit of Figure 1, 3/5 = V+ (Note 1)
MAX1626, 3/5 = V+, output forced to 5V
SHDN = 0V or V+
MAX1627, includes hysteresis
CONDITIONS
mV4 10REF Load Regulation
V1.27 1.30 1.33VREFReference Voltage
mV/A15Load Regulation
mV/V5Line Regulation
%100EXT Duty-Cycle Limit
µs1.5 2.0 2.5
8 10 12Minimum EXT Off Time
Ω10EXT Resistance
µA±13/5 Leakage Current
V0.53/5 Input Voltage Low
VV+ - 0.53/5 Input Voltage High
µA±1SHDN Input Current
V0.4SHDN Input Voltage Low
µA1
I+Supply Current into V+70 90
V3.0 16.5V+Input Voltage Range
V1.6SHDN Input Voltage High
mV85 100 115VCSCS Threshold Voltage
µA0 10CS Input Current
nA0 35FB Leakage Current
V2.7 2.8Undervoltage Lockout
4.85 5.00 5.15
µA24 37 50IOUTOUT Input Current
V1.27 1.30 1.33FB Threshold Voltage
UNITSMIN TYP MAXSYMBOLPARAMETER
V+ = 3V to 16.5V, ILOAD = 0µA µV/V10 100REF Line Regulation
Circuit of Figure 1, 3/5 = 0V (Note 1)V
3.20 3.30 3.40VOUTOutput Voltage
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_______________________________________________________________________________________ 3
ELECTRICAL CHARACTERISTICS(V+ = +3V to +16.5V, SHDN = 3/5 = 0V, TA = -40°C to +85°C, unless otherwise noted.) (Note 2)
Note 1: V+ must exceed VOUT to maintain regulation.Note 2: Specifications from 0°C to -40°C are guaranteed by design, not production tested.
V+ = SHDN = 16.5V (shutdown)
Operating, no load
ILOAD = 0µA
MAX1627
Circuit of Figure 1, 3/5 = V+
MAX1626, 3/5 = V+, output forced to 5V
MAX1627, includes hysteresis
CONDITIONS
µA2
IOUTSupply Current into V+100
V3.0 16.5V+Input Voltage
V1.25 1.35Reference
mV80 120CS Threshold Voltage
nA0 50FB Leakage Current
V2.9Undervoltage Lockout
4.80 5.20
µA24 50IOUTOUT Input Current
V1.25 1.35FB Threshold Voltage
UNITSMIN TYP MAXSYMBOLPARAMETER
Circuit of Figure 1, 3/5 = 0VV
3.16 3.44VOUTOutput Voltage
__________________________________________Typical Operating Characteristics(Circuit of Figure 1, TA = +25°C, unless otherwise noted.)
100
00.1m 100m 11m 10m 10
EFFICIENCY vs. LOAD CURRENT (VOUT = +3.3V)
20
10
MAX
1626
-05
LOAD CURRENT (A)
EFFI
CIEN
CY (%
)
40
30
60
50
70
80
90
A: V+ = +4.3V B: V+ = +5V C: V+ = +8V D: V+ = +10V E: V+ = +12V F: V+ = +15V
CIRCUIT OF FIGURE 1
AB C
D E F
100
00.1m 100m 11m 10m 10
EFFICIENCY vs. LOAD CURRENT (VOUT = +5V)
20
10
MAX
1626
-03
LOAD CURRENT (A)
EFFI
CIEN
CY (%
)
40
30
60
50
70
80
90
A: V+ = +6V B: V+ = +8V C: V+ = +10V D: V+ = +12V E: V+ = +15V CIRCUIT OF FIGURE 1
AB C
D E
0.45
00 0.5 1.0 1.5 2.0 2.5
DROPOUT VOLTAGE vs. LOAD CURRENT
0.05
0.10
0.35
0.25
0.30
0.40
MAX
1626
-11
LOAD (A)
DROP
OUT
VOLT
AGE
(V)
0.20
0.15
3.3V SETTING VOUT = +3.17V
5V SETTING VOUT = +4.8V
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4 _______________________________________________________________________________________
____________________________Typical Operating Characteristics (continued)(Circuit of Figure 1, TA = +25°C, unless otherwise noted.)
0-60 -40
MAX1626 SHUTDOWN CURRENT vs. TEMPERATURE
MAX
1626
-04
TEMPERATURE (°C)
SHUT
DOW
N CU
RREN
T (µ
A)
0.1
0.3
0.2
0.5
0.4
0.6
0.7
0.8
-20 0 20 40 60 80 100 120 140
E
C
B
A
D
APPLICATION CIRCUIT SHUTDOWN CURRENT: A: V+ = +15V B: V+ = +10V C: V+ = +4V MAX1626 SHUTDOWN CURRENT: D: V+ = +16V E: V+ = +4V
400
00 2000 4000
EXT RISE AND FALL TIMES vs. CAPACITANCE
50
100
300
250
350 MAX
1626
-10
CAPACITANCE (pF)
t RIS
E AN
D t F
ALL
(ns)
200
150
tRISE, V+ = +15V
tFALL, V+ = +5V
tRISE, V+ = +5V
tFALL, V+ = +15V
50
0-60 -40 -20 40 60 140
EXT RISE AND FALL TIMES vs. TEMPERATURE
10
5
15
40
35
45
MAX
1626
-09
TEMPERATURE (°C)
t RIS
E AN
D t F
ALL
(ns)
0 20 80
25
20
30
100 120
tFALL, V+ = +5V
CEXT = 1nF 3/5 = 0V OUT = 50kHz, 0.3Vp-p, 3.3VDC
tRISE, V+ = +5V
tRISE, tFALL, V+ = +15V
60-60 -40
MAX1626 V+ QUIESCENT CURRENT
vs. TEMPERATURE
MAX
1626
-01
TEMPERATURE (°C)
I Q (µ
A)
62
64
66
68
70
72
-20 0 20 40 60 80 100 120 140
3/5 = 0V OUT FORCED TO 3.4V
V+ = +16V
V+ = +10V
V+ = +4V
12
00 1 2 3 4 5
MAX1626 EXT OFF TIME vs. OUTPUT VOLTAGE
2
8
10
MAX
1626
-02
OUTPUT VOLTAGE (V)
EXT
OFF
TIM
E (µ
s)
6
4
V+ = +5V
3/5 = GND
3/5 = V+
12
00 0.2 0.4 0.6 0.8 1.0 1.2 1.4
MAX1627 EXT OFF TIME vs. FB PIN VOLTAGE
2
8
10
MAX
1626
-03
FB PIN VOLTAGE (V)
EXT
OFF
TIM
E (µ
s)
6
4
V+ = +5V
115
85-60 -40 -20 0 20 40 60 80 100 120 140
CS TRIP LEVEL vs. TEMPERATURE
90
105
110
MAX
1626
-12
TEMPERATURE (°C)
CS T
RIP
LEVE
L (m
V)
100
95
OUT = 0V
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5V/3.3V or Adjustable, 100% Duty-Cycle, High-Efficiency, Step-Down DC-DC Controllers
_______________________________________________________________________________________ 5
1.310
1.280-60 -40 -20 0 20 40 60 80 100 120 140
REFERENCE OUTPUT VOLTAGE vs. TEMPERATURE
1.285
1.300
1.305M
AX16
26-1
3
TEMPERATURE (°C)
REFE
RENC
E OU
TPUT
VOL
TAGE
(V)
1.295
1.290
IREF = 10µAIREF = 0µA
IREF = 50µA
IREF = 100µA
100µs/div
MAX1626 LOAD-TRANSIENT RESPONSE
MAX
1626
-15
V+ = 8V, VOUT = 3.3V, LOAD = 30mA to 2A A: OUT, 50mV/div, 3.3V DC OFFSET B: LOAD CURRENT, 1A/div
B
A
5ms/div
MAX1626 LINE-TRANSIENT RESPONSE
MAX
1626
-16
VOUT = 5V, LOAD = 1A, CIN = 33µF A: OUT, 100mV/div, 5V DC OFFSET B: V+ 6V to 12V, 2V/div
B
A
5ms/div
LINE-TRANSIENT RESPONSE FROM 100% DUTY CYCLE
MAX
1626
-17
VOUT = 3.3V, LOAD = 1A, CIN = 47µF A: OUT, 100mV/div, 3.3V DC OFFSET B: V+ 3.3V to 15V, 5V/div
B
A
____________________________Typical Operating Characteristics (continued)(Circuit of Figure 1, TA = +25°C, unless otherwise noted.)
500µs/div
MAX1626 SHUTDOWN RESPONSE TIME AND SUPPLY CURRENT
MAX
1626
-14
V+ = 8V, VOUT = 5V, LOAD = 1A A: OUT, 2V/div B: SUPPLY CURRENT, 1A/div C: SHDN, 5V/div
B
C
A
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6 _______________________________________________________________________________________
______________________________________________________________Pin Description
Ground88
1.3V Reference Output. Can source 100µA. Bypass with 0.1µF.44
Positive Supply Input. Bypass with 0.47µF.55
Current-Sense Input. Connect current-sense resistor between V+ and CS. ExternalMOSFET is turned off when the voltage across the resistor equals the current-limittrip level (around 100mV).
66
Gate Drive for External P-Channel MOSFET. EXT swings between V+ and GND.77
Active-High Shutdown Input. Device is placed in shutdown when SHDN is drivenhigh. In shutdown mode, the reference, output, and external MOSFET are turned off.Connect to GND for normal operation.
33
3.3V or 5V Selection. Output voltage is set to 3.3V when this pin is low or 5V when itis high.
—2
Feedback Input for adjustable-output operation. Connect to an external voltagedivider between the output and GND (see the Setting the Output Voltage section).
2—
Sense input for fixed 5V or 3.3V output operation. OUT is internally connected to anon-chip voltage divider (MAX1626). It does not supply current. Leave OUT uncon-nected during adjustable-output operation (MAX1627).
11
GND
REF
V+
CS
EXT
SHDN
3/5
FB
OUT
MAX1626
C5 0.47µF
P
D1
RSENSE 0.04Ω
U1 LOGIC-LEVEL MOSFET
C4 0.1µF
3/5
SHDN
REF
V+
EXT
CS
GND OUT
INPUT
L1 22µH, 3A
C1 220µF LOW-ESR TANTALUM
C2 68µF LOW-ESR
TANTALUM
C3 68µF LOW-ESR TANTALUM
L1: SUMIDA CDRH125-220 D1: NIHON NSQ03A03 U1: MORTOLA MMSF3PO2HD
OUTPUT
Figure 1. MAX1626 Typical Operating Circuit
MAX1626 MAX1627 MINIMUM ON-TIME
ONE-SHOT
S
OUT
(FB)
3/5
R3
R2
R1
SHDN
V+CS
R
TRIGQ
Q
MINIMUM OFF-TIME ONE-SHOT
CURRENT-SENSE COMPARATOR
ERROR COMPARATORREF
1.5V
EXT REF
( ) MAX1627 ONLY MAX1626 ONLY
TRIGQ
Figure 2. Simplified Functional Diagram
PIN
MAX1626FUNCTION
MAX1627NAME
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_______________________________________________________________________________________ 7
_______________Detailed DescriptionThe MAX1626/MAX1627 are step-down DC-DC con-trollers designed primarily for use in portable comput-ers and battery-powered devices. Using an externalMOSFET and current-sense resistor allows design flexi-bility and the improved efficiencies associated withhigh-performance P-channel MOSFETs. A unique, cur-rent-limited, pulse-frequency-modulated (PFM) controlscheme gives these devices excellent efficiency overload ranges up to three decades, while drawing around90µA under no load. This wide dynamic range opti-mizes the MAX1626/MAX1627 for battery-poweredapplications, where load currents can vary consider-ably as individual circuit blocks are turned on and off toconserve energy. Operation to a 100% duty cycleallows the lowest possible dropout voltage, extendingbattery life. High switching frequencies and a simplecircuit topology minimize PC board area and compo-nent costs. Figure 1 shows a typical operating circuitfor the MAX1626.
PFM Control SchemeThe MAX1626/MAX1627 use a proprietary, third-genera-tion, current-limited PFM control scheme. Improvementsinclude a reduced current-sense threshold and operationto a 100% duty cycle. These devices pulse only as need-ed to maintain regulation, resulting in a variable switchingfrequency that increases with the load. This eliminates thecurrent drain associated with constant-frequency pulse-width-modulation (PWM) controllers, caused by switchingthe MOSFET unnecessarily.
When the output voltage is too low, the error compara-tor sets a flip-flop, which turns on the external P-chan-nel MOSFET and begins a switching cycle (Figures 1and 2). As shown in Figure 3, current through theinductor ramps up linearly, storing energy in a magnet-ic field while dumping charge into an output capacitorand servicing the load. When the MOSFET is turned off,the magnetic field collapses, diode D1 turns on, andthe current through the inductor ramps back down,transferring the stored energy to the output capacitorand load. The output capacitor stores energy when theinductor current is high and releases it when the induc-tor current is low.
The MAX1626/MAX1627 use a unique feedback andcontrol system to govern each pulse. When the outputvoltage is too low, the error comparator sets a flip-flop,which turns on the external P-channel MOSFET. TheMOSFET turns off when the current-sense threshold isexceeded or when the output voltage is in regulation. Aone-shot enforces a 2µs minimum on-time, except incurrent limit. The flip-flop resets when the MOSFET
turns off. Otherwise the MOSFET remains on, allowing aduty cycle of up to 100%. This feature ensures the low-est possible dropout. Once the MOSFET is turned off,the minimum off-time comparator keeps it off. The mini-mum off-time is normally 2µs, except when the output issignificantly out of regulation. If the output is low by30% or more, the minimum off-time increases, allowingsoft-start. The error comparator has 0.5% hysteresis forimproved noise immunity.
In the MAX1626, the 3/5 pin selects the output voltage(Figure 2). In the MAX1627, external feedback resistorsat FB adjust the output.
Operating ModesWhen delivering low and medium output currents, theMAX1626/MAX1627 operate in discontinuous-conduc-tion mode. Current through the inductor starts at zero,rises as high as the peak current limit set by the cur-rent- sense resistor, then ramps down to zero duringeach cycle (Figure 3). Although efficiency is still excel-lent, output ripple increases and the switch waveformexhibits ringing. This ringing occurs at the resonant fre-quency of the inductor and stray capacitance, due toresidual energy trapped in the core when the commuta-tion diode (D1 in Figure 1) turns off. It is normal andposes no operational problems.
When delivering high output currents, the MAX1626/MAX1627 operate in continuous-conduction mode(Figure 4). In this mode, current always flows throughthe inductor and never ramps to zero. The control cir-cuit adjusts the switch duty cycle to maintain regulationwithout exceeding the peak switching current set bythe current-sense resistor. This provides reduced out-put ripple and high efficiency.
100% Duty Cycle and DropoutThe MAX1626/MAX1627 operate with a duty cycle upto 100%. This feature extends usable battery life byturning the MOSFET on continuously when the supplyvoltage approaches the output voltage. This servicesthe load when conventional switching regulators withless than 100% duty cycle would fail. Dropout voltageis defined as the difference between the input and out-put voltages when the input is low enough for the out-put to drop out of regulation. Dropout depends on the MOSFET drain-to-source on-resistance, current-senseresistor, and inductor series resistance, and is propor-tional to the load current:
Dropout Voltage =
I x R + R + ROUT DS(ON) SENSE INDUCTOR[ ]
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EXT Drive Voltage RangeEXT swings from V+ to GND and provides the gatedrive for an external P-channel power MOSFET. A high-er supply voltage increases the gate drive to the MOSFET and reduces on-resistance (RDS(ON)). SeeExternal Switching Transistor section.
Quiescent CurrentThe device’s typical quiescent current is 70µA.However, actual applications draw additional current tosupply MOSFET switching currents, OUT pin current, orexternal feedback resistors (if used), and both the diodeand capacitor leakage currents. For example, in the cir-cuit of Figure 1, with V+ at 7V and VOUT at 5V, typicalno-load supply current for the entire circuit is 84µA.When designing a circuit for high-temperature opera-tion, select a Schottky diode with low reverse leakage.
Shutdown ModeWhen SHDN is high, the device enters shutdown mode.In this mode, the feedback and control circuit, reference,and internal biasing circuitry are turned off. EXT goeshigh, turning off the external MOSFET. The shutdownsupply current drops to less than 1µA. SHDN is a logic-level input. Connect SHDN to GND for normal operation.
ReferenceThe 1.3V reference is suitable for driving external loads,such as an analog-to-digital converter. It has a guaran-teed 10mV maximum load regulation while sourcing loadcurrents up to 100µA. The reference is turned off during
shutdown. Bypass the reference with 0.1µF for normaloperation. Place the bypass capacitor within 0.2 inches(5mm) of REF, with a direct trace to GND (Figure 7).
Soft-StartSoft-start reduces stress and transient voltage slumpson the power source. When the output voltage is nearground, the minimum off-time is lengthened to limit peakswitching current. This compensates for reduced nega-tive inductor current slope due to low output voltages.
________________Design InformationSetting the Output Voltage
The MAX1626’s output voltage can be selected to 3.3Vor 5V under logic control by using the 3/5 pin. The 3/5pin requires less than 0.5V to ensure a 3.3V output, ormore than (V+ - 0.5)V to guarantee a 5V output. Thevoltage sense pin (OUT) must be connected to the out-put for the MAX1626.
The MAX1627’s output voltage is set using two resis-tors, R2 and R3 (Figure 5), which form a voltage dividerbetween the output and GND. R2 is given by:
where VREF = 1.3V. Since the input bias current at FBhas a maximum value of 50nA, large values (10kΩ to200kΩ) can be used for R3 with no significant accuracyloss. For 1% error, the current through R2 should be at
R2 = R3 x VVOUT
REF −
1
5V/3.3V or Adjustable, 100% Duty-Cycle, High-Efficiency, Step-Down DC-DC Controllers
8 _______________________________________________________________________________________
10µs/div
CIRCUIT OF FIGURE 1, V+ = 8V, VOUT = 5V, LOAD = 100mA A: MOSFET DRAIN, 5V/div B: OUT, 50mV/div, 5V DC OFFSET C: INDUCTOR CURRENT, 1A/div
B
0AC
A
Figure 3. Discontinuous-Conduction Mode, Light-Load-CurrentWaveform
10µs/div
CIRCUIT OF FIGURE 1, V+ = 8V, VOUT = 5V, LOAD = 1.5A A: MOSFET DRAIN, 5V/div B: OUT, 50mV/div, 5V DC OFFSET C: INDUCTOR CURRENT, 1A/div
B
0AC
A
Figure 4. Continuous-Conduction Mode, Heavy-Load-CurrentWaveform
least 100 times FB’s input bias current. Capacitor CR2is used to compensate the MAX1627 for even switch-ing. Values between 0pF and 330pF work for manyapplications. See the Stability and MAX1627 FeedbackCompensation section for details.
Current-Sense-Resistor SelectionThe current-sense comparator limits the peak switchingcurrent to VCS/RSENSE, where RSENSE is the value ofthe current-sense resistor and VCS is the current-sensethreshold. VCS is typically 100mV, but can range from85mV to 115mV. Minimizing the peak switching currentwill increase efficiency and reduce the size and cost ofexternal components. However, since available outputcurrent is a function of the peak switching current, thepeak current limit must not be set too low.
Set the peak current limit above 1.3 times the maximumload current by setting the current-sense resistor to:
Alternatively, select the current-sense resistor for 5Vand 3.3V output applications using the current-senseresistor graphs in Figures 6a and 6b. The current-senseresistor’s power rating should be 20% higher than:
Standard wire-wound resistors have an inductancehigh enough to degrade performance, and are not rec-ommended. Surface-mount (chip) resistors have verylittle inductance and are well suited for use as current-
sense resistors. Power metal-strip resistors feature1/2W and 1W power dissipation, 1% tolerance, andinductance below 5nH. Resistance values between10mΩ and 500mΩ are available.
Inductor SelectionThe essential parameters for inductor selection areinductance and current rating. The MAX1626/MAX1627operate with a wide range of inductance values. In manyapplications, values between 10µH and 68µH take bestadvantage of the controller’s high switching frequency.
Calculate the minimum inductance value as follows:
where 2µs is the minimum on-time. Inductor valuesbetween two and six times L(MIN) are recommended.
L = V+ - V
(MIN)(MAX) OUT( )
( )
x s
V
RCS MIN
CS
2µR =
V
RPOWER RATING (W)
2
CS
CS MAX( )
R = V
1.3 x ICSCS(MIN)
OUT(MAX)
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5V/3.3V or Adjustable, 100% Duty-Cycle, High-Efficiency, Step-Down DC-DC Controllers
_______________________________________________________________________________________ 9
R3
CR2 R2
FROM OUTPUT
TO FB
Figure 5. Adjustable-Output Operation Using the MAX1627
4.5 5.55.0 6.0 1210 14 16
INPUT VOLTAGE (V)
MAX
IMUM
OUT
PUT
CURR
ENT
(A)
3.0
3.5
2.5
2.0
1.5
1.0
0
0.5
VOUT = 5V RSENSE = 0.03Ω
RSENSE = 0.04Ω
RSENSE = 0.05Ω
RSENSE = 0.1Ω
Figure 6a. MAX1626 5V-Operation Current-Sense ResistorGraph
3.0 4.03.5 4.5 1210 14 16
INPUT VOLTAGE (V)
MAX
IMUM
OUT
PUT
CURR
ENT
(A)
3.0
3.5
2.5
2.0
1.5
1.0
0
0.5
VOUT = 3.3VRSENSE = 0.03Ω
RSENSE = 0.04Ω
RSENSE = 0.05Ω
RSENSE = 0.1Ω
Figure 6b. MAX1626 3.3V-Operation Current-Sense ResistorGraph
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begin continuous-conduction operation at a lower frac-tion of the full load (see Detailed Description). Low-valueinductors may be smaller and less expensive, but theyresult in greater peak current overshoot due to current-sense comparator propagation delay. Peak-currentovershoot reduces efficiency and could cause the exter-nal components’ current ratings to be exceeded.
The inductor’s saturation and heating current ratingsmust be greater than the peak switching current to pre-vent overheating and core saturation. Saturation occurswhen the inductor’s magnetic flux density reaches themaximum level the core can support, and inductancestarts to fall. The heating current rating is the maximumDC current the inductor can sustain without overheating.The peak switching current is the sum of the current limitset by the current-sense resistor and overshoot duringcurrent-sense comparator propagation delay.
1µs is the worst-case current-sense comparator propa-gation delay.
Inductors with a core of ferrite, Kool Mu™, METGLAS™,or equivalent, are recommended. Powder iron coresare not recommended for use with high switching frequencies. For optimum efficiency, the inductor wind-ings’ resistance should be on the order of the current-sense resistance. If necessary, use a toroid, pot-core,
or shielded-core inductor to minimize radiated noise.Table 1 lists inductor types and suppliers for variousapplications.
External Switching TransistorThe MAX1626/MAX1627 drive P-channel enhancement-mode MOSFETs. The EXT output swings from GND tothe voltage at V+. To ensure the MOSFET is fully on,use logic-level or low-threshold MOSFETs when theinput voltage is less than 8V. Tables 1 and 2 list recom-mended suppliers of switching transistors.
Four important parameters for selecting a P-channelMOSFET are drain-to-source breakdown voltage, cur-rent rating, total gate charge (Qg), and RDS(ON). Thedrain-to-source breakdown voltage rating should be atleast a few volts higher than V+. Choose a MOSFETwith a maximum continuous drain current rating higherthan the peak current limit:
The Qg specification should be less than 100nC toensure fast drain voltage rise and fall times, and reducepower losses during transition through the linear region.Qg specifies all of the capacitances associated withcharging the MOSFET gate. EXT pin rise and fall timesvary with different capacitive loads, as shown in theTypical Operating Characteristics. RDS(ON) should beas low as practical to reduce power losses while theMOSFET is on. It should be equal to or less than thecurrent-sense resistor.
I D(MAX LIM MAXCS MAX
SENSEI
V
R) ( )( )
≥ =I = VR
V V 1 s
LPEAKCS
CS
OUT
++ −( ) × µ
5V/3.3V or Adjustable, 100% Duty-Cycle, High-Efficiency, Step-Down DC-DC Controllers
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KOOL Mu is a trademark of Magnetics.METGLAS is a trademark of Allied Signal.
PRODUCTIONMETHOD INDUCTORS CAPACITORS DIODES CURRENT-SENSE
RESISTORS MOSFETS
Surface Mount
AVXTPS series
Sprague595D series
MotorolaMBRS340T3
NihonNSQ series
DaleWSL series
IRC LRC series
Miniature Through-Hole
SumidaRCH875-470M (1.3A)
SanyoOS-CON serieslow-ESR organicsemiconductor
IRCOAR series Motorola
Low-CostThrough-Hole
CoilCraftPCH-45-473 (3.4A)
Motorola1N5817 to1N5823
MotorolaTMOS power MOSFETs
SumidaCDRH125-470 (1.8A)CDRH125-220 (2.2A)
CoilCraftDO3316-473 (1.6A)DO3340-473 (3.8A)
SiliconixLittle Foot series
Motorolamedium-powersurface-mount products
NichiconPL serieslow-ESR electrolytics
United Chemi-ConLXF series
Table 1. Component Selection Guide
Diode SelectionThe MAX1626/MAX1627’s high switching frequencydemands a high-speed rectifier. Schottky diodes, suchas the 1N5817–1N5822 family or surface-mount equiva-lents, are recommended. Ultra-high-speed rectifierswith reverse recovery times around 50ns or faster, suchas the MUR series, are acceptable. Make sure that thediode’s peak current rating exceeds the peak currentlimit set by RSENSE, and that its breakdown voltageexceeds V+. Schottky diodes are preferred for heavyloads due to their low forward voltage, especially inlow-voltage applications. For high-temperature applica-tions, some Schottky diodes may be inadequate due totheir high leakage currents. In such cases, ultra-high-speed rectifiers are recommended, although a Schottkydiode with a higher reverse voltage rating can oftenprovide acceptable performance.
Capacitor SelectionChoose filter capacitors to service input and outputpeak currents with acceptable voltage ripple.Equivalent series resistance (ESR) in the capacitor is amajor contributor to output ripple, so low-ESR capaci-tors are recommended. Sanyo OS-CON capacitors are
best, and low-ESR tantalum capacitors are secondbest. Low-ESR aluminum electrolytic capacitors are tol-erable, but do not use standard aluminum electrolyticcapacitors.
Voltage ripple is the sum of contributions from ESR andthe capacitor value:
To simplify selection, assume initially that two-thirds ofthe ripple results from ESR and one-third results fromcapacitor value. Voltage ripple as a consequence ofESR is approximated by:
Estimate input and output capacitor values for givenvoltage ripple as follows:
where I∆L is the change in inductor current (around0.5IPEAK under moderate loads).
These equations are suitable for initial capacitor selec-tion; final values should be set by testing a prototype orevaluation kit. When using tantalum capacitors, usegood soldering practices to prevent excessive heatfrom damaging the devices and increasing their ESR.Also, ensure that the tantalum capacitors’ surge-currentratings exceed the start-up inrush and peak switchingcurrents.
Pursuing output ripple lower than the error compara-tor’s hysteresis (0.5% of the output voltage) is not prac-tical, since the MAX1626/MAX1627 will switch asneeded, until the output voltage crosses the hysteresisthreshold. Choose an output capacitor with a workingvoltage rating higher than the output voltage.
The input filter capacitor reduces peak currents drawnfrom the power source and reduces noise and voltageripple on V+ and CS, caused by the circuit’s switchingaction. Use a low-ESR capacitor. Two smaller-valuelow-ESR capacitors can be connected in parallel forlower cost. Choose input capacitors with working volt-age ratings higher than the maximum input voltage.
Place a surface-mount ceramic capacitor very close toV+ and GND, as shown in Figure 7. This capacitorbypasses the MAX1626/MAX1627, and prevents spikesand ringing on the power source from obscuring the
CLI
V V
CLI
V VV
V V
INL
RIPPLE CIN IN
OUTL
RIPPLE COUT OUT
IN
IN OUT
=
=−
12
2
12
2
∆
∆
,
,
V RIPPLE,ESR ≈ ( )( )RESR IPEAK
V RIPPLE ≈ + , ,V VRIPPLE ESR RIPPLE C
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Table 2. Component Suppliers
COMPANY PHONE FAX
(803) 946-0690AVX USA or (803) 626-3123
(800) 282-4975Coilcraft USA (847) 639-6400 (847) 639-1469Coiltronics USA (516) 241-7876 (516) 241-9339Dale USA (605) 668-4131 (605) 665-1627International USA (310) 322-3331 (310) 322-3332RectifierIRC USA (512) 992-7900 (512) 992-3377Motorola USA (602) 303-5454 (602) 994-6430Nichicon USA (847) 843-7500 (847) 843-2798
Japan 81-7-5231-8461 81-7-5256-4158Nihon USA (805) 867-2555 (805) 867-2698
Japan 81-3-3494-7411 81-3-3494-7414Sanyo USA (619) 661-6835 (619) 661-1055
Japan 81-7-2070-6306 81-7-2070-1174(408) 988-8000
Siliconix USA or (408) 970-3950(800) 554-5565
Sprague USA (603) 224-1961 (603) 224-1430Sumida USA (847) 956-0666 (847) 956-0702
Japan 81-3-3607-5111 81-3-3607-5144United USA (714) 255-9500 (714) 255-9400Chemi-Con
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recommended. Increase the value as necessary inhigh-power applications.
Bypass REF with 0.1µF. This capacitor should beplaced within 0.2 inches (5mm) of the IC, next to REF,with a direct trace to GND (Figure 7).
Layout ConsiderationsHigh-frequency switching regulators are sensitive to PCboard layout. Poor layout introduces switching noise intothe current and voltage feedback signals, resulting in jit-ter, instability, or degraded performance. The current-sense resistor must be placed within 0.2 inches (5mm)of the controller IC, directly between V+ and CS. Placevoltage feedback resistors (MAX1627) next to the FB pin(no more than 0.2") rather than near the output. Placethe 0.47µF input and 0.1µF reference bypass capacitorswithin 0.2 inches (5mm) of V+ and REF, and routedirectly to GND. Figure 7 shows the recommended lay-out and routing for these components.
High-power traces, highlighted in the Typical OperatingCircuit (Figure 1), should be as short and as wide aspossible. The supply-current loop (formed by C2, C3,RSENSE, U1, L1, and C1) and commutation-current loop(D1, L1, and C1) should be as tight as possible toreduce radiated noise. Place the anode of the commuta-tion diode (D1) and the ground pins of the input andoutput filter capacitors close together, and route them toa common “star-ground” point. Place components androute ground paths so as to prevent high currents fromcausing large voltage gradients between the ground pinof the output filter capacitor, the controller IC, and thereference bypass capacitor. Keep the extra copper onthe component and solder sides of the PC board, ratherthan etching it away, and connect it to ground for use asa pseudo-ground plane. Refer to the MAX1626Evaluation Kit manual for a two-layer PC board example.
Stability and MAX1627 FeedbackCompensation
Use proper PC board layout and recommended exter-nal components to ensure stable operation. In one-shot, sequenced PFM DC-DC converters, instability ismanifested as “Motorboat Instability.” It is usuallycaused by excessive noise on the current or voltagefeedback signals, ground, or reference, due to poor PCboard design or external component selection.Motorboat instability is characterized by groupedswitching pulses with large gaps and excessive low-frequency output ripple. It is normal to see somegrouped switching pulses during the transition fromdiscontinuous to continuous current mode. This effectis associated with small gaps between pulse groups
and output ripple similar to or less than that seen dur-ing no-load conditions.
Instability can also be caused by excessive stray capaci-tance on FB when using the MAX1627. Compensate forthis by adding a 0pF to 330pF feed-forward capacitoracross the upper feedback resistor (R2 in Figure 5).
MAX1626/MAX1627 vs. MAX1649/MAX1651 vs.
MAX649/MAX651The MAX1626/MAX1627 are specialized, third-genera-tion upgrades to the MAX649/MAX651 step-down con-trollers. They feature improved efficiency, a reducedcurrent-sense threshold (100mV), soft-start, and a100% duty cycle for lowest dropout. The MAX649/MAX651 have a two-step (210mV/110mV) current-sense threshold. The MAX1649/MAX1651 are second-generation upgrades with a 96.5% maximum duty cyclefor improved dropout performance and a reduced cur-rent-sense threshold (110mV) for higher efficiency,especially at low input voltages. The MAX1649/MAX1651 are preferable for special applications wherea 100% duty cycle is undesirable, such as flyback andSEPIC circuits.
Since the MAX1626’s pinout is similar to those of theMAX649 and MAX1649 family parts, the MAX1626 canbe substituted (with minor external component valuechanges) into fixed-output mode applications, providedthe PC board layout is adequate. The MAX1627 canalso be substituted when MAX649 or MAX1649 familyparts are used in adjustable mode, but the feedbackresistor values must be changed, since the MAX1627has a lower reference voltage (1.3V vs. 1.5V). Reducethe current-sense resistor value by 50% when substitut-ing for the MAX649 or MAX651.
5V/3.3V or Adjustable, 100% Duty-Cycle, High-Efficiency, Step-Down DC-DC Controllers
12 ______________________________________________________________________________________
MAX1626
yyyyyyyyyyyyyyyyyyyyyyyyyyyyyyyyyyyyyyyyyyyyy
yyyyyyyyyyyyyyyy
yyyyyyyyyyyy
yyyyyy
C REF
C V+ BYPASS
4x SCALE
R SENSE
Figure 7. Recommended Placement and Routing of theCurrent-Sense Resistor, 0.1µF Reference, and 0.47µF InputBypass Capacitors
________________________ApplicationsThe MAX1626/MAX1627 typical operating circuits(Figures 1 and 8) are designed to output 2A at a 5Voutput voltage. The following circuits provide examplesand guidance for other applications.
Micropower Step-Down ConverterWhen designing a low-power, battery-based applica-tion, choose an external MOSFET with low gate capaci-tance (to minimize switching losses), and use a lowpeak current limit to reduce I2R losses. The circuit inFigure 9 is optimized for 0.5A.
High-Current Step-Down ConverterThe circuit in Figure 10 outputs 6A at 2.5V from a 5V or3.3V input. High-current design is difficult, and boardlayout is critical due to radiated noise, switching tran-sients, and voltage gradients on the PC board traces.Figure 11 is a recommended PC board design. Choosethe external MOSFET to minimize RDS(ON). Keep thegate-charge factor below the MAX1626/MAX1627’sdrive capability (see Ext Rise and Fall Times vs.Capacitance graph in the Typical OperatingCharacteristics). Otherwise, increased MOSFET riseand fall times will contribute to efficiency losses. Forhigher efficiencies, especially at low output voltages,the MAX796 family of step-down controllers with syn-chronous rectification is recommended.
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MAX1627
C5 0.47µF
P
D1
RSENSE 0.15Ω
U1 LOGIC-LEVEL MOSFET
C4 0.1µF
OUTN.C.
SHDN
REF
V+
EXT
CS
GND FB
INPUT
L1 22µH, 3A
R3 R2
CR2
C1 220µF LOW-ESR TANTALUM
C2 68µF LOW-ESR
TANTALUM
C3 68µF LOW-ESR
TANTALUM
L1: SUMIDA CDRH125-220 D1: NIHON NSQ03A03 U1: MOTOROLA MMSF3P02HD
ADJUSTABLE OUTPUT
Figure 8. MAX1627 Typical Operating Circuit
MAX1626
C3 0.47µF
P
D1
RSENSE 0.15Ω
U1 LOGIC-LEVEL MOSFET
C4 0.1µF
3/5
SHDN
REF
V+
EXT
CS
GND OUT
INPUT
L1 68µH, 0.7A OUTPUT
C1 100µF LOW-ESR TANTALUM
C2 68µF LOW-ESR
TANTALUM
L1: SUMIDA CDR1053-680 D1: MOTOROLA MBRS130T3 U1: MOTOROLA MMSF3P02HD
Figure 9. 0.5A Step-Down Converter
MAX1627
POUTPUT 2.5V, 6A
R3 21.5k, 1%
R2 20k, 1%
D1
CR2 220pF
Q1 LOGIC-LEVEL MOSFET
C10 0.1µF
OUT
3V TO 6V
SHDN
N.C.
REF
V+
EXT
CS
GND FB
INPUT
L1 2.7µH >8A
C4 100µF
C5 100µF
C9 1.0µF
RCS1, RCS2 0.025Ω
C8 1.0µF
C7 0.1µF
C3 220µF
C2 220µF
C1 220µF
C6 0.1µF
C1–C3: SANYO OS-CON 220µF, 6.3V C4, C5: SANYO OS-CON 100µF, 20V RCS1, RCS2: 0.025Ω DALE WSL-2512 Q1: MOTOROLA MTB50PO3HDL D1: NIEC C10T04Q L1: SUMIDA CDRH127-2R7NC
Figure 10. 6A Step-Down Converter
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VIA
VIA
VIAS
COMPONENT PLACEMENT GUIDE—COMPONENT SIDE COPPER ROUTING—FRONT SIDE
COPPER ROUTING—BACK SIDE
Figure 11. Recommended PC Board Design for 6A Step-Down Converter
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___________________Chip Topography
( ) ARE FOR MAX1627
TRANSISTOR COUNT: 375
SUBSTRATE CONNECTED TO V+
REF
0.105" (2.63mm)
0.081" (2.06mm)
VCC
CS
EXT
GNDGNDOUT
SHDN
3/5 (FB)
DIM
A A1 B C E e H L
MIN 0.053 0.004 0.014 0.007 0.150
0.228 0.016
MAX 0.069 0.010 0.019 0.010 0.157
0.244 0.050
MIN 1.35 0.10 0.35 0.19 3.80
5.80 0.40
MAX 1.75 0.25 0.49 0.25 4.00
6.20 1.27
INCHES MILLIMETERS
21-0041A
Narrow SO SMALL-OUTLINE
PACKAGE (0.150 in.)
DIM
D D D
MIN 0.189 0.337 0.386
MAX 0.197 0.344 0.394
MIN 4.80 8.55 9.80
MAX 5.00 8.75 10.00
INCHES MILLIMETERSPINS
8 14 16
1.270.050
L
0°-8°
HE
D
e
A
A1 C
0.101mm 0.004in.
B
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16 ______________________________________________________________________________________
Recommended