Emi protection for_communication_systems

EMI Protection forCommunication Systems

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EMI Protection forCommunication Systems

Kresimir Malaric

a r techhous e . c om

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Contents

Preface xiii

CHAPTER 1Communications Systems 1

1.1 Components of Communications Systems 11.2 Transmitter Systems 2

1.2.1 Transmitter 31.2.2 Randomization 41.2.3 Encryption 51.2.4 Encoder 51.2.5 Interleaving 91.2.6 Modulation 101.2.7 Mixer (Upconverter) 101.2.8 Filter 11

1.3 Receiver Systems 111.3.1 Filter 111.3.2 Mixer (Downconverter) 121.3.3 Demodulator 121.3.4 Deinterleaver 121.3.5 Decoder 131.3.6 Decryptor 151.3.7 Derandomizer 151.3.8 Demultiplexer 161.3.9 Received Power 16

1.4 User Interface 181.4.1 Graphical User Interface (GUI) 181.4.2 Voice User Interface (VOI) 19

1.5 Antenna Systems 191.5.1 Duplexer 191.5.2 Antenna 20

1.6 Power Supplies 221.6.1 Power Supply Types 231.6.2 Power Amplifier 23

v

1.7 Considerations for Voice Versus Data 231.7.1 Text 231.7.2 Images 241.7.3 Voice 241.7.4 Video 24Selected Bibliography 24

CHAPTER 2Electromagnetic Spectrum Used for Communications 27

2.1 Electromagnetic Spectrum 272.1.1 Extra Low Frequency (ELF) 282.1.2 Super Low Frequency (SLF) 282.1.3 Ultra Low Frequencies (ULF) 292.1.4 Very Low Frequency (VLF) 292.1.5 Low Frequency (LF) 292.1.6 Medium Frequency (MF) 292.1.7 High Frequency (HF) 292.1.8 Very High Frequency (VHF) 292.1.9 Ultra High Frequency (UHF) 292.1.10 Super High Frequency (SHF) 302.1.11 Extra High Frequency (EHF) 302.1.12 Infrared (IR) 302.1.13 Visible 30

2.2 Spectrum Division 30Selected Bibliography 33

CHAPTER 3Electromagnetic Properties of Communications Systems 35

3.1 Fundamental Communications System Electromagnetics 353.1.1 Smith Chart 393.1.2 Snell’s Law of Reflection and Refraction 42

3.2 Wave Generation and Propagation in Free Space 443.2.1 Maxwell’s Equations 443.2.2 Wave Propagation 463.2.3 Wave Polarization 473.2.4 Fresnel Knife-Edge Diffraction 483.2.5 Path Loss Prediction 51

3.3 Wave Generation and Propagation in the Terrestrial Atmosphere 533.3.1 Absorption and Scattering 533.3.2 Wave Propagation in the Atmosphere 54Selected Bibliography 55

CHAPTER 4Electromagnetic Interference 57

4.1 Electromagnetic Interference with Wave Propagation and Reception 574.1.1 Additive White Gaussian Noise (AWGN) 57

vi Contents

4.1.2 Thermal Noise 584.1.3 Shot Noise 584.1.4 Flicker (1/f ) Noise 584.1.5 Burst Noise 594.1.6 Noise Spectral Density 594.1.7 Effective Input Noise Temperature 59

4.2 Natural Sources of Electromagnetic Interference 594.2.1 Lightning and Electrostatic Discharge 594.2.2 Multipath Effects Caused by Surface Feature Diffractionand Attenuation 644.2.3 Attenuation by Atmospheric Water 654.2.4 Attenuation by Atmospheric Pollutants 674.2.5 Sunspot Activity 68

4.3 Manmade Sources of Electromagnetic Interference 694.3.1 Commercial Radio and Telephone Communications 694.3.2 Military Radio and Telephone Communications 744.3.3 Commercial Radar Systems 744.3.4 Industrial Sources 754.3.5 Intentional Interference 76Selected Bibliography 77

CHAPTER 5Filter Interference Control 79

5.1 Filters 795.1.1 Lowpass Filter 805.1.2 Highpass Filter 805.1.3 Bandpass Filter 815.1.4 Bandstop Filter 835.1.5 Resonator 83

5.2 Analog Filters 855.2.1 Butterworth Filter 855.2.2 Chebyshev Filters 865.2.3 Bessel Filters 875.2.4 Elliptic Filters 885.2.5 Passive Filters 885.2.6 Active Filters 91

5.3 Digital Filters 915.3.1 FIR Filters 935.3.2 IIR Filters 94

5.4 Microwave Filters 975.4.1 Lumped-Element Filters 975.4.2 Waveguide Cavity Filters 985.4.3 Dielectric Resonator 100Selected Bibliography 101

Contents vii

CHAPTER 6Modulation Techniques 103

6.1 Signal Processing and Detection 1036.2 Modulation and Demodulation 105

6.2.1 Analog Modulations 1056.2.2 Digital Modulation 112

6.3 Control of System Drift 120Selected Bibliography 120

CHAPTER 7Electromagnetic Field Coupling to Wire 123

7.1 Field-to-Wire Coupling 1237.1.1 Skin Effect 1237.1.2 Unshielded Twisted Pair (UTP) 1257.1.3 Ferrite Filter 126

7.2 Electric Field Coupling to Wires 1287.3 Magnetic Field Coupling to Wires 1317.4 Cable Shielding 132

7.4.1 Tri-Axial Cable 1337.4.2 Cable Termination 1337.4.3 Shielded Twisted Pair Cables 134Selected Bibliography 136

CHAPTER 8Electromagnetic Field-to-Aperture Coupling 137

8.1 Field-to-Aperture Coupling 1378.1.1 Shielding Effectiveness (SE) 1388.1.2 Multiple Apertures 1388.1.3 Waveguides Below Cutoff 140

8.2 Reflection and Transmission 1418.2.1 Electric Field 1458.2.2 Magnetic Field 146

8.3 Equipment Shielding 1478.3.1 Gasketing 1478.3.2 PCB Protection 1488.3.3 Magnetic Shield 149Selected Bibliography 151

CHAPTER 9Electrical Grounding and Bonding 153

9.1 Grounding for Safety 1549.1.1 Shock Control 1549.1.2 Fault Protection 155

9.2 Grounding for Voltage Reference Control 1569.2.1 Floating Ground 1569.2.2 Single Point Ground 157

viii Contents

9.2.3 Multipoint Ground 1589.2.4 Equipotential Plane 158

9.3 Bonding for Current Control 1599.3.1 Bonding Classes 1609.3.2 Strap Bond for Class R 1609.3.3 Resistance Requirements 162

9.4 Types of Electrical Bonds 1629.4.1 Welding and Brazing 1639.4.2 Bolting 1639.4.3 Conductive Adhesive 164

9.5 Galvanic (Dissimilar Metal) Corrosion Control 164Selected Bibliography 166

CHAPTER 10Emissions and Susceptibility—Radiated and Conducted 167

10.1 Control of Emissions and Susceptibility—Radiated and Conducted 16710.1.1 Sources of Electromagnetic Interference 16710.1.2 Test Requirements for Emission and Susceptibility 17110.1.3 Standard Organizations 173

10.2 Commercial Requirements 17710.3 Military Requirements 178

10.3.1 Specific Conducted Emissions Requirements Mil-Std 461E 17810.3.2 Specific Conducted Susceptibility Requirements Mil-Std 461E 17910.3.3 Radiated Emissions Requirements Mil-Std 461E 18110.3.4 Radiated Susceptibility Requirements Mil-Std 461E 182Selected Bibliography 182

CHAPTER 11Measurement Facilities 185

11.1 Full Anechoic and Semianechoic Chambers 18511.1.1 Absorbers 18711.1.2 Ferrite Tiles 189

11.2 Open Area Test Site (OATS) 19111.3 Reverberation Chamber 19311.4 TEM Cell 195

11.4.1 Characteristic Impedance 19611.4.2 Higher-Order Modes 19711.4.3 TEM Cell Construction 19811.4.4 Parameter Measurements 200

11.5 GTEM Cell 20111.5.1 GTEM Cell Characteristics 20311.5.2 GTEM Cell Construction 20311.5.3 GTEM Cell Parameter Measurement 20411.5.4 Current Distribution at Septum 211Selected Bibliography 212

Contents ix

CHAPTER 12Typical Test Equipment 215

12.1 LISN—Line Impedance Stabilization Network 21512.2 Coupling Capacitor 21612.3 Coupling Transformer 21712.4 Parallel Plate for Susceptibility Test 21712.5 Coupling Clamps and Probes 218

12.5.1 Capacitive Coupling Clamp 21912.5.2 Current Probe 220

12.6 Injection Clamps and Probes 22112.6.1 Current Injection Probe 22112.6.2 EM Clamp 22112.6.3 Electrostatic Discharge (ESD) Generator 223

12.7 EMI Receiver 22412.8 Spectrum Analyzer 22512.9 Oscilloscopes 225

Selected Bibliography 225

CHAPTER 13Control of Measurement Uncertainty 227

13.1 Evaluation of Standard Uncertainty 22713.1.1 Type A Evaluation of Standard Uncertainty 22713.1.2 Type B Evaluation of Standard Uncertainty 228

13.2 Distributions 22813.2.1 Normal (Gaussian) Distribution 22913.2.2 Rectangular Distribution 22913.2.3 U-Shaped Distribution 23013.2.4 Combined Standard Uncertainty 23013.2.5 Expanded Uncertainty 231

13.3 Sources of Error 23113.3.1 Stability 23113.3.2 Environment 23113.3.3 Calibration Data 23113.3.4 Resolution 23213.3.5 Device Positioning 23213.3.6 RF Mismatch Error 232

13.4 Definitions 232Selected Bibliography 232

Appendix A Communication Frequency Allocations 235

A.1 Frequency Allocation in the United States 235A.2 International Frequency Allocation 245

x Contents

Appendix B List of EMC Standards Regarding Emission and Susceptibility 255

B.1 Cenelec 255B.2 Australian Standards 256B.3 Canadian Standards 256B.4 European Standards 258B.5 Other Standards 259

Acronyms and Abbreviations 261

Glossary 265

About the Author 267

Index 269

Contents xi

PrefaceCommunication today is not as easy as it was in the past. Protecting numerous com-munication services, which are operating in the same or adjacent communicationchannels, has become increasingly challenging. Communication systems have to beprotected from both natural and manmade interference. Electromagnetic interfer-ence can be radiated or conducted, intentional or unintentional. Understandingphysical characteristics of wave propagation is necessary to comprehend the mecha-nisms of electric and magnetic coupling in the communication signal paths. Differ-ent modulating techniques, as well as encoding and encrypting, can improve biterror rates (BER) and signal quality. Communication systems must be designedproperly, so that the performance of their system capabilities is not subject to degra-dation or complete loss due to electromagnetic interference.

Although there are numerous books available on electromagnetic compatibil-ity, signal processing, and electromagnetic theory, there is no book offering a com-prehensive description of technologies for the protection of communicationsystems, which includes discussions on the improvement of existing communicationsystems and the creation of new systems. The book provides laymen with basicinformation and definitions of problems regarding electromagnetic interference incommunication systems. In addition, it gives an experienced practitioner knowl-edge of how to solve possible problems in both digital and analog communicationsystems. The examples given in the book are intended for an easier comprehensionof otherwise demanding electromagnetic problems. The book’s primary audienceincludes designers, researchers, and graduate students in the area ofcommunications.

The book is organized into 13 chapters dealing with fundamental concerns ofdevelopers and users of communication systems.

• Chapter 1 gives an overview of communication system components.• Chapter 2 deals with the use of the electromagnetic spectrum for communica-

tions.• Chapter 3 describes wave propagation in free space and the terrestrial atmo-

sphere.• Chapter 4 discusses natural sources of electromagnetic interference, such as

attenuation of atmospheric water or lightning, as well as numerous manmadesources of electromagnetic interference.

• Chapter 5 covers analog, digital, and microwave filters.• Chapter 6 deals with signal processing and modulation/demodulation issues

in communication systems.

xiii

• Chapters 7 through 9 are devoted to electromagnetic field-to-wire and aper-ture coupling, as well as to electrical grounding and bonding.

• Chapter 10 gives the commercial and military requirements for radiated andconducted emission and susceptibility. Facilities for EMI measurement suchas TEM and GTEM cells, open area test sites, and reverberation chambers arecovered in Chapter 11.

• Chapter 12 includes the description of coupling capacitors and transformers,coupling and injection clamps and probes, and other test equipment.

• Chapter 13 deals with the control of measurement uncertainties.• Appendix A gives a list of communication frequencies, and Appendix B gives

a list of EMC standards.

The program TEM-GTEM on the CD-ROM accompanying this book works inthe LabVIEW environment on a PC Windows operating system. The TEM-GTEMprogram requires prior installation of LabVIEW Run-Time Engine 8.6 – Windows2000/Vista x64/Vista x86/XP on any computer which does not already haveLabVIEW 8.6 installed. The Run-Time engine can be found on the CD-ROM in thefolder: runtimeengine. The second folder, tem-gtem, contains the tem-gtem.exe aswell as Installation.doc and Instructions.doc file.

The application TEM-GTEM has two programs: TEM and GTEM. The firstprogram, TEM, calculates the characteristic impedanceZ0 (in ohms), and cutoff fre-quencies (fc) for modes TE01, TE10, TE11, TM11, TE02, TE12, TM12, and TE20 depend-ing on the TEM-cell dimensions. The second program, GTEM, calculates the cutoff(fc) and the associated stimulated resonant frequencies (fr1, fr2, and fr3) in mega-hertz for higher-order modes H10, H01, H11, H20, E11, and E21. The resonances arealso shown graphically.

A detailed explanation on the theory for the TEM cell and the GTEM cell can befound in Sections 11.4 and 11.5, respectively.

I wish to thank Artech House editors Lindsey Gendall, Barbara Lovenvirth, andMark Walsh for their encouragement in writing this book. Finally, I thank my wifeBlazenka and my parents Marija and Vladimir for their love and support through-out this project.

xiv Preface

C H A P T E R 1

Communications Systems

1.1 Components of Communications Systems

A communication system usually consists of the information source, transmitter,channel, receiver, antenna systems, amplifier, and end user (Figure 1.1). It convertsinformation into a format appropriate for the transmission medium.

A transmitting antenna’s purpose is to effectively transform the electrical signalinto radiation energy, whereas the receiving antenna’s purpose is to effectivelyreceive the radiated energy and its electric signal transformation for further process-ing at the receiver.

Communication systems can be either analog or digital. Historically, analogsystems (Figure 1.2) are simpler but less resilient to interference. Analog communi-cation systems convert (modulate) analog signals into modulated signals. Signalsthat are analog are converted into digital bits by sampling and quantization, alsocalled digitization and coding.

Digital systems can reconstruct original information, are better protected frominterference, and have the potential to code signals, thus enabling larger amount ofdata transportation.

It is important that the information sent from the source and the informationsent to the end user are similar as possible (i.e., identical in the case of digital infor-mation). Even though digital systems are used more for communications, analogsystems will not be neglected in this and other chapters.

With digital systems, there is a source and channel coder instead of signal pro-cessing in the transmitter, as is the case when dealing with analog systems. In thereceiver, there are also channel and source decoders instead of the signal processingof analog systems.

The modulator and demodulator should be designed to lessen the distortion andnoise from the channel. The channel transports the signals using electromagneticwaves. There is always noise in the channel along with the useful signal.

The source coder (Figure 1.3) converts the analog information to digital bitsusing analog to digital conversion (A/D). The transmitter converts the signal (ana-log) or bits (digital) into a format that is appropriate for channel transmission. Dis-tortion, noise, and interference are brought into the channel. The receiver decodesthe received signal back into the information signal and then the source decoderdecodes the signal back to the original information (analog or digital).

1

1.2 Transmitter Systems

A transmitter system is used to send information/data from one user to another. Thesource information can be analog or digital. Digital systems are usually more com-plex than the analog ones and require more modules. Both the transmitter andreceiver systems should have the same complexity. For example, if we have a modu-lator on the transmitter side, there should be a demodulator on the receiver side.The same applies for the coder, multiplexer, and so forth. Usually, transmitter sys-tems have a multiplexer, randomization, encryption, an encoder, interleaving, amodulator, a mixer (upconverter), a power amplifier, a filter, a duplexer, and a

2 Communications Systems

Transmitter in baseband

Receiver in baseband

User

Source

Processing

Processing

Demodulator

Modulator

RF stage

RF stage

Channel

Figure 1.2 Analog communication system.

Source ofinformation

Transmitter

Channel

Receiver User

Figure 1.1 Communication system.

Source Sourcecoder

Channelcoder

Digitalmodulator

Digitaldemodulator

RF stage

RF stage

Channel

Receiver in baseband

Transmitter in baseband

User Decoder Decoder

Figure 1.3 Digital communication system.

transmitting antenna. Not all systems require all of the mentioned modules; thisdepends on the type of communication used and/or on the required security level.

1.2.1 Transmitter

The multiplexer provides multiple dedicated channels to users and combines thedata (bits) from every channel into one combined stream of data bits. These bits areorganized into frames; a frame has a fixed length of bits. Every user is allocated aspecific position in the frame. There must be a synchronization code or sequence ofbits inside the frame in order to provide the ability to identify each frame and the rel-ative position of the bit inside the frame. There should also be a clock for the correcttransmission of the data.

In this way several signals can share one communication line, instead of havingone line for every signal (Figure 1.4).

Multiplexers (MUX) can range from two input signals up to sixteen or more.For more input signals, a cascade (consisting of simpler multiplexers) is used. Typi-cally they are 2/1, 4/1, 8/1, or 16/1, indicating the number of input signals and onlyone output signal. Figure 1.5 shows a 4/1 multiplexer. This means that four inputsignals share one communication line. The input signals are I0, I1, I2, and I3. Theoutput is Z. Which signal will pass from input to output is decided on the basis ofcontrol signals a1 and a0 as shown in Table 1.1. Input E enables (E = 1) or disables (E= 0) the multiplexer.

1.2 Transmitter Systems 3

DEMUXMUX

Conversation 5

Conversation 4

Conversation 3

Conversation 2

Conversation 1

Conversation 5

Conversation 4

Conversation 3

Conversation 2

Conversation 1

Figure 1.4 Use of one line for several communications.

MUX 4/1I0

I3

I2

I1

a1 a0

Z

E

Figure 1.5 Multiplexer 4/1.

1.2.2 Randomization

Randomization is used to ensure the even number of 0s and 1s in the data informa-tion, which should be randomly distributed. The process of randomization is car-ried out by the exclusive or adding (XOR), which adds a bit from a selected bitsequence to each bit within the multiplexer frame, except for the synchronizationbits. The bit sequence that is used to randomize is called pseudorandom orpseudonoise (PN) sequence. The random distribution of the bit sequence matchesthe Gaussian distribution. This function happens simultaneously with eachmultiplexing frame. PN codes can be generated using a series of shift registers andlogic gates in feedback as shown in Figure 1.6.

There is also a modulo-2 adder (adding without carry). The shift registersreceive a clock signal every Tc seconds. The feedback lines can be used to obtain dif-ferent output codes. For r shift registers, a maximum of 2r − 1 bit sequence can beproduced. This means that with four shift registers, a maximum of fifteen bitsequences can be achieved. After that, the combinations will start repeating them-selves. It is possible to connect the feedback gates to produce a shorter sequence, butshorter sequences are less random and will repeat more often. A circuit configuredto produce the maximum sequence of nonrepeating bits for a given number of shiftregisters is called a maximal length PN code generator. The main characteristic ofthis maximal length code is that the Modulo 2 sum of any sequence with a shiftedversion of itself will produce another shifted version of the same sequence. All com-binations will appear only once except all the 0 combinations, as this state willcause no changes to occur in the shift register values or in the output. The number of1s will always be 1 larger than the number of 0s, independent of the length of thecode.


Table 1.1 CombinationTable for Multiplexer 4/1

E a1 a0 Z

0 x x 0

1 0 0 I0

1 0 1 I1

1 1 0 I2

1 1 1 I3

Clock signal

Modulo 2 adder

PN signal out

1 2 3 4 r

Shift register

Figure 1.6 Pseudonoise generator.

This type of PN generator is used in the transmitter to modulate a continuouswave signal, as well as in the receiver, where an identical PN generator is used todemodulate the received signal.

1.2.3 Encryption

Encryption is used to protect the data should it be intercepted. Usually the bitstreamis changed (encrypted) in such a way that it would be difficult to reconstruct theoriginal bitstream without a decryption device. A problem develops if there is anerror in the received bitstream, which results in an additional error in the decryptionprocess. This is called error extension. Encryption has been used in wars and forinformation protection for a long time now. It can be used in computer systems andcommunication systems for authorization, copyright protection, and other applica-tions. For the encryption process, an encryption key is required. It is usually 40 to256 bits long. The longer the key (cipher strength), the harder it is to break the code.

There are two methods available: the secret and the public key. With the secretkey, both sender and receiver use the same key to encrypt and decrypt the bitstream.This is the fastest method, but there is the problem of getting the secret key to thereceiving side. With the public key, each recipient has a private key that is keptsecret and a public key known to everyone. The sender uses the public key toencrypt the data, whereas the recipient uses the private key to decrypt the data. Inthis manner the private key is never transmitted, and thus is not vulnerable to inter-ception. The most spread encryption standards are the Data Encryption Standard(DES) and the Advanced Encryption Standard (AES).

DES (Figure 1.7) is the most widely used encryption standard, dating from the1970s. It has blocks of 64 bits at a time, and the key length is 56 bits.

The 64 bits of the input block to be enciphered are first subjected to initial per-mutation. The permutated input block becomes the input to a complex key-depend-ent computation. The output of that computation, called the preoutput, is thensubjected to permutation, which is the inverse of the initial permutation. The com-putation which uses the permuted input block as its input to produce the preoutputblock consists of 16 iterations of a calculation depending on the cipher function.

Today DES is considered insecure because a key of 56 bits is not long enough.That is why in 2002, AES was adopted, which is capable of processing data blocksof 128 bits using cipher keys with lengths of 128, 192, and 256 bits. More on AEScan be found in “Announcing the Advanced Encryption Standard (AES),” which isfree to download from the Internet [National Institute of Standards and Technol-ogy (NIST), http://csrc.nist.gov/publications/fips/fips197/fips-197.pdf].

1.2.4 Encoder

Encoders are used in transmitter systems for detection and correction of errors thatmay occur during transmission due to noise or interference. Coding can also be usedfor compressing the information. Most encoders add the redundant (known) bitsexpanding the data (information bits). This slows the traffic. How many redundantbits will be added depends on the surrounding of our communication service (inter-ference) and on the importance of the information being transmitted in real time.


The differential encoder, convolutional encoder, Reed Solomon coding, and Golayencoder are most often used in communication systems.

1.2.4.1 Differential Encoder

Differential encoding of data is required for modulations such as duobinary and dif-ferential phase shift keying. These modulation types are used for optical links andhigh data rates of 10 to 40 Gbps.

The principle of differential encoding is shown in Figure 1.8.


Finalization

Round 16

Round 2

Round 1

Output (64 bits)

Final permutation

Permutation

Cipherfunction

Binary rotation Binary rotation

Cipherfunction

Cipherfunction

Left half (32 bits) Left half (28 bits)Right half (32 bits) Right half (28 bits)

Initial permutation Key permutation

Input (64 bits) Key (64 bits)

Initialization

Subkey #16 (48 bits)

Subkey #2 (48 bits)

Subkey #1 (48 bits)

Permutation


Permutation


Figure 1.7 Data Encryption Standard algorithm.

XOR

1 bitperioddelay

c c d=k k−1

ck−1

dk

Figure 1.8 Differential encoder.

Let dk be a sequence of binary bits that are the input to a differential encoderand let ck be the output of the differential encoder. Then we have

c c dk k k= ⊕−1 (1.1)

where ⊕ is the modulo 2 addition. The direct implementation of the above equationis the use of an exclusive –OR (XOR) gate with a delay in the feedback path of 1 bitperiod delay. At 40 Gbps, 1 bit period is equal to 25 ps.

1.2.4.2 Convolutional Encoder

Information data is susceptible to errors. For useable data, there are methods ofencoding information. This means organizing the 0s and 1s so that errors can becorrected. Convolutional encoding is applied to the data link signal in order to cor-rect bit errors that might occur during transmission, which results in coding gain forthe system. Through the convolutional encoding/decoding process, the majority oftransmission errors will be corrected before they are passed onto the decryptionprocess.

Codes have three primary characteristics: length, dimension, and minimum dis-tance of a code. The code’s length is the amount of bits per code word. The codedimension is the amount of actual information bits contained within each codeword and the minimum distance is the minimum number of information differencesbetween each code word. Convolutional codes are commonly specified by threeparameters: (n, k, m) where n is the number of output bits, k is the number of inputbits and m is the number of memory registers. The quantity k/n is called the coderate R and is a measurement of coding efficiency:

Rkn

= (1.2)

Commonly k and n parameters range from 1 to 8 and the code rate accordinglyfrom 1/8 to 7/8. Memory registers, m, can range from 2 to 10. Another parameter,the constraint length K, is defined by:

( )K k m= ⋅ − 1 (1.3)

which represents the number of bits in the encoder memory that affects the genera-tion of n output bits.

A convolutional encoder can be made with a K-stage shift register and nmodulo-2 adders, where K is called the constraint length of the code. An example ofsuch an encoder with K = 3 and n = 2 is shown in Figure 1.9.

For each bit entering into the register, the output switch samples n = 2 code bitsout (u1 and u2); hence the rate of the code k/n is 1/2. Each output code bit will be afunction of the input bit (located in the leftmost stage of the register) plus two of theearlier bits (in the rightmost stages).

The larger the constraint length K, the greater the number of past bits that havean effect on each output code word.


1.2.4.3 Reed-Solomon Coding

As with convolutional encoding, RS coding adds redundant bits and creates codewords that enable the decoding process to correct errors. RS differs fromconvolutional encoding by performing block encoding (using bytes) rather thanbitwise encoding. Because of the block encoding, RS is eight times faster thanconvolutional encoding. The incoming data stream is first packaged into smallblocks, which are treated as a new set of k symbols to be packaged into asuper-coded block of n symbols, by appending the calculated redundancy. Suchsymbols can either be comprised of one bit or of several bits (symbol code). There-fore, the information transfer rate is reduced by a factor called code rate (R), and themodulator is expanded by the ratio:

1R

nk

= (1.4)

A Reed-Solomon decoder can correct up to t symbols that contain errors in acode word, where

2t n k= − (1.5)

A Reed-Solomon code word is generated using a special polynomial. All validcode words are exactly divisible by the generator polynomial. The general form ofthe generator polynomial is

( ) ( )( ) ( )g x x a x a x ai i t= − − −+ +1 1 2 (1.6)

The code word is constructed using:

( ) ( ) ( )c x g x i x= (1.7)

where g(x) is the generator polynomial, i(x) is the information block, c(x) is a validcode word, and a is referred to as the primitive element of the field.


Input bitm

Secondcode bit

Outputbranch word

Firstcode bit

u2

u1

Figure 1.9 Convolutional encoder: K = 3, rate = 1/2.

1.2.4.4 Golay Code

Marcel J. E. Golay discovered the possible existence of a perfect binary (23, 12, 7)code, with error-correcting capability t = 3, that is, capable of correcting all possiblepatterns of three errors in 23 bit positions, at the most. So the Golay (23, 12, 7) codeis a perfect linear error-correcting code consisting of 212 = 4,096 code words oflength 23 and a minimum distance of 7. Golay also defined the parity check matrixfor this code as:

( )H MI= 11 (1.8)

where I11 is the 11 × 11 identity matrix and M is a 11 × 12 defined matrix. Since thecode’s length is relatively small (length = 23), the number of redundant bits is 11,and the dimension is 12, the Golay (23, 23, 7) code can be encoded by simply usinglook up tables (LUTs). A look up table is an array that holds a set of precomputedresults for a given operation. This array provides access to results faster than com-puting the result of the given operation each time. Beside the perfect binary Golaycode, there is the extended binary Golay code that encodes 12 bits of data in a wordwith a length of 24 bits, so that a triple-bit error can be corrected and a quadru-ple-bit error detected.

1.2.5 Interleaving

Interleaving is used to intermix the bits of the code words generated throughconvolutional encoding. The motivation for interleaving is to compensate for burstor sequential errors, which can otherwise exceed the capability of the decoder tocorrect errors. Each code word generated through convolutional encoding can onlycorrect a limited number of errors that occur in that code word. Sequential errorscan cause multiple errors in a single code word, which can exceed the error-correct-ing capability of the decoding process. Interleaving distributes bits in such a waythat, if sequential errors do occur, they will be distributed over multiple code words.For example, seven errors in a single code word will be distributed during interleav-ing into seven code words each having a single error. While the decoder may not beable to recover data in a code word with seven errors, it can easily recover a singleerror in seven code words.

The disadvantage of interleaving is the delay created by writing a block of bitsinto memory, intermixing the bits, and then pulling the bits from memory. Thisdelay is dependent on the number of bits that are interleaved at a time and the datarate of the aggregate bitstream. Interleaving is performed only on a finite block ofbits at a time. Similar to multiplexing, interleaving requires framing the aggregatebitstream and adding synchronization bits.

Interleavers are divided into periodic and pseudorandom. In periodicinterleavers, symbols of the transmitted sequence are scrambled as a periodic func-tion of time. Periodic interleavers can be either block or convolutional.


1.2.6 Modulation

Modulation is the process of changing one or more parameters of an auxiliary sig-nal, depending on the signal that carries the information. This auxiliary signal iscalled the transmission signal. The signal that carries the information (and controlsthe parameter changes of the transmission signal) is called the modulation signal.The result of the modulation is the modulated signal. The process is performed in adevice called the modulator, which converts the total digital bitstream into the radiofrequency (RF) analog signal. The digital bitstream is usually modulated into theintermediate frequency (IF), which after amplification is upconverted to thetransmit frequency.

There are many analog and digital modulations that are used in communicationsystems. Analog modulations include: amplitude modulation (AM), frequencymodulation (FM), phase modulation (PM), and several others. Digital modulationsinclude frequency shift keying (FSK), phase shift keying (PSK), amplitude shift key-ing (ASK), quadrature amplitude modulation (QAM), pulse code modulation(PCM), and others.

Chapter 6 will discuss more on modulation and demodulation.

1.2.7 Mixer (Upconverter)

Mixers are used in transmitter systems for easier processing of the signal. It is muchcheaper and easier to amplify the signal at a lower intermediate frequency (IF) thanat a higher radio frequency (RF).

The mixer inputs two different frequencies (one of them is a local oscillator fre-quency) and mixes them. The result is the sum and difference of the input signals.The frequency that is not needed must be filtered out. Figure 1.10 shows the mixer,which has a local oscillator frequency added to or subtracted from the inputfrequency.

For upconversion of the frequency, the local oscillator frequency fLO is added tothe input signal frequency fin:

f f fout in= + LO (1.9)

True systems mixers will produce more than just the sum and difference of theinput signals. There will be intermodulation products from the input signals.

If a second signal fin2 arrives at the input with the fin, the mixer will generateintermodulation products at its output due to inherent nonlinearity, in the form


Outputsignal

Inputsignal

Mixer

Localoscillator

Figure 1.10 The mixer.

± ⋅ ± ⋅m f n fiin in2 (1.10)

where m and n are positive integers, which can assume any value from 1 to infinity.The order of the intermodulation is defined as m + n. Accordingly, 2 fin − fin2, 2 fin2 −fin, 3 · fin and 3 · fin2 are third order products by definition. The first two products arecalled two-tone third-order products as they are generated when two tones areapplied simultaneously at the input. Two-tone third-order products are very closeto the desired signals and are very difficult to filter out.

1.2.8 Filter

Filtering of the frequency range is an important part of every communication sub-system—hence the transmitter. Filtering is the ability to select the frequency rangewe wish to process and to block all other frequencies. Filters can be analog or digi-tal. Figure 1.11 shows the symbols used for lowpass, bandpass, and highpass filters,depending on which frequency range needs to be processed further.

On lower frequencies, LC filters are used, and on higher frequencies (such asmicrowave) the microstrip is used.

Filters will be discussed in more detail in Chapter 5.

1.3 Receiver Systems

The receiver system largely depends on the transmitter system. If in a communica-tion system a multiplexer is used on the transmitter’s side, there must be ademultiplexer on the receiving side. The same applies for other blocks mentioned inthe previous section, which on the receiving side are placed in reverse order. Somereceivers must deal with very small signals. Better and more expensive receivers willintroduce very little noise themselves. Again, depending on the complexity of thecommunication system, the following blocks are optional: filter, downconverter,demodulator, deinterleaver, decoder, derandomizer, and demultiplexer.

1.3.1 Filter

The filter is part of every receiving system. It selects the frequency band of use to beprocessed further and it stops signals on all other frequencies. The selectivity of thefilter is shown by Q factor which can be calculated as

Qf

f fc=−2 1

(1.11)

1.3 Receiver Systems 11

Figure 1.11 Lowpass, bandpass, and highpass filters.

where f2 and f1 are frequencies where the power drops by 50% (3 dB), and fc is thecentral or resonant frequency as shown in Figure 1.12.

1.3.2 Mixer (Downconverter)

On the receiving end of the communication system there is also a mixer, which inthis case serves as a downconverter. It is necessary to downconvert the received RFfrequency because it is much easier to amplify the signal at intermediate frequencies(IF) than at RF frequencies. Here, again, the local oscillator is necessary, and theoutput frequency is obtained as

f f fout in= − LO (1.12)

where the input frequency fin must be greater than fLO; otherwise an error will occur.The other result, that is, the adding of the two frequencies, will be filtered out.Again, intermodulation products may occur here. That is why it is necessary to takeinto consideration all possible transmitters in the vicinity (depending on the applica-tion, this can be up to 50 km) and calculate the intermodulation in order to deter-mine whether additional filtering is required.

1.3.3 Demodulator

The demodulator converts an analog RF signal into a digital bitstream. It extractsthe original information from the modulated carrier wave. There are different typesof demodulation, such as envelope detection, differential, coherent, and synchro-nous demodulation.

Demodulation and demodulators will be discussed in greater detail in Chapter 6.

1.3.4 Deinterleaver

If a bitstream was interleaved in the transmission process, deinterleaving is requiredin the receiving process to reassemble the code words created by the encoder.Should any errors have occurred before deinterleaving, they will be distributeddepending on the selected algorithm. Figure 1.13 shows the deinterleaving of anarray of three element structures. Synchronization bits must be present to recognizewhen one frame is finished and the other is starting.


Bandwidth −3dB

fcf1 f2

Figure 1.12 Selectivity of the filter.

1.3.5 Decoder

The decoder in the receiver system must match the encoder that was used in thetransmitter system.

1.3.5.1 Differential Decoder

If differential encoding was used in the transmitter system, differential decodingmust be performed in the receiving system. The differential encoding process doesnot introduce redundant bits, but transforms the waveform by converting the spacesignal (zeros) into transitions. Accordingly, the decoding process converts transi-tions back to spaces. Since a single bit error affects two transitions, the differentialdecoding process doubles any bit error, corresponding to a 3-dB loss to the system.

The decoder decodes the binary input signal. The output is the logical differencebetween present and previous input. The input and the output are related with

( ) ( )( )

m t d t

m t dk

0 0=

=

XOR initial condition parameter value

( ) ( )t d tk kXOR −1

(1.13)

where d is the differentially encoded input, m is the output message, tk is the kth timestep, and XOR is the logical exclusive-or operator.


X[3] C

X[2] C

X[1] C

X[0] C

X[3] B

X[2] B

X[1] B

X[0] B

X[3] A

X[2] A

X[1] A

X[0] A

A 3

B 3

C3

A 2

B 2

C2

A 1

B 1

C1

A 0

B 0

C0 Y2

Y1

Y0

Figure 1.13 Deinterleaving an array of three element structures.

1.3.5.2 Viterbi Decoder

The Viterbi decoder is used together with convolutional coding, and is applied tothe data link signal to correct errors that may have occurred during the RF transmis-sion. The encoding/decoding process adds what is referred to as coding gain, whichmay be necessary for the successful data link transmission. The decoding correctserrors before the decryption process of total bitstream. Correction of the errorsoccurs because the convolutional encoder (or transmit side of the data link) createscode words, which contain data bits with added redundant bits. The redundant bitsallow the decoder to detect and correct errors that may exist in each code word. Theprocess of decoding is much more complicated than the encoding process, and limitsthe speed of the bitstream that needs to be decoded.

If the convolutional code uses 2n possible symbols, the input vector length is K ·n for positive integer K. If decoded data uses 2k possible output symbols, the outputlength will be K · n. The integer number K is the number of frames processed in eachstep. The entry into the decoder input can be a real number (positive real is logicalzero, while negative real is logical one), 0 and 1 (0 is logical zero, 1 is logical 1). Thelatter is called a hard decision. The third possible input is a soft decision. It can beany integer between 0 and 2b − 1, where b is the number of the soft decision bitparameter. Here 0 is the most confident decision for logical zero, 2b − 1 is the mostconfident decision for logical one, and other values are less confident decisions.Table 1.2 shows the decisions for three bits.

1.3.5.3 Reed-Solomon Decoding

The Reed-Solomon encoder and decoder are commonly used in data transmissionand storage applications, such as: broadcast equipment, wireless LANs, cablemodems, xDSL, satellite communications, microwave networks, and digital TV.The block diagram of the RS decoder is shown in Figure 1.14.

The received code word r(x) is the original (transmitted) code word c(x) plusadditional errors:

( ) ( ) ( )r x c x e x= + (1.14)


Table 1.2 3-Bit Soft Decision

InputValue Decision

0 Most confident zero

1 Second most confident zero

2 Third most confident zero

3 Least confident zero

4 Least confident one

5 Third most confident one

6 Second most confident one

7 Most confident one

The decoder will try to identify the position and the magnitude of maximum terrors (or 2t erasures) and correct the errors and erasures. A Reed-Solomon codeword has 2t syndromes (Si) that depend only on errors and not on the transmittedcode word. The syndrome is calculated by substituting the 2t roots of the generatorpolynomial g(x) into r(x). To find the symbol error location, solving simultaneousequations with t unknowns is necessary. First, the error locator polynomial (L(x)) isfound using the Berklekamp-Massey or Euclid’s algorithms, with v being the num-ber of errors. The roots of the polynomial [i.e., the error locations (Xi)], are foundwith the Chien search algorithm. Next, the symbol error values (Yi) are found usingthe Fornay algorithm.

1.3.5.4 Golay Decoder

The Golay coding can detect up to four bit errors in 24 bits (12 information bits)and correct up to three bit errors in 24 bits. If in 24 received bits there are three orless errors, the Golay decoding algorithm will detect the errors and correct them. Iffour errors appear, they will be detected but the exact pattern will not be deter-mined. An error message will be displayed. If there are more than four errors, theGolay decoding will not provide the actual error pattern, and the information in the12 bits will be lost.

1.3.6 Decryptor

Decryption is the process that reconstructs the original signal, which was alteredthrough the encrypter in the transmitter. Decryption is required in the receiver sys-tem only if the encrypter was used in the transmitting system. Encryption is used asa protection means from signal interception. Error extension is possible duringdecryption (where multiple errors will be added for every error bit received).

For decryption of the Data Encryption Standard (DES), the same encryp-tion algorithm (Figure 1.7) is used, with the same key, but reversed key schedule(16, ..., 1).

1.3.7 Derandomizer

When randomization is used in the transmitting system, a derandomization of thedata bitstream must be done in the receiving system. If synchronization bits are notrandomized, they do not need to be derandomized. Synchronization bits identify themultiplexing frame, which is derandomized by Modulo 2 adding the same PNsequence that was used to randomize the frame to the bits within the frame.


Input

r x( ) SiSyndromecalculator

Errorpolynomial

Errorlocations

Errorcorrector

Yi

Xi c x( )L x( )

v

Errormagnitudes

Output

Figure 1.14 Reed-Solomon decoder.

1.3.8 Demultiplexer

The demultiplexer (Figure 1.15) is a device that receives data from one input anddistributes it on 2n possible outputs, where n is the number of control bits.

Table 1.3 shows the combination table for the demultiplexer 4/1. The data ccoming to input “Z” will be distributed to four outputs I0, I1, I2, and I3 according tothe controlling combinations of a1 and a0. All other outputs will have 0 as the outputvalue.

The demultiplexer recreates the user channels from the total bitstream. Thebitstream is organized into multiplexing frames with a fixed bit length. Every userchannel is allocated a specific bit position inside the frame. Inside the frame thereare synchronization bits, which are used in the demultiplexing process, in which thebits are distributed to the appropriate user channel.

This is not all that has to be thought of when considering the receiver system.Received power, sensitivity, required ratio of signal to noise, and noise factor arejust some of the important parameters that have to be taken into considerationwhen planning a communication link.

1.3.9 Received Power

Received power (Pr) at the receiving point is calculated using the effective area of anantenna (λ2/4π) and power density (Pt /4 · π · d2) as

PP

dP

drt

t=⋅

⋅⋅ ⋅

=⋅ ⋅

⎛⎝⎜

⎞⎠⎟

λ

π π

λ

π

2

2

2

4 4 4(1.15)

where Pr is the received power, d is the distance from the transmitter to the receiver,Pt is the transmitted power, and λ is the wavelength of the signal. There are transmit-ting and receiving antenna gains Gt and Gr, for the antenna so the previous expres-sion can be written as

P P G Gdr t r t= ⋅ ⋅

⋅ ⋅⎛⎝⎜

⎞⎠⎟

λ

π4

2

(1.16)


DEMUX 1/4

I0

I3

I2

I1

Z

a1 a0

Figure 1.15 Demultiplexer 1/4.

1.3.9.1 Receiver Sensitivity

Receiver sensitivity is the minimum level of signal at the input of the receiver, whichis required to achieve a sufficient level of signal-to-noise ratio for the demodulation.The sensitivity is determined with thermal noise Pterm, required ratio of signal tonoise (S/N)req for demodulation, and noise factor (NF) as

P PSN

NFr termreq

min = ⋅ ⎛⎝⎜

⎞⎠⎟

⋅ (1.17)

Receivers have the lowest level of signal strength required to process the infor-mation without loss of data. With digital systems, a lower received signal strengthwill result in a lower rate of received information. Typically receivers have a sensi-tivity ranging from −60 to −94 dBm (10−9 to 4 × 10−13 W).

1.3.9.2 Thermal Noise

The thermal noise of the receiver is defined as

P k T Bterm = ⋅ ⋅ (1.18)

where k is the Boltzmann constant 1,38⋅10−23 J/K, T is the temperature in Kelvin(290–300K), and B is the frequency range width in hertz. The density of the thermalnoise at room temperature (290K) is 204 dBw/Hz. The width of the frequency chan-nel B is determined by the receiving filter width.

Required ratio signal to noise in the receiver is the ratio of signal to noiserequired for a certain quality of the link (i.e., relative number of bits or frames witherrors). The ratio of signal to noise is the difference between received signal andnoise:

[ ] ( ) ( )S N S NdB = −10 10log log (1.19)

For analog systems, the S/N ratio must always be above zero. In digital systems(spread spectrum), the signal can be buried in the noise. The higher the bit rate, thelarger the signal to noise ratio must be.

1.3.9.3 Noise Factor

The noise factor of the receiver (NF) is the ratio of signal to noise at the input andoutput of the receiver:


Table 1.3 Combination Tablefor Demultiplexer 1/4

a1 a0 Z I0 I1 I2 I3

0 0 c 0 0 0 c

0 1 c 0 0 c 0

1 0 c 0 c 0 0

1 1 c c 0 0 0

NFS N

S Nin

out

= (1.20)

This ratio can be from a fraction of a decibel for low noise microwave convert-ers [0.3 dB for low noise block downconverters (LNB) for satellite applications] to30 or 40 dB for spectral analyzers; typically the ratio ranges from 2 to 10 dB. This isactually the noise the receiver itself introduces into the system.

The noise threshold is the sum of the thermal noise and the noise factor.

1.4 User Interface

The user interface is the means for people to interact with the communication sys-tem. It consists of input of some sort and output. The input can be a command usinga keyboard, voice, or text. We have heard of the phrase “user-friendly,” whichmeans that it is simple to operate a certain device. When designing an application, alot of care is taken to make a suitable user interface. Nowadays, there are hearingaids and other tools available for individuals with a handicap. Normally, the userinterface is graphical (GUI), but it can also be operating via voice or touch.

1.4.1 Graphical User Interface (GUI)

The graphical user interface interacts with electronic devices through icons or visualindicators. Touch screens are one of the GUI types. There are also the command lineand text user interfaces, which use a keyboard to type the commands. Maintouchscreen technologies are resistive and capacitive. Resistive LCD touchscreenmonitors rely on a touch overlay, which is composed of a flexible top layer and arigid bottom layer separated by insulating dots attached to a touchscreen controller.The inside surface of each of the two layers is coated with a transparent metal oxidecoating that facilitates a gradient across each layer when voltage is applied. Pressingthe flexible top sheet creates electrical contact between the resistive layers, andcloses a switch in the circuit. The control electronics alternate voltage between thelayers and pass the resulting X and Y touch coordinates to the touchscreen control-ler. The touchscreen controller data is then passed on to the computer operating sys-tem for processing. Capacitive touch screens work by placing a very small charge ateach of the four corners of the screen. When a finger touches the screen, the touchcontroller determines the change of capacitance of the screen from each of the fourpoints and provides a touch value at the correct location. Surface acoustic wavetouch screen technology is based on sending acoustic waves across a clear glasspanel with a series of transducers and reflectors. When a finger touches the screen,the waves are absorbed, causing a touch event to be detected at that point.

Because the panel is all glass, there are no layers that can be worn, which resultsin durability. Infrared technology is based on the interruption of an infrared lightgrid in front of the display screen. The touch frame contains a row of infrared LEDsand photo transistors, each mounted on two opposite sides to create a grid of invisi-ble infrared light.


1.4.2 Voice User Interface (VOI)

The voice user interface can be activated through speech. Today hands-free com-mands are possible. The possibility of error when inputing a command or data byvoice is higher than when entering it through a keyboard. Figure 1.16 shows a possi-ble user interface for a communication device.

1.5 Antenna Systems

Antenna systems consist of a duplexer and an antenna used to transmit and receiveinformation from one user to another. There are many types of antennas that can beused for a communication system, depending on the frequency of use, power, appli-cation, and even international standard regulations. Most communication systemsuse the same antenna for transmitting and receiving a signal. This normally requirestwo antennas, which have to be physically separated. This is impractical, except insome cases of high interference when antenna diversity could be an option. That iswhy in most cases a single antenna is used for both transmitting and receiving thesignal. This is possible with the use of a duplexer.

1.5.1 Duplexer

The duplexer makes it possible for receiver and transmitter systems to use the sameantenna; otherwise it would be necessary to use two antennas. The duplexer has fil-ters, which isolate the transmitting frequency from the receiving frequency. Sincethe transmitting and receiving frequency are usually not the same (because of inter-ference), there must be a separation between them. The duplexer must be designedto operate in the frequency band used by both the receiver and the transmitter. Italso must be able to operate on the power from the power amplifier. When workingat the transmitting frequency, it must reject the noise from the receiver and viceversa.

The duplexer can be made with a hybrid ring, cavity notch, and a band-pass/band-reject design. Figure 1.17 shows the design of a reject duplexer using notchcavities.

1.5 Antenna Systems 19

Keyboard

Screen

Audioprocessor

Signalprocessor Transmitter

Userinterface

Communicationinterface

Figure 1.16 User interface.

Using only two notch cavities would probably not provide sufficient isolationfor most situations. The cavity in the transmitter is tuned to the receiver frequency,and the cavity in the receiver is tuned to the transmitter frequency. That means thatthe cavity in the transmitter area will pass the transmitting frequency and notch(reject) the receiving frequency. The same applies for the cavity in the receiver area,which will pass the receiver frequency and notch the transmitter frequency. If thereare other strong signals present, this design will not be enough. A more advanceddesign must include four to six cavities. Two or three cavities in each leg are moreeffective than just one. Usually the cavities require tuning with a spectrum analyzeror wattmeter.

1.5.2 Antenna

The antenna is a device that transforms a guided electromagnetic wave from thetransmission line (waveguide or cable) into a space wave in free space. The antennaactually makes a transition between the guided wave in the transmission line andthe space wave in free space. The most important characteristics of an antenna are:radiation pattern, directivity, impedance, gain, and affective area.

1.5.2.1 Radiation Pattern

Electric field intensity falls with 1/d, where d is the distance from the antenna. Tomeasure the electric or magnetic field from the antenna, we have to be far enoughfrom the antenna (only the radiating field exists). This happens at the distance d,

dD= 2 2

λ(1.21)

where D is the largest dimension of the antenna and λ is the wavelength of the sig-nal. Then, knowing the electric field E, the magnetic field H can be calculated from

HE=η

(1.22)


To transmitter To receiver

Cavity tuned totransmitter frequency

To antenna

Optimal cable length

Cavity tuned toreceiver frequency

Figure 1.17 Reject duplexer with notch cavities.

where η is the impedance of free space, that is 120π or 377Ω.Usually the radiation pattern is given in two perpendicular planes: horizontal

and vertical; it usually has one main lobe and several sidelobes.

1.5.2.2 Directivity

Often the goal of an antenna is to have most of the radiation in just one directionwith much less radiation in other directions. The directivity angle is calculated asthe angle where the power density is one half of the maximum and the field densitydrops for a factor of 1 2/ . Directivity is defined as the ratio of the power densityradiated by the antenna in the direction of maximum intensity and the power den-sity radiated by the isotropic radiator. The isotropic radiator radiates equally in alldirections. Antennas with higher directivity are used to radiate as much energy tothe receiver as possible. At the same time, dispersion of the signal in unwanteddirections is diminished, thus making the interference to other systems smaller.

1.5.2.3 Antenna Impedance

Antenna impedance is the ratio of the voltage and current at the antenna. The mostpower from the generator will be given to the antenna if the antenna and generatorimpedances are complex conjugates (i.e., Z Za G= * ). That means that theirresistances must be equal, whereas their reluctances must be equal in magnitude butof opposite signs. Usually generators have an output impedance of 50Ω or 75Ω, sothe antenna will have to be of the same impedance if possible.

1.5.2.4 Gain

Gain is related to the power received from the generator and represents the numbershowing how much larger the power from the isotropic radiator must be comparedto the received power of the antenna, in order for the radiation from the isotropicradiator to be the same as the radiation from the observed antenna in the directionof maximum radiation. For an ideal antenna without losses, the gain would beequal to directivity. Gain of the antenna is usually given in decibels.

1.5.2.5 Effective Area

The effective area of a receiving antenna, Aeff, is defined as the ratio of receivedpower, Pr, absorbed on a matched load connected to the antenna, and power den-sity of the incident electromagnetic wave, Sr:

AP

Seffr

r

= (1.23)

The power density of the transmitting antenna in the maximum direction isequal to

1.5 Antenna Systems 21

SG P

dr

t t=⋅ ⋅4 2π

(1.24)

where Gt and Pt are gain of the transmitting antenna and power of the transmittingantenna.

The relation that connects the effective area and gain for all antennas is

A Geff =⋅

⋅λ

π

2

4(1.25)

1.5.2.6 Antenna Types

There are many types of antennas, depending on the type of the application needed.They can be divided into four groups: electrically small antennas, wideband anten-nas, resonant antennas, and aperture antennas.

Electrically small antennas are much smaller in dimension than the wavelengthassociated with the frequency on which they operate. They have small directivityand radiation effectiveness. They include Hertz’s dipole and monopole. To increasedirectivity, antenna arrays can be built. By changing the phase of the supplying cur-rents, different radiation patterns can be obtained.

Wideband antennas have a stable radiation pattern, gain, and impedance in thewide frequency range. The gain is small to medium. The biconical antenna andlog-periodic antenna are examples of this type of antenna. Resonant antennas oper-ate in one or more selective frequency ranges. They have a small to medium gain.The microwave microstrip antenna is a resonant antenna. Aperture antennasreceive and radiate electromagnetic waves through an aperture. They have largegain, which increases with the frequency. The horn antenna and parabolic dish areexamples of this type of antenna.

1.5.2.7 Smart Antenna Systems

A smart antenna system uses multiple antenna elements including signal processingto optimize its radiation pattern depending on the signal environment. The smartantenna interference is smaller, which enables reuse of the frequency more often.This can also improve the capacity of the link. Greater signal gain will result inlower power requirements at the receiving system with a smaller size and battery.The power amplifier used can be cheaper, with less total power consumption.

1.6 Power Supplies

Power in communication systems is necessary for the operation of electronic com-ponents. For simple systems, a DC supply is sufficient. For the high power of atransmitter, a power amplifier is necessary. The transmitter system usually requiresmore power than the receiving system. The majority of power is used to amplify thesignal before reaching the antenna. In calculating the communication link, a freespace loss must also be taken into consideration. In addition, cable loss and match-


ing losses require the transmitted power to be raised. For small transmitting power,a simple 12-V DC supply is sufficient. Larger power requires power amplifiers.

1.6.1 Power Supply Types

Linear power uses a transformer to convert the voltage from the mains to a lowervoltage. Converters, which transform 120-V or 220-V AC into a lower DC voltage(typically 12V or 24V), are often used for electronic circuits. There are many typesavailable. An uninterruptible power supply (UPS) must be used in applications forwhich a constant power supply is necessary. UPS usually takes the power from theAC mains and charges its own battery at the same time. If there is a loss of power,the battery will provide the necessary power for some time. There are solutionswhere the UPS charges a battery with energy generated from internal combustionengines or turbines. Batteries are also often used for mobile communications. Insome situations solar power might be used, especially in areas with a lot of sun.

1.6.2 Power Amplifier

Power amplifiers are used to increase the level of the signal, both in transmitter andreceiver systems.

They are used to amplify the low-level signal to a higher value. Power gain isdescribed as

( )GP

Pout

in

dB =⎛⎝⎜

⎞⎠⎟10 10log (1.26)

where Pin is input power and Pout output power.Power amplifiers can be divided in classes A, B, AB, C, D, and E. Class A uses

100% of the input signal. This amplifier is inefficient and is used for small signals orlow power amplification. Class B uses 50% of the input signal. It is more efficientthan class A, but subject to signal distortions. Class AB is a combination of class Aand class B. It uses more than 50% of the signal. Class C uses less than 50% of thesignal. Distortions are high, but so is the efficiency. Class D uses switching (on/off)for high efficiency. It can be used in digital circuits. There are also some otherspecial classes.

1.7 Considerations for Voice Versus Data

Input information into the communication system can be voice or data (text, pic-tures, video, and so forth). In this section just some of the codecs are mentioned.There are many more; some of them are obsolete, while others are being developed.

1.7.1 Text

ASCII uses 7 bits per character. Extended ASCII uses 8 bits per character.

1.7 Considerations for Voice Versus Data 23

1.7.2 Images

The Graphics Interchange Format (GIF) (lossless compression) uses 8 bits per pixeland a 256 color palette. The Joint Photographic Exchange Group (JPEG) (lossycompression) format most often uses a 10:1 compression.

1.7.3 Voice

Pulse code modulation (PCM) has 8,000 samples per second—with 8 bits per sec-ond it results in 64 Kbits per second. Compression techniques are adaptive differen-tial pulse code modulation (ADPCM) (32 Kbps) and residual excited linearpredictive coding (8–16 Kbps). Audio music requires 32–384 Kb/s.

The audio signal is sensitive to delay and jitter. Latency is the end-to-end delayfrom mouth to ear. It must not exceed 100 ms for excellent quality. For acceptablequality it should not exceed 150 ms. For higher delays an echo canceller is required.

There is a propagation delay in free space, which depends on the frequency usedand the distance between the transmitter and receiver. Packetization delay is thetime required to create an audio packet and send it on a network—it depends on thecodec. Table 1.4 gives the data rates for some audio codecs. The G.711 codec usedin telephony works at 64 Kbps.

1.7.4 Video

H.261 coding uses 176 by 144 or 352 by 258 frames at 10–30 frame/sec. MPEG-2and HDTV use 1,920 by 1,080 frames at 30 frames/sec.

Selected Bibliography

Balanis, C. A., Antenna Theory—Analysis and Design, New York: John Wiley & Sons, 2005.Brown, S., and Z. Vranesic, Fundamentals of Digital Logic with VHDL Design, New York:McGraw-Hill, 2001.Couch, L. W., Digital and Analog Communication Systems, 6th ed., Upper Saddle River, NJ:Prentice-Hall, 2001.Dunlop, J., and D. G. Smith, Telecommunication Engineering, London, U.K.: Chapman and Hill,1994.Diffie, W., and M. E. Hellman, “Privacy and Authentification: An Introduction to Cryptogra-phy,” Proceedings of the IEEE, Vol. 67, No. 3, March 1979, pp. 397–428.Hanna, S. A., “Convolutional Interleaving for Digital Radio Communications,” Proc. 2nd Inter-national Conference on Personal Communications: Gateway to the 21st Century, 1993, Vol. 1,pp. 443–447.


Table 1.4 Data Rates for Audio Codecs

ADPCM G.711 G.729a

Sample Rate 8 KHz 8 KHz 8 KHz

Effective Sample Size 8 bits 4 bits 1 bit

Data Rate 64 Kbps 32 Kbps 8 Kbps

Federal Information Processing Standards Publications 197: “Announcing the Advanced Encryp-tion Standard (AES),” http://csrc.nist.gov/publications/fips/fips197/fips-197.pdf.Gardiol, F. E., Introduction to Microwaves, Dedham, MA: Artech House, 1984.Morelos-Zaragoza, R. H., The Art of Error Correcting Coding, New York: John Wiley & Sons,2006.Sklar, B., Digital Communication: Fundamentals and Applications, Upper Saddle River, NJ:Prentice-Hall, 2001.Xiong, F., Digital Modulation Techniques, Norwood, MA: Artech House, 2000.

1.7 Considerations for Voice Versus Data 25

C H A P T E R 2

Electromagnetic Spectrum Used forCommunications

2.1 Electromagnetic Spectrum

The electromagnetic (EM) spectrum of an object is the distribution of electromag-netic radiation from that object. The EM spectrum (Figure 2.1) covers frequenciesfrom 3 Hz (ELF) to gamma rays (30 ZHz) and beyond (cosmic rays).

The corresponding wavelengths λ can range from thousands of kilometers to afraction of an atom size (Table 2.1). The frequency and the wavelength are relatedby the following expression:

λ = cf

(2.1)

where c is the speed of light—approximately 30,000,000 m/s.The energy of the particular range is defined as

E h f= ⋅ (2.2)

where f is the frequency in hertz and h is the Planck’s constant, 6.62606896e−34 Js.Energy can be expressed in eV, where 1 eV is approximately 1.60217653e−19 J. OneeV is equal to the amount of energy gained by a single unbound electron when itaccelerates through an electrostatic potential difference of 1 volt. It is also theenergy needed to break the chemical bond in the cell. The higher the frequency, thehigher the energy in each photon (Table 2.1).

Table 2.2 gives the prefix converters used in Table 2.1.The spectrum is divided in decades. The radio spectrum (including microwaves)

is considered to cover frequencies from 9 kHz to 300 GHz, that is, from VLF toSHF. Most communications take place in the radio spectrum, but the infrared andthe visible spectrum can be used as well. The use of frequency bands for communi-cation is discussed latter in Sections 2.1.1 to 2.1.13.

27

2.1.1 Extra Low Frequency (ELF)

Most sources in the ELF band are natural or accidental. However, ELF can be usedfor submarine communications, since signals with a transmitter power of 100 MWcan penetrate up to several hundred meters deep. However, the messages are veryshort.

2.1.2 Super Low Frequency (SLF)

The SLF band, like ELF, can be used for submarine communications. Unwantedsources can occur from power lines (50 or 60 Hz), which run for kilometers. Thissignal is called hum. Another natural source example is the interaction of solar windwith the ionosphere.

28 Electromagnetic Spectrum Used for Communications

ELF VLF LF MF HF VHF UHF SHF EHF IR

Visi

ble

UV X-ray

SOFT HARD

Gamma-ray

SOFT HARD

ULFSLF

Figure 2.1 Electromagnetic spectrum.

Table 2.1 Electromagnetic Spectrum

Range Frequency Wavelength Energy (eV)

Extremely low frequency (ELF) 3 Hz–30 Hz 108m–107m 1.24 · 10−14–1.24 · 10−13

Super low frequency (SLF) 30 Hz–300 Hz 107m–106m 1.24 · 10−13–1.24 · 10−12

Ultra low frequency (ULF) 300 Hz–3 kHz 106m–105m 1.24 · 10−12–1.24 · 10−11

Very low frequency (VLF) 3 kHz–30 kHz 105m–104m 1.24 · 10−11–1.24 · 10−10

Low frequency (LF) 30 kHz–300 kHz 104m–103m 1.24 · 10−10–1.24 · 10−9

Medium frequency (MF) 300 kHz–3 MHz 103m–102m 1.24 · 10−9–1.24 · 10−8

High frequency (HF) 3 MHz–30 MHz 102m–101m 1.24 · 10−8–1.24 · 10−7

Very high frequency (VHF) 30 MHz–300 MHz 101m–1m 1.24 · 10−7–1.24 · 10−6

Ultra high frequency (UHF) 300 MHz–3 GHz 1m–10−1m 1.24 · 10−6–1.24 · 10−5

Super high frequency (SHF) 3 GHz–30 GHz 10−1m–10−2m 1.24 · 10−5–1.24 · 10−4

Extremely high frequency (EHF) 30 GHz–300 GHz 10−2m–10−3m 1.24 · 10−4–1.24 · 10−3

Infrared (IR) 0.3 THz–400 THz 10−3m–750 · 10−9m 1.24 · 10−3–1.65

Visible 400–790 THz 750 · 10−9m–380 · 10−9m 1.65–3.27

Ultraviolet (UV) 750 THz–30 PHz 400 · 10−9m–10 · 10−9m 3.10–124

X-ray 30 PHz–30 EHz 10 · 10−9m–0.01 · 10−9m 124–124,000

Gamma ray 30 EHz–30 ZHz 0.01 · 10−9m–10 · 10−15m 0.124–124 · 103

Table 2.2 Prefix Converters

Symbol Z E P T G M k m µ n p f

Prefix zetta exa peta tera giga mega kilo milli micro nano pico femto

Factor 1021 1018 1015 1012 109 106 103 10−3 10−6 10−9 10−12 10−15

2.1.3 Ultra Low Frequencies (ULF)

Along with ELF and SLF, the ULF band can also be used for submarine communica-tions. ULF is used in mines as well. Only slow modulation can be applied (Morsecode), limiting the amount of information. If only the phase is required, as is thecase with navigation systems, a limited amount of information is not adisadvantage.

2.1.4 Very Low Frequency (VLF)

Similar to ULF, VLF is used for navigation systems and communication over largedistances. The information capacity is small with VLF. Communication with sub-marines is possible only near the surface. Lightning also happens in this band. Fre-quencies below 9 kHz are not allocated by the International TelecommunicationUnion and can be used freely for communications in some countries.

2.1.5 Low Frequency (LF)

Communication in this band is possible around the Earth by refraction from theionosphere and reflection from the Earth’s surface. It can be used for navigation,AM radio, and radio frequency identification (RFID).

2.1.6 Medium Frequency (MF)

Like the LF band, MF can use refraction from the ionosphere—but only at night. Itis used for AM radio, amateur radio, and navigation.

2.1.7 High Frequency (HF)

The HF band is also known as the short wave band. It is used for medium- andlong-range communications, such as marine and aviation communications, ama-teur radio, and RFID. More information can be sent in channels in the HF bandsthan in the previously described bands.

2.1.8 Very High Frequency (VHF)

The VHF band is used for radio (FM) and television at short distances (little morethan line of sight (LOS). VHF antennas are usually one quarter or one half wave-length long. VHF can also be used for land mobile communications, radio astron-omy, cordless telephones, amateur radio, navigation, satellite communications, andrailways.

2.1.9 Ultra High Frequency (UHF)

UHF is used for television, mobile phones, satellites, radar, RFID, the global posi-tioning system (GPS), Bluetooth, WLAN, and so forth. The communication ispoint-to-point over line-of-sight (LOS). For larger distances, a repeater is necessary.

2.1 Electromagnetic Spectrum 29

UHF is strongly affected by rain. The antenna size in this frequency range is about awavelength.

2.1.10 Super High Frequency (SHF)

SHF is used for satellite communications, microwave links, and radar. It is used forline-of-sight communications.

2.1.11 Extra High Frequency (EHF)

The EHF band is mostly used for satellite communications, but not yet for othertypes of communications as it is hard to modulate and demodulate high frequencieson the band.

2.1.12 Infrared (IR)

IR is used for short-range wireless communications and in astronomy. Computers,PDAs, and remote controls use Infrared Data Association (IrDA) technology.Devices must be in line-of-sight (LOS) and the data transmitted must be short.

2.1.13 Visible

Optical fiber is suitable for large distances because light propagates with little atten-uation. It is used for a large amount of data traffic. Visible light communications(VLC) is a new technology that uses light that is visible to human eyes. It must beline-of-sight and suffers from interference from other light sources.

2.2 Spectrum Division

The International Telecommunication Union (ITU) is the leading United Nations’agency for information and communication technologies. It has three sectors: radiocommunications, standardization, and development. ITU manages internationalradio frequencies, allocating the spectrum and frequencies in order to avoid inter-ference between radio stations of different countries. In recent years, radio commu-nication systems have expanded largely. The radio frequency spectrum is a naturalresource, and its allocation has to be planned well ahead.

Apart from the traditional division shown in the previous section, there are anumber of other divisions, such as radar, satellite, and military frequency band des-ignations, given in Tables 2.3–2.8.

The bands for TV receive only (TVRO) are given in Table 2.4. TVRO is a satel-lite technology for receiving satellite TV programs from fixed service satellites.

Military secret radar bands originate from World War II and were used forradars. After the war, the secrecy was lifted. IEEE adopted the codes, and today theyare in use in radar, satellite, countermeasures, and terrestrial communications.

ITU bands are subbands of military designations.


2.2 Spectrum Division 31

Table 2.5 Military ElectronicCountermeasures BandDesignations (NATO)

Frequency Band

30–250 MHz A

250–500 MHz B

500–1,000 MHz C

1–2 GHz D

2–3 GHz E

3–4 GHz F

4–6 GHz G

6–8 GHz H

8–10 GHz I

10–20 GHz J

20–40 GHz K

40–60 GHz L

60–100 GHz M

Table 2.4 Satellite TVROBand Designations

Frequency Band

1.7–3 GHz S

3.7–4.2 GHz C

10.9–11.75 GHz Ku1

11.75–12.5 GHz Ku2(DBS)

12.5–12.75 GHz Ku3

18.0–20.0 GHz Ka

Table 2.3 IEEE Radar Band Designations Bands(According to IEEE Standard 521-2002)

Frequency Wavelength Band

3–30 MHz 100–10m HF

30–300 MHz 10–1m VHF

300–1000 MHz 100–30 cm UHF

1–2 GHz 30–15 cm L

2–4 GHz 15–7.5 cm S

4–8 GHz 7.5–3.75 cm C

8–12 GHz 3.75–2.50 cm X

12–18 GHz 2.5–1.67 cm Ku

18–27 GHz 1.67–1.11 cm K

27–40 GHz 11.1–7.5 mm Ka

40–75 GHz 7.5 mm–4 mm V

75–110 GHz 4 mm–2.73 mm W

110–300 GHz 2.73 mm–1 mm mm

300–3,000 GHz 1 mm–100 µm µm


Table 2.6 Traffic Radar Designations(Police)

Frequency Band

2.455 GHz S

10.525 GHz ± 25 MHz X

13.450 GHz Ku

24.125 GHz ± 100 MHz K

24.150 GHz ± 100 MHz K

33.4–36.0 GHz Ka

332 THz IR (Infrared)

Table 2.7 Military Radar

Frequency Band

3–30 MHz HF

30–300 MHz VHF

300–1,000 MHz UHF

1–2 GHz L

2–4 GHz S

4–8 GHz C

8–12 GHz X

12–18 GHz Ku

18–27 GHz K

27–40 GHz Ka

40–300 GHz Mm

Table 2.8 ITU Radar Bands

Frequency Band

138–144 MHz216–225 MHz

VHF

420–450 MHz890 - 942 MHz

UHF

1.215–1.400 GHz L

2.3–2.5 GHz2.7–3.7 GHz

S

5.250–5.925 GHz C

8.500–10.680 GHz X

13.4–14.0 GHz15.7–17.7 GHz

Ku

24.05–24.25 GHz K

33.4–36.0 GHz Ka

59.0–64.0 GHz V

76.0–81.0 GHz92.0–100.0 GHZ

W

126.0–142.0 GHz144.0–149.0 GHz231.0–235.0 GHz238.0–248.0 GHz

mm

The selected United States radio frequency allocation from 30 MHz to 300 GHzaccording to the FCC’s “Online Table of Frequency Allocations,” 47 C.F.R., 2.106; and the selected European radio frequency allocation from 30 MHz to 300GHz according to the ERC Report 25, “The European Table of Frequency Alloca-tions and Utilizations in the Frequency Range 9 kHz to 1000 GHz”; are given at theend of this book in the Appendix A.


The European Table of Frequency Allocations and Utilizations Covering the Frequency Range 9kHz to 275 GHz, ERC Report 25, Copenhagen 2004, http://www.erodocdb.dk/docs/doc98/offi-cial/pdf/ErcRep025.pdf.“FCC Online Table of Frequency Allocations 47 C.F.R. § 2.106,” Revised on September 23,2008, http://www.fcc.gov/oet/spectrum/table/fcctable.pdf.Manual of Regulations and Procedures for Federal Radio Frequency Management, National Tele-communications and Information Administration, http://www.ntia.doc.gov/.

2.2 Spectrum Division 33

C H A P T E R 3

Electromagnetic Properties ofCommunications Systems

3.1 Fundamental Communications System Electromagnetics

This chapter will focus on the free space relations, leading into basic propagationtheory.

Electromagnetic waves are mostly characterized by their wavelength, as wehave seen in Chapter 2. They are also characterized by their frequency and energy.Every electromagnetic source whose characteristics change (oscillate) with time willproduce waves with certain properties. An electromagnetic wave is a propagatingelectromagnetic field through a medium. The speed of the wave depends on themedium through which it propagates. The wave is polarized depending on the ori-entation of its oscillation. The waves can carry energy from the source into themedium through which they propagate. Radiation is an example of this energytransfer. Electromagnetic waves propagate via reflection, refraction, diffraction,and dispersion.

The electromagnetic wave can propagate through different types of mediums:partial conductor, perfect dielectric (insulator), free space, and good conductor. Thewave consists of both electric and magnetic fields. The ratio of these two fields(impedance) depends on the losses in the medium.

If we are dealing with a partial conductor (i.e., seawater), the wave impedancewill be

ηωµ

σ ωε=

+j

j(3.1)

where σ is the conductivity in S/m, µ is the permeability (4π · 10−7 H/m), and ε is thepermittivity (8.852 · 10−12 F/m) of the medium. The phase velocity, ω equals 2πf,where f is the frequency of the wave. The angle between the electric and magneticfield, θ, is 0º < θ < 45º. The velocity of the wave is obtained from:

v = =

+ ⎛⎝⎜

⎞⎠⎟

+⎛

⎝⎜⎜

⎞

⎠⎟⎟

ω

βµε σ

ωε

1

21 1

2

(3.2)

where β = 2π/λ is the phase constant. Wavelength, λ, is calculated from:

35

λπ

βω

µε σ

ωε

= =

+ ⎛⎝⎜

⎞⎠⎟

+⎛

⎝⎜⎜

⎞

⎠⎟⎟

2 1

21 1

2

(3.3)

When dealing with a perfect dielectric, where the conductivity σ = 0, (3.1) issimplified, the wave impedance becomes

ηµ

ε= ∠0º (3.4)

In this case, there is no attenuation of the electric or magnetic component of theelectromagnetic wave, and they are in phase all the time, that is, θ = 0°.

Phase velocity is equal to

v = =ω

β µε

1(3.5)

and the wavelength

λπ

β

π

ω µε= =2 2

(3.6)

If the electromagnetic wave propagates through free space, then permeabilityand permittivity are:

µ µ π

ε ε

= = ⋅

= = ⋅

−

−

07

012

4 10

885 10

H m

F m.(3.7)

The wave impedance in this case is:

η π= ≈120 377Ω (3.8)

and the velocity is equal to the speed of light, that is,

v c= ≈ ⋅3 108 m s (3.9)

It is valid for a good conductor: σ >>ωε. Then, the spreading constant γ can bewritten as:

γ α β= + j (3.10)

where α is the attenuation constant and β is the phase constant as given before. Bothof them are equal to

α βωµσ

π µσ= = =2

f (3.11)

36 Electromagnetic Properties of Communications Systems

The wave impedance can be written as:

ηωµ

σ= ∠45º (3.12)

The wave entering the conductor is attenuated very fast. The intensity is attenu-ated with the factor e−αy. Both the electric and magnetic fields are attenuated to thevalue of 1/e, or 36.8% of the surface value at the depth for which αy = 1. This spe-cial value is defined as the skin depth δ and is calculated as

δπ µσ

= 1

f(3.13)

In summary, the medium is a good conductor if the loss tangent is large (σ >>ωε). If the loss tangent is very small (σ >> ωε), the medium is a good dielectric. Equa-tions for calculating the attenuation constant, phase constant, and impedance forvarious medium types are given in Table 3.1.

When a wave comes to the border of two mediums, one part will be reflected,and the other part will go through into the second medium. For the electric field inany point,

E E e E eE

eEj l j l g j l= ′ + ′ = +

⎡

⎣⎢

⎤

⎦⎥ + −

⎡

⎣+ − +γ γ γη

η

η

η21

211

2

1

2

R⎢

⎤

⎦⎥

−e j lγ (3.14)

is valid, where vectors E´ and E´´ represent the components of incident and reflectedwave on the border of two mediums, and l is the distance from the border. There-fore, the component of the incident and reflected wave can be written as:

′ = +⎡

⎣⎢

⎤

⎦⎥ ′′ = −

⎡

⎣⎢

⎤

⎦⎥E

EE

ER

RR

R

21

211

2

1

2

η

η

η

η, (3.15)

where η1 and η2 are the impedances of the medium 1 and 2. The reflection coefficientis obtained from

3.1 Fundamental Communications System Electromagnetics 37

Table 3.1 Attenuation Constant, Phase Constant, and Impedance for VariousMedium Types

Medium with Losses GoodConductor

GoodDielectric

FreeSpace

AttenuationConstant α ω

µε σ

ω21 1

2

+ ⎛⎝⎜

⎞⎠⎟

−⎡

⎣⎢⎢

⎤

⎦⎥⎥e

ωµσ

2≈ 0 0

PhaseConstant β ω

ωε σ

ωε21 1

2

+ ⎛⎝⎜

⎞⎠⎟

+⎡

⎣⎢⎢

⎤

⎦⎥⎥

ωµσ

2ω µε ω µ ε0 0

Impedance η jj

ωµ

σ ωε+( )ωµσ

21 + j µ

ε377

rE

ER

R

R

=′′

′=

−+

→

→

η η

η η2 1

2 1

(3.16)

and the transmission coefficient from

rE

ET

R

R

= ′′′

′=

+

→

→

2 2

1 2

η

η η(3.17)

The vector of the transmitted electric field is equal to the sum of the vector ofincident and the reflected wave:

′′′ = ′ + ′′→ → →

E E ER R R (3.18)

The power density of the electromagnetic wave is equal to

PE H

→

→ →

= =

2 2

2 2ηϕ η ϕcos cos (3.19)

where ϕ is the angle between the electric and magnetic field. The free space imped-ance is equal to

η =E

Hx

z

(3.20)

The wave propagates in the z direction, and electric (x) and magnetic (y) com-ponents are perpendicular to each other and to the direction of the wave propaga-tion. Power densities are correlated as follows:

′ = ′′+ ′′′→ → →

P P PR R R (3.21)

which means that the incident power is divided into reflected and transmittedpower.

If there is a need to calculate the input impedance on some distance l from theborder of two mediums (Figure 3.1), the following expression will be used

Z Zj l

j lin B= =++

ηη η β

η η β12 1 1

1 2 1

tan

tan(3.22)

The input impedance can be solved with the Smith chart as the propagationthrough the free space can be substituted with the transmission line, or more pre-cisely, on the circle of constant attenuation.


3.1.1 Smith Chart

Let a load ZR be connected to some voltage source Ug, with inner impedance Zg (Fig-ure 3.2). The line has impedance Z0. From the theory of networks, it is well knownthat most of the energy from the source will be given to the load if inner sourceimpedance is equal to the load impedance, and both of them are equal to the imped-ance of the line (Zg = Z0 = ZR). If a load is different on the transmission line (wave-guide or cable in real situations) from the line characteristic impedance (usually50Ω or 75Ω) connected, then not all of the energy from the source will be transmit-ted to the load.

There will be a reflection, which might even damage the source (generator). Theaim is to adapt the load to the generator with compensation elements that are eitherinductive or capacitive in character. In some special cases, only an open or shortedtransmission line of a certain length will be sufficient. This will perform the adapta-tion and then there will be no reflection on the load (usually an antenna) and noreturn of the power back to the transmitter.

The input impedance in the line can be found from the following expression:


A B

ZB

η1

ε2 ε1

η2

Figure 3.1 Electromagnetic wave at the border of two mediums.

ZR

Z0

Z0

Zin

Ug

Figure 3.2 Voltage source Ug, with inner impedance Zg.

Z ZZ jZ l

Z jZ linR

R

=++0

0

0

tan

tan

β

β(3.23)

where βl is the electric length of the line.The reflection coefficient, rR, determines how much of the power is reflected on

the load:

rZ Z

Z ZRR

R

=−+

0

0

(3.24)

The incident and the reflected wave result in the standing wave. The standingwave ratio (SWR or ρ) defines how much a load is adapted to the source:

SWR = =+−

E

E

r

rR

R

max

min

1

1(3.25)

SWR can also be defined as the ratio of the maximum electric field and mini-mum electric field. When SWR is equal to 1, it means that the reflection coefficientis equal to 0—which means that we have a perfect match. SWR cannot be smallerthan 1. This is the center of the Smith chart (Figure 3.3).

In the Smith chart, the upper part represents the inductive character (+j), and thebottom part the capacitive character (−j). The impedance of the short circuit is equalto 0 and can be found at the left side of the chart next to the wavelength of 0λ. Theadmittance of the short circuit is equal to ∞ and is found at the far right side next tothe wavelength of 0.25λ. The open line, in turn, has an impedance of ∞, next to thewavelength of 0.25λ, and the admittance is 0 with the wavelength of 0λ.


0λ

0

0.25λ

00

0.125λ

+ 1j

0.375λ

−j1

SWR=3

Curve ofconstant real part

1.0 3.0

Figure 3.3 The Smith chart.

If the load is reactance, it will be placed on the circle of reactive reactance (orcapacitive susceptance) on the top or bottom of the circle, depending on the value.The goal of matching (or adapting) the load is to get it as close as possible to the cen-ter of the chart. Usually, the values that are inserted into the chart are divided withcharacteristic impedance of the line, that is, Y=1/Z0. Moving along the transmissionline corresponds to moving along the circle with a center in the center of the Smithchart. This circle is called the circle of constant attenuation and is not drawn in theSmith chart. An individual solving the problem must draw it on his or her own. Thecircles of the constant real part (0.1, 0.3, . . . 1.0, 3.0, . . . 10.0, . . .) are drawn in thechart; they represent the point in the transmission line and differ only in reactance.The parts of the circles in the upper and lower part, which have a common reactancepart (± j) and different real part, are also shown.

The matching of the load to the generator impedance can be performed by add-ing one or more compensation elements, and in some cases with λ/4 transformers.These elements can be open line, short circuited, or have some reactance (C or L).Their purpose is to create a standing wave on the compensation element, so that thereflected wave from the compensation element and the load nullify each other(Figure 3.4).

Here, the circulation of the energy is at hand. Theoretically, there are no losseson the transmission line, since we assume that the line itself has no losses.

If the load has only resistors (no reactance) and is used at a fixed frequency, λ/4,transformers (Figure 3.5) can be used for impedance matching.

The input power (voltage and current are in phase) is equal to the one dissipatedon the load:

P U I U I I Z I ZU

Z

U

Zul Rul R

= = = = = =1 1 2 2 12

22 1

222

(3.26)

The characteristic impedance of the line is calculated from

Z Z Zin R0 = ⋅ (3.27)


Z0

Z0 Z0ZR

Energycirculation

Figure 3.4 Circulation of the energy on the compensation element.

3.1.2 Snell’s Law of Reflection and Refraction

If the wave is entering from one medium into the other at an angle rather than per-pendicular, there will be a transmitted component as well as a reflected componentof the incident wave, and their intensities will depend on the angle of incidence.

Snell’s law of reflection says that the angle of incidence θi will be equal to theangle of reflection θr, that is,

θ θi r= (3.28)

while Snell’s law of refraction says that the angle of refraction (transmitted wave),θt, and the angle of incidence, θi, will be

sin

sin

θ

θ

µ ε

µ εi

t

= 2 2

1 1

(3.29)

Total reflection appears when θt = 90º , which can happen only when a wavetravels from an electrically denser medium into a less dense medium. For instance,when the wave from the Teflon ( r = 2,1, µr = 1) enters free space, the critical anglewhen the total reflection will appear is

θε

εc = = =− −sin sin,

.1 2

1

1 12 1

4364º (3.30)

The electric component of the wave can be vertically or horizontally polarizedin regard to the incident plane. If it is vertically polarized (Figure 3.6), it is parallel tothe incident plane and the wave will be totally or partially reflected.

With horizontal polarization (Figure 3.7), the electric component lies in theincident plane, and with µ1 = µ2, there can be an incident angle where there will beno reflected wave (total transmission).

Total transmission exists only when a wave travels from an electrically lessdense to an electrically denser medium. This angle is called the Brewster’s angle andis defined as:


Zin

ZRU1 U2

I1

I2

λ/4

Figure 3.5 A λ/4 transformer.

θε

εB = −tan 1 2

1

(3.31)

For instance, when a wave travels from air to glass (εr = 5, µr = 1), the incidentwave at which there will be no reflection (total transmission) will be

θε

εB = = =− −tan tan .1 2

1

1 51

6591º (3.32)

If the angle of incidence is at an angle other than perpendicular, there are twocomponents of the wave: one which is transmitted, and the other vibrating at theborder, so that the following expression cannot be used for the reflectioncoefficient:

rR =−+

η η

η η2 1

2 1

(3.33)


H”

H’”

H’

θ

θθ

l

tr

E”E’”

E’

x

y

1 2

Figure 3.6 Vertical wave polarization.

H”

H’”

H’

θ

θθ

l

tr

E”

E’”

E’

x

y

1 2

Figure 3.7 Horizontal wave polarization.

Instead, the following can be used:

rZ Z

Z ZR =−+

η η

η η

2 1

2 1

(3.34)

The values Z η are different for vertical and parallel polarization. For verticalpolarization, it will be:

Z

Z

η

η

η

θ

η

θ

1

2

1

1

2

2

=

=

cos

cos

(3.35)

and thus, the reflection coefficient,

( )( )

rZ Z

Z ZR

n

n

=−

+=

−+

2

2

2 1

2 1

η

η

1

1

θ θ

θ θ

sin

sin(3.36)

For parallel polarization it will be:

Z

Zη

η

η θ

η θ1

2

1 1

2 2

=

=

cos

cos(3.37)

and the reflection coefficient is

( )( )

rZ Z

Z ZR =−+

= −−−

η η

η η

θ θ

θ θ2 1

2 1

2 1

2 1

tan

tan(3.38)

Other types of electromagnetic wave propagation are diffraction and disper-sion. Diffraction is the bending of waves around small obstacles and the spreadingout of waves past small openings. With dispersion, an electromagnetic wave is sepa-rated into components with different wavelengths due to refraction, interference, ordiffraction. Diffraction will be covered in more detail in Section 3.2.4.

3.2 Wave Generation and Propagation in Free Space

The propagation of electromagnetic waves deals with the way the wave travels fromthe transmitting antenna to the receiving antenna. The electromagnetic waves cantravel through guided structures like transmission lines, waveguides, and free space.This will be the subject of interest in this section.

3.2.1 Maxwell’s Equations

Maxwell’s equations can be written in integral or differential form. Written in inte-gral form they look as follows:


E dAq⋅ =∫ ε0

(3.39)

B dA⋅ =∫ 0 (3.40)

E dld

dtB⋅ = −∫

Φ(3.41)

B dl ic t

E dA⋅ = + ⋅∫∫ µ∂

∂0 2

1(3.42)

Written in differential form the equations look as follows:

∇ ⋅ =D ρ (3.43)

∇ ⋅ =B 0 (3.44)

∇ × = −EBt

∂

∂(3.45)

∇ × = +H JDt

∂

∂(3.46)

where E is the electric field strength in V/m; H is the magnetic field strength in A/m;D is the electric flux density in C/m2 (Coulombs); B is the magnetic flux density inWb/m2 (Webers); J is the conduction current in A/m2; and v is the electric chargedensity in C/m3. Additional equations describing the relation for the medium are:

D E= ε (3.47)

B H= µ (3.48)

J E= σ (3.49)

where ε = ε0εr is the permittivity, µ = µ0µr is the permeability, and σ is the conductiv-ity of the medium.

The first Maxwell equation (3.39) is also called Gauss’ law for electricity. It saysthat the electric flux out of any closed surface is proportional to the total chargeenclosed within the surface. In the second (3.40), Gauss’ law for magnetism saysthat the net magnetic flux out of any closed surface is zero. Equation (3.41), or Fara-day’s law of induction, says that the line integral of the electric field around a closedloop is equal to the negative of the rate of magnetic flux change through the areaenclosed by the loop. Equation (3.42), or Ampere’s law, says that in the case of a

3.2 Wave Generation and Propagation in Free Space 45

static field, the line integral of the magnetic field around a closed loop is propor-tional to the electric current through the loop.

3.2.2 Wave Propagation

If it is assumed that the wave propagates in the z direction and the wave is polarizedin the x direction, the values of the electric and magnetic field will depend on the dis-tance and time according to:

( ) ( )E z t E e t z azx, cos= −−

0α ω β (3.50)

( ) ( )H z tE

e t z azy, cos= − −−0

ηω β θα

η (3.51)

where η is the intrinsic impedance of the medium calculated from

ηµ ε

σ

ωε

θσ

ωεθ=

+ ⎛⎝⎜

⎞⎠⎟

= ≤ ≤

4 1

2 0 451

4

, tan , º (3.52)

Equations (3.50) and (3.51) show the attenuation of the electromagnetic wavewhile it propagates through a medium with the factor e az (Figure 3.8).

Power density of the electromagnetic wave is

PE

e axz= −0

22

2 ηθα

ηcos (3.53)

In free space the E and H fields are perpendicular to each other and to the direc-tion of wave propagation.

The loss of the electromagnetic wave in the z direction happens due to the rela-tive permittivity of the medium.


E

z

e−αz

H

y

x

Figure 3.8 Components of the electric and magnetic fields in a lossy medium.

3.2.3 Wave Polarization

The electric field can have various orientations depending on the transmittingantenna. The polarization is the orientation of the tip of the electric field in a planeperpendicular to the direction of the propagation at some point in space as a func-tion of time. The types of polarization are linear (vertical or horizontal), circular,and elliptic. Under extreme conditions in the atmosphere (e.g., rain), electromag-netic wave depolarization is possible. Depolarization can also happen fromreflections.

With linear polarization, the orientation of the field is constant in space andtime. For a wave traveling in the z direction, the electric field can be written as

E E a E ax x y y= + (3.54)

where

( )E a t kzx a= − +cos ω φ (3.55)

( )E b t kzy b= − +cos ω φ (3.56)

The trajectory of the electric field vector E will be drawn on the x, y plane. Thetip of the electric field vector moves as time goes by.

The polarization will be linear when phase angles a and b are equal and thetrajectory is a line. The phase difference between the angles must be

∆φ φ φ π= − = =b a n n, , , ,0 1 2 (3.57)

Linearly polarized waves can be generated using simple antennas such asdipoles. Figure 3.9 shows the linear polarization.

Circular polarization will have different phase angles φa and φb, but the ampli-tudes a and b will be the same. This results in a circle trajectory. The phase differ-ence between the angles is

∆φ φ φ π= − = ± +⎛⎝⎜

⎞⎠⎟

=b a n n12

2 0 1 2, , , , (3.58)


x

y

E

Figure 3.9 Linear polarization.

Circularly polarized waves can be generated by a helically wound wire antennaor with two linear sources perpendicular to each other. Figure 3.10 shows the circu-lar polarization.

Both the linear and circular polarizations are special cases of elliptical polariza-tion. Elliptical polarization will occur when the phase angles φa and φb, as well asthe amplitudes a and b, are different. The trajectory in this case is elliptical. Thephase difference between the angles is the same as in the circular polarization. Fig-ure 3.11 shows the elliptical polarization.

3.2.4 Fresnel Knife-Edge Diffraction

Diffraction occurs when an electromagnetic wave encounters an obstacle. The wavewill bend around the obstacle and continue to spread. If there is no obstacle, theelectromagnetic wave will travel in a straight line from the transmitter to thereceiver. However, if there are obstacles near the path, they will influence the waveby possible power reduction or phase distortion. Fresnel’s zones are ellipsoids (Fig-ure 3.12) where obstacles can create signals that will be out of phase.

The first Fresnel zone creates signals that are 0° to 90° out of phase; the secondzone creates signals that are 90° to 270° out of phase; the third zone creates signalsthat are 270° to 450° out of phase, and so forth. Odd number zones are construc-


x

y

E

Figure 3.10 Circular polarization.

x

y

E

Figure 3.11 Elliptic polarization.

tive, since they reinforce the signal, and even numbered zones are destructive, sincethey destroy the signal. The obstacles in the first zone are potentially the mostdangerous ones.

The radius r1 of the first Fresnel zone is calculated from

rd d

d d11 2

1 2

=+

λ(3.59)

where d1 and d2 are the distance between the obstacle and the transmitter andreceiver. The above expression is valid when the distances d1 and d2 are much largerthan r1. To achieve communication, it is desirable to have the first Fresnel zone clearof any obstacles—to be more precise 60% of the first Fresnel zone should be clear ofobstacles, meaning a radius of 0.6 r1.


T

First Fresnel zone

Second Fresnel zone

Third Fresnel zone

Fourth Fresnel zone

Rr1

d1 d2

Figure 3.12 Fresnel zones.

T Rh<0

d2d1

α<0

T Rh<0

d2d1

α<0

Figure 3.13 Knife-edge diffraction.

Figure 3.13 shows the knife-edge diffraction from an obstacle in the line ofsight. This obstacle can be above or below the line of sight.

The Fresnel-Kirchoff diffraction parameter v is the dimensionless quantity,which can be calculated from

( )v h

d d

d d=

+2 1 2

1 2λ(3.60)

which depends on the distances from the transmitter and receiver to the tip of theobstacle, height of the obstacle, and wavelength. The value of h can be positive ornegative.

The value of v can also be calculated from the angle of diffraction α, which canbe either positive or negative, like the height h.

( )v

d d

d d=

+α

λ

2 1 2

1 2

(3.61)

The diffraction loss for knife-edge obstacle can be calculated from

( )

( )( )

A v

v

v v

e vv

=

≤ −− − ≤ ≤⋅ ≤ ≤−

0 1

20 05 062 1 0

20 05 0 1

2

0 95

log . .

log . .

( )0 04 01184 038 01 1 2 4

200225

2log . . . . .

log.

− − −⎛⎝⎜

⎞⎠⎟ ≤ ≤

⎛

v v

v⎝⎜⎞⎠⎟

≤

⎧

⎨

⎪⎪⎪⎪

⎩

⎪⎪⎪⎪

⎫

⎬

⎪⎪⎪⎪

⎭

⎪⎪⎪⎪

v 2 4.

dB (3.62)

The results of losses (dB) are shown in Figure 3.14.


−25

−20

−15

−10

−5

−2,0 −1,0 0,0 1,0 2,0 3,0

0

v

Av(),

dB

Figure 3.14 Knife-edge diffraction loss.

Most obstacles in real situations are large in comparison to the signal wave-length and are not knife-edge. In such situations different models for path loss pre-dictions are used. With higher frequencies, the first Fresnel zone gets smaller andsmaller. However, when the Fresnel zones are smaller, the diffraction loss willbecome greater if the receiver antenna is lowered.

3.2.5 Path Loss Prediction

Path loss or attenuation is the reduction of the power density of an electromagneticwave as it travels through space. When calculating the link budget, the path loss isof great importance. It is calculated for free space, but different factors such asreflection, absorption, and refraction influence its value. Terrain is also of impor-tance, so different models are used for urban, semiurban, or rural terrain.

Path loss is calculated in free space from

Ld

f = ⎛⎝⎜

⎞⎠⎟

204

10logπ

λ(3.63)

where d is the distance between the transmitter and the receiver and λ is the wave-length. It is usually given in dB/km or dB/m. In closed areas (buildings) the addi-tional loss is 1 dB/m (i.e., an office). The exact path loss will depend on the actualsituation, width of the walls, and so forth. The path loss for 2.4 GHz is shown inFigure 3.15 for free space and inside the building.

There are several models for path loss in an urban area. The most popularmodel is the Hata model for urban areas. In cities, there is almost never a line ofsight (LOS) between the transmitter and a receiver. The Hata model parameters are:

• d: the distance from the transmitter to the receiver (1–20 km);


L(d

B)f

d (m)

0 20

20

40

40

60

60

80

80

100

In building

Free space

120

140

160

Figure 3.15 Path loss for 2.4 GHz.

• f: the frequency in MHz (100–1,500 MHz);• hb: the base station height (30–200m);• hm: the mobile station height (1–10m).

The mean path loss is given by empirical equation:

( )( )[ ] ( )

L f h

h d a

S b

b

= + − +

− −

6955 2616 1382

449 655

. . log . log

. . log log ( )h Lm x− α

(3.64)

in an open, suburban, or medium-size city

( ) ( )[ ] ( )a h f h fm m= − − +11 07 156 08. log . . log . (3.65)

and in a big city

( ) ( )( )a h

h f

h fmm

m

=− ≤

−829 154 11 300

32 1175 497

2

2

. log . .

. log . .

MHz

≥⎧⎨⎩ 300 MHz

(3.66)

The correction factor is

( )( )L

f

fcor =+

−2 28 5 4

478 1833

2

2

log .

. log . log

in the suburbs

( )f +⎧⎨⎩ 4094. in the open

(3.67)

Figure 3.16 depicts an example of path losses for open space, a suburbanmedium sized city, and a large sized city, versus distance (in kilometers). The heightsof the transmitter and receiver in this example are 50m and 2m, respectively. Thefrequency is 900 MHz. The medium and large city path losses are almost the samefor this example.


080

100

L(d

B)S

120

140

160

180

2 4 6 8 10 12 14 16 18 20

d (km)

Big city

Medium city

Suburbs

Open area

Figure 3.16 Hata model path loss.

The reflections from the obstacles or objects produce multiple paths or fading.Usually 30 dB is considered enough to raise the transmitted power (or some othermeans) for compensation. Spatial or frequency diversity is used to battle this prob-lem. Many other models are used to calculate the path loss, including the IrregularTerrain Model (also known as the Longley-Rice code), which is a model of radiopropagation for frequencies between 20 MHz and 20 GHz. The Longley-Ricemodel predicts the median attenuation of a radio signal as a function of distance andthe variability of the signal in time and in space. A more precise evaluation can onlybe obtained with test measurements in the field. This is more expensive but givesbetter insight into the problem.

3.3 Wave Generation and Propagation in the Terrestrial Atmosphere

In the atmosphere there are gases: mainly nitrogen, oxygen, and carbon dioxide.The atmosphere is also influenced by gravity. Near the Earth’s surface, density andpressure are higher than at higher altitudes away from the Earth. The influence onthe propagation is the highest closest to the Earth. The atmosphere is divided intolayers of which the most important are: the troposphere, stratosphere, and iono-sphere. The troposphere is about 11 km high, depending on geographical latitude. Itis the warmest, wettest, and densest layer of the atmosphere, with the greatest influ-ence on communication systems, mostly due to rain. The next layer is the strato-sphere, which reaches up to 50 km high. It is much colder than the troposphere anddoes not influence microwave transmissions very much. The ionosphere is the nextlayer, and has three parts: the D, E, and F regions. The D region is 75 to 95 km awayfrom the Earth’s surface and has weak ionization. The E region is 95 to 150 kmaway from the Earth’s surface. The F region is 150 to 6,000 km from the Earth’ssurface and has the most electrons of all three regions. It is the most importantregion for communications.

The density and refractive index, which change with altitude and weather con-ditions, and the curvature of the Earth can influence the communication links forsatellite and microwave applications over large distances. This means that commu-nication is possible, even if there is no line of sight (LOS) between the transmitterand the receiver.

3.3.1 Absorption and Scattering

Absorption and scattering are the main sources of losses in the troposphere.Absorption occurs because atmospheric gas molecules resonate at some frequencies(i.e., water vapor molecules resonate at 22.235 GHz and oxygen molecules at 60GHz). The absorption is always present, although it can depend on the humidity.More about absorption in the atmosphere will be discussed in Chapter 4.

Scattering occurs when an electromagnetic wave collides with atmosphere par-ticles. If these particles are smaller than the wavelength, Rayleigh scattering willhappen. The particles reflect some of the energy, depending on the size and dielec-tric property. They can be dust, nitrogen, or oxygen molecules. In addition, Miescattering occurs when the particles in the atmosphere are about the same size as the

3.3 Wave Generation and Propagation in the Terrestrial Atmosphere 53

wavelength. These particles are dust, pollen, water vapor, and smoke and are usu-ally present in the lower parts of the atmosphere, especially if there are clouds. Thelast type is nonselective scattering, which occurs when the particles, usually largedust or rain drops, are much larger than the wavelength.

3.3.2 Wave Propagation in the Atmosphere

There are three possible types of wave propagation over the Earth (Figure 3.17):

• Surface propagation along the surface of the earth;• Wave propagation through the troposphere;• Propagation by reflection from the ionosphere.

The ionosphere refracts the wave back to the earth in a frequency range up toapproximately 50 MHz. The surface wave can be used up to 5 MHz. The surfacewave is attenuated more than the wave traveling through free space, so the transmit-ters in these bends must have a higher transmitting power. In free space, beside thedirect wave there is usually at least one reflected wave.


Transmitter Receiver

Earth

Surface wave

Reflected wave

Troposphere Direct wave

Reflection from ionosphere

Ionosphere

Figure 3.17 Wave propagation in the atmosphere.


Barclay, L. W., Propagation of Radio Waves, 2nd ed., London, U.K.: Institution of ElectricalEngineers, 2003.Deal, W. R. et al., “Guided Wave Propagation and Transmission Lines,” in RF and MicrowaveHandbook, M. Golio, (ed.), Boca Raton, FL: CRC Press, 2001.Lee, W. C. Y., Mobile Communications Engineering, New York: McGraw-Hill, 1982.Magnusson, P. C., et al., Transmission Lines and Wave Propagation, 4th ed., Boca Raton, FL:CRC Press, 2001.Rappaport, T. S., Wireless Communications: Principles and Practice, Upper Saddle River, NJ:Prentice-Hall, 2001.Rothwell, E. K., and M. J. Cloud, Electromagnetics, Boca Raton, FL: CRC Press, 2001.Sadiku, M.N.O., and K. Demarest, “Wave Propagation,” in Electrical Engineering Handbook,Dorf, R. C., (ed.), Boca Raton, FL: CRC Press, 2000.Solheim, F. S., et al., “Propagation Delays Induced in GPS Signals by Dry Air, Water Vapor,Hydrometeors, and Other Particulates,” Journal of Geophysical Research, Vol. D8, April 1999,pp. 9663–9670.Smrkic, Z., Mikrovalna Elektronika, Skolska Knjiga, Zagreb, 1986.

3.3 Wave Generation and Propagation in the Terrestrial Atmosphere 55

C H A P T E R 4

Electromagnetic Interference

4.1 Electromagnetic Interference with Wave Propagation andReception

Electromagnetic interference exists in every communication link. It manifests itselfas noise, which degrades the quality of the application. In analog systems, tradition-ally the signal-to-noise (S/N) ratio is used to show the quality of the communicationlink. In every case, the signal level should be above the noise for communication tobe possible. How much above depends on the quality of the receiver used. In digitalsystems, especially where the spread spectrum is used, the ratio S/N is not the bestparameter to evaluate link quality, since the signal is almost always buried in thenoise—but this does not mean that communication will be impossible. In this case,other parameters such as the energy of the bit compared to the noise spectral density(Eb/N0), are much better to use regarding the quality of the communication.

Any signal, although intentional and useful, is considered noise to other signalsin the same channel or frequency band. This is why careful planning and good fre-quency allocation is necessary. In some cases even neighboring countries must worktogether, because electromagnetic signals are not bound to national borders.

There are several types of interference or noise, which are either natural or man-made. Natural interference includes phenomena such as lightning or electrostaticdischarge, atmosphere effects, sunspot activity, and reflections from the roughEarth surface. Manmade interference comes from both commercial and militarycommunications such as radar, radio, television, and cell phone communications.Industry can also create interference. All this interference is unintentional, but therecan also be intentional interference, especially during a war.

4.1.1 Additive White Gaussian Noise (AWGN)

Additive white Gaussian noise (AWGN) is a statistically random noise in the widefrequency range (very low frequencies up to 1012 Hz) with constant spectral density.AWGN can come from many sources such as thermal noise, shot noise, noise fromSun radiation, and others. It is a background noise in the communication channel.

If in the communication channel (Figure 4.1) a signal s(t) is introduced, it will beadded by additive white Gaussian noise n(t):

( ) ( ) ( )r t s t n t= + (4.1)

57

At the receiver the signal r(t) will be received. There, in the process of detection(see Section 6.1), the decision about the value of the signal will be made.

It is possible to use this model only in deep space communications (i.e., betweensatellites), where the only degradation in the channel is caused by the thermal noisein electronic devices. In real situations multipath, fading, dispersion, and other fac-tors must be included.

4.1.2 Thermal Noise

Conductor resistivity used for the flow of electrons depends on temperature. Thus,temperature will have an influence on the noise in the communication channel. Thethermal noise Pterm, in [W] (sometimes defined as Nt), is defined as

P kTBterm = (4.2)

where k is the Boltzmann 1.38 · 10−38, T is the temperature in [K], and B is the fre-quency bandwidth in [Hz]. Thermal noise exists in every communication systemand cannot be avoided.

4.1.3 Shot Noise

Shot noise appears in electrical circuits where direct current (DC) flows. It repre-sents small variations of the current. This noise does not depend on temperature.The noise current In, in [A], is defined as

I qI Bn DC= 2 (4.3)

where q is charge of the electron, 1.6 · 10−19C, IDC is a DC bias current in the electriccircuit, and B is the frequency bandwidth in [Hz].

4.1.4 Flicker (1/f ) Noise

Flicker noise is proportional to the bias current and decreases with frequency. Itspower density is proportional to 1/f, and falls by approximately 10 dB per decade.Flicker noise is weak above several kilohertz and is sometimes called pink noise.

58 Electromagnetic Interference

r t( )s t( ) Receiver

n t( )

Figure 4.1 AWGN channel communication model.

4.1.5 Burst Noise

Burst noise appears in semiconductors and is also called popcorn noise. It is causedby defects in the manufacturing process like heavy metal ion contamination or sur-face contamination. The noise increases with the bias current level and is propor-tional to 1/f2.

4.1.6 Noise Spectral Density

Noise spectral density, N0, is the noise in the frequency range of 1 Hz:

NP

BkTterm

0 = = (4.4)

In digital systems, energy per bit, Eb, is often used with noise spectral density forevaluating data (bit) error rate performance (BER). It can be found using the signalto noise ratio by:

E

NSN

BR

b

0

= ⋅ (4.5)

where R is the data rate and B is the frequency bandwidth.

4.1.7 Effective Input Noise Temperature

The effective input noise temperature, Te, is defined as the temperature at which theinput impedance has to be placed in order to generate the observed noise power atthe output of a two-port network or amplifier. It is calculated as

( )T NFe = −290 1 (4.6)

where NF is the noise factor [defined in (1.19)] at 290K.This parameter is often used for satellite communications where antennas are

pointed to the cold sky, and the temperature of 290K (used for most terrestrial com-munications) is not applicable.

4.2 Natural Sources of Electromagnetic Interference

4.2.1 Lightning and Electrostatic Discharge

Lightning and electrostatic discharge are examples of transients. Transients can becreated from guided or radiated emissions from electromechanical or electronicdevices, or from natural interference or discharges. They often appear as a result ofcurrent changes in inductive loads such as engines or relays. They can be created byradar as well as isolators in high voltage conductors during bad weather and can bedangerous, as the semiconductor could burn, the capacitor could explode, and thewire or transformer isolation could break down. Transients rise quickly and fallslowly (ratio of one to hundred). The rise time ranges from a nanosecond to a milli-

4.2 Natural Sources of Electromagnetic Interference 59

second. The amplitudes can be from below one volt up to more than one hundredkilovolts.

4.2.1.1 Lightning

Lightning is a transient electric discharge, the path of which is measured in kilome-ters. It appears when a part of the atmosphere becomes electrically charged enoughto allow electric breakdown in the air. It is the strongest natural force. In most cases,lightning appears in clouds, but it is also possible in snowstorms, desert storms, andabove erupting volcanoes. It can very rarely appear on mountains or tall TV towers.It can strike the same place several times during the same storm.

The lightning waveform is shown in Figure 4.2. An understanding of the wave-form is necessary to create the protection system. The pulse can be divided into threeparts (I to III). The first component (initial stroke) is a pulse of strong DC current,which can reach more than 200 kA and last about 200 µs. The rise speed is about 3 ·1010 m/s. The second component is an intermediate phase with a current level of sev-eral kA. It lasts about 5 ms. The third component has a current of around 400A andlasts about 0.75 second. After that, the first component can appear again (restrike)with an intensity of half as much as the initial stroke and of the same length. Usuallythere are a few of restrikes, each of lower intensity.

A lightning strike can cause potential difference between buildings. This poten-tial difference can be up to 1 MV. Figure 4.3 shows the potential difference betweentwo buildings as the result of a lightning strike.

When lightning strikes a high voltage post 150m from the building, is theregoing to be any damage to the cable connecting the two buildings? Let us assumethat the other building is 75m away from the first building. If the resistivity of theground, ρ, is 1 kΩ/m and the current is 200 kA, what will be the potential differencebetween the two buildings?

The potential is calculated from the following equation:


0.4

~5

200

I, kA

I

t, ns

II

III

0 200 5 10×3

0.75 10×6

Figure 4.2 Lightning waveform.

VI

d d d= −

+⎛⎝⎜

⎞⎠⎟

ρ

π21 1

1 1 2

(4.7)

where d1 is the distance from the first building to the post and d2 is the distancebetween the two buildings.

The potential for the above values will be 70.77 kV. This can be enough to dam-age the isolation of a communication cable between the buildings, which can be pre-vented by connecting the buildings with a conductor having a small impedance inthe frequency range of 300 kHz (lightning strike). Inside this conductor all the com-munication cables are placed. Thus, the lightning currents will flow on the outersurface, which protects the interior and the communication cables. The previouslymentioned skin depth (3.13) prevents the currents from going too deep into theconductor.

Lightning protection grounding must be performed with care. Typically it con-sists of guides with small impedance. There can be several going in parallel from thetop to the bottom of the building. Usually they are made of aluminum and copper,not only because of their electrical characteristics, but also because they are rustresistant. Transient voltages can enter a building through electrical, cable TV,phone, or internet lines. If the antenna on the roof is protected, the rest of the build-ing is not. The cables through which lightning current flows produces a magneticfield, and if it is large enough to encompass other conductors, magnetic couplingcan occur. The cables can also conduct the lightning currents to other electronicsystems and damage them.

If a tree is close to a building and lightning strikes the tree, it can conduct cur-rents to the building. Trees have a relatively large impedance compared to thegrounding protection. The diagram in Figure 4.4 shows the situation when light-ning hits a tree.

The lightning is a current source, and the tree has impedance Z, so potential willoccur on the tree. If it exceeds one million volts, the current can go to the objects inthe vicinity. If the typical current is 20 kA and the impedance of the tree 100Ω, thevoltage on the tree will be 2 · 106V, which can propel a person standing even 2mfrom the tree.


Building 1 Building 2

Grounded metal conductor

d2d1

Figure 4.3 Potential difference between buildings from lightning.

4.2.1.2 Electrostatic Discharge (ESD)

Electrostatic discharge is a fast spontaneous transmission of the electrostatic chargeinduced from the electrostatic field. The charge is transferred over the spark (staticdischarge) between two bodies with different electrostatic potentials when they areclose to each other.

Electrostatic discharge exists everywhere in our surroundings. How many timeshave we felt it when we touch a door handle or a metal chair? Even though this dis-charge cannot harm humans, it can be devastating to electronic equipment sensitiveto ESD. Even the ESD that we do not feel at all is dangerous to equipment. Table 4.1shows the typical sources of static electricity, and Table 4.2 gives the typical situa-tions that generate the electrostatic voltages.


d

Z

I

Figure 4.4 Lightning hitting a tree.

Table 4.2 Typical Situations that Generate Electrostatic Voltages

Static Discharge TypeRelative Humidity10%–20%

Relative Humidity65%–90%

Walking on the carpet 35,000V 1,500V

Walking on the pvc floor 12,000V 250V

Plastic foil 7,000V 600V

Worker at the desk 6,000V 100V

Table 4.1 Typical Sourcesof Static Electricity

Object Material

Floor PVCConcrete

Clothes ShoesWorksuit

Chair WoodPlastic

Packagingroom

Cathode rays

Moisture is an important factor. It is best to have the relative humidity between40% and 60% in the working area.

The damage from ESD occurs when a person or an object comes into contactwith an electronic device sensitive to electrostatic discharge. If this discharge hasenough energy, overheating damage can occur. Generally, the more sensitive theequipment is, the more vulnerable it will be to ESD. The damage can be immediate,where the electronic device is damaged or destroyed right after ESD, or latent,where the electronic device appears to be working normally, but the circuitry isdamaged and could stop operating at any moment.

Protection can be done on several levels. The first is in the working area. Elec-tronic devices sensitive to ESD should be operated in places where there is no ESD.Antistatic wrist tape (Figure 4.5) should be worn if available.

Additionally, an air ionizer can be used. Ions are created in nature by oceanwaves, waterfalls, and so forth. They purify the air from dust, smoke, or pollen. Ifwe had an EMC laboratory in the open near a waterfall, there would be no troublewith ESD. However, in big cities pollution is greater, so additional air ionizers couldbe an option, as they are commercially available. Normally, sources of static elec-tricity should be at least 1m away from sensitive equipment.

Figure 4.6 shows a work area protected from ESD.The table is covered with material absorbing static charge through a 1 MΩ

resistor, which protects the operator from shock if the Earth becomes electricallyalive. The mat under the table is also of a similar material to the cover on the table.Ground points (Gp) can also have connectors for the antistatic wrist tape and forother electronic equipment that are being tested.

The materials used for carpet and table surfaces should not be made of a metal,like stainless steel, because of their low resistivity, which could lead to transient dis-charges of electricity. Fast discharge is much more dangerous for electronic devicesthan discharge through static dissipative materials, which should have resistivityvalues in the range of 105 to 1011Ωm.

Second, before operating sensitive equipment, a person should discharge him-self or herself from all static electricity. This can be done with the previously men-tioned antistatic wrist tape or by touching a conductive surface. There are alsoantistatic suits that can be worn.

Last, devices sensitive to ESD should be placed in antistatic bags or containersduring transportation and storage.


Figure 4.5 Antistatic wrist tape.

Figure 4.7 shows the symbol for ESD danger, which is placed on packages con-taining sensitive electronic equipment.

Electrostatic field measurement equipment is also commercially available. Itcan usually measure voltages up to 30 kV.

4.2.2 Multipath Effects Caused by Surface Feature Diffraction andAttenuation

The path of the electromagnetic wave between the transmitter and receiver is rarelydirect. There is almost always a multipath. The propagation with the presence of amultipath is different from the propagation in ideal free space conditions. There areat least two paths: the direct path and the reflected path (from the atmosphere andEarth) as shown in Figure 4.8.

The reflected component has two parts: coherent and noncoherent. The coher-ent part is determined in regards to amplitude, phase, and direction. It follows theSnell law. The noncoherent part is subject to random characteristics of the scatter-ing terrain and is not deterministic. It is not a plane wave and does not follow theSnell law. It does not come from a certain direction but from the continuum. Table4.3 gives the electric properties of various types of terrain. The surface wave propa-gates best over sea water, and worst over dry terrain.


Gp

Figure 4.6 Working area protected from ESD.

Figure 4.7 ESD danger symbol.

The phase difference, ∆, between the direct and reflected path is calculated from

( )∆ = + −21 2

π

λd d d (4.8)

where d1 and d2 are the reflected paths and d is the direct path.Whether the terrain is smooth or not will depend on the Rayleigh criterion:

h ≥ λ

8sin Ψ(4.9)

where h is the height of the terrain roughness, λ is the wavelength, and Ψ is the angleof wave incidence (Figure 4.9).

If the above condition is fulfilled, the terrain is rough—otherwise it is smooth.In other words, if h is small enough, the dominant reflection will be coherent.

4.2.3 Attenuation by Atmospheric Water

The effect of atmospheric hydrometeors is of major concern for satellite to Earthpropagation. The main hydrometeors that exist are rain, snow, and dust particles.Rain is the major obstacle because it causes attenuation, phase difference, and depo-larization of radio waves. For analog signals, rain is most significant at frequenciesabove 10 GHz, and for digital signals above 3 GHz. The loss due to rain is given by


Transmitter

Flat earth

Reflected path

Direct path

Receiver

Curved earth

d

h

h

2

1

d

d

2

1

Figure 4.8 Multipath from Earth.

Table 4.3 Dielectric Properties of Various Earth Types

Earth Type Permittivity r Conductivity

Sea water 80 5

Fresh water (river, lakes) 80 0.005

Moist Earth 15–30 0.005–0.01

Rocky terrain 7 0.001

Dry terrain 4 0.001–0.01

( ) ( ) ( )L R l R p Rc= ⋅ ⋅γ (4.10)

where γ is the attenuation per unit length at rain rate R, le is the equivalent pathlength at rain rate R, and p(R) is the probability in percentage of rainfall rate R.

The attenuation depends on the rain rate, size, temperature, and refractiveindex of the water. Attenuation is calculated from

( ) [ ]γ R a Rb= ⋅ dB km (4.11)

where a and b are constants depending on the frequency. At 0°C, the values of a andb are obtained from

a G f

b G f

aE

bE

a

b

= ⋅

= ⋅(4.12)

where the values for Ga, Ea, Gb, and Eb are given in Tables 4.4 and 4.5.The effective length le(R) is used because the rain intensity is not the same over

the whole path. It depends on the local climate conditions. It can be approximatedfrom

( ) ( )[ ]I R R Re = + −−

00007 0232 0000180 766 1. . . sin. θ (4.13)


h

Figure 4.9 Reflection from the rough terrain.

Table 4.4 Values of Ga and Ea

Frequency Ga Ea

f < 2.9 GHz 6.39 · 10−5 2.03

2.9 GHz ≤ f ≤ 54 GHz 4.21 · 10−5 2.42

54 GHz ≤ f ≤ 180 GHz 4.09 · 10−2 0.6999

f 180 GHz 3.38 −0.151

Table 4.5 Values of Gb and Eb

Frequency Ga Ea

f < 8.5 GHz 0.158 0.158

2.9 GHz ≤ f ≤ 54 GHz 1.41 −0.078

25 GHz ≤ f ≤ 164 GHz 2.63 −0.272

f > 164 GHz 0.616 −0.0126

where θ is the elevation angle.The probability of the rainfall rate R in percentage is determined by:

( ) ( )p RM

e e eR R= ⋅ ⋅ + − ⋅ + ⋅− − −

8766003 02 1 1860 03 0 258 1 6

.. . .. . .β β ( )[ ]3 ⋅R (4.14)

where M is the mean annual rainfall accumulation in [mm] and β is theRice–Holmberg thunderstorm ratio.

Other hydrometeors like snow, vapor, or ice have similar characteristics as rain,but are at least one order of magnitude smaller. Figure 4.10 shows the rain attenua-tion versus frequency and rainfall rate.

The size of raindrops depends on weather conditions and rainfall rate. They aregiven in Table 4.6. The larger the size of the raindrops, the higher the attenuation.

The effects of rain can be lessened by using the circle polarization. With circlepolarization, the polarization of the electromagnetic wave changes during oneperiod of the wave. It can be clockwise or counterclockwise. In both cases, the trans-mitter and receiver must be synchronized. Circle polarization is widely used in satel-lite communications.

4.2.4 Attenuation by Atmospheric Pollutants

Beside rain and other hydrometeors, other particles can also influence the propaga-tion of the electromagnetic wave. The atmosphere consists of several gasses given inTable 4.7.

As can be seen from the Table 4.7, the atmosphere mainly consists of nitrogenand oxygen. While hydrometeors influence the propagation of the electromagnetic


Frequency (GHz)10

10

Att

enua

tion

(dB/

km)

100

1

2.5 mm/hr

5 mm/hr

12.5 mm/hr

25 mm/hr

50 mm/hr

100 mm/hr

Rain rate 150 mm/hr= 20 degrees Celsiust

100 30010.1

Figure 4.10 Rain attenuation versus frequency and rainfall rate.

wave by attenuation and scattering, gasses mostly only attenuate the EM waves.Oxygen has a small permanent magnetic moment, which results in small attenua-tion above 30 GHz. Water vapor can also attenuate EM waves above 10 GHzbecause it has a permanent electric dipole. The amount of water vapor can varyfrom 1 mg/m3 in cold dry climates to 30 g/m3 in hot humid climates. This means thatin deserts there is almost no water vapor present, whereas in rain forests it can makeup about 4% of the atmosphere. Nitrogen, on the other hand, has no permanentelectric or magnetic dipole, so it does not attenuate EM waves. Most gases have anegligible influence below 30 GHz.

4.2.5 Sunspot Activity

A sunspot is a region on the Sun surface near the equator consisting of magneticactivity with reduced surface temperature (4,000K compared to the surrounding5,800K). They are visible from the Earth without a telescope. Their numbers andsize rise and fall every 11 years. The sunspots have influenced Earth’s climatethroughout history. They do not influence solar radiation much, but their magneticactivity influences the ultraviolet and soft X-ray emission levels. They also emitions, the amount of which depends on the sunspot activity. Both X-rays and ions are


Table 4.7 Dry Atmosphere Constituentsfrom Sea Level to 90 km High

ParticleVolumePercentage

WeightPercentage

Nitrogen 78.088 75.527

Oxygen 20.949 23.143

Argon 0.93 1.282

Carbon dioxide 0.03 0.0456

Neon 1.8 × 10−3 1.25 × 10−3

Helium 5.24 × 10−4 7.24 × 10−3

Methane 1.4 × 10−4 7.75 × 10−5

Krypton 1.14 × 10−4 3.30 × 10−4

Nitrogen oxide 5 × 10−5 7.60 × 10−5

Xenon 8.6 × 10−6 3.90 ×10−5

Hydrogen 5 × 10−5 3.48 × 10−6

Table 4.6 The Size of Raindrops for DifferentTypes of Precipitation

Condition Raindrop Size ( m

Haze 0.01–3

Fog 0.01–100

Clouds 1–50

Light rain 3–800

Medium rain (4 mm/hr) 3–1,500

Heavy rain (16 mm/hr) 3–3,000

charged particles, which can interfere with radio electromagnetic waves near thesurface of the Earth. They have a strong influence on the ionosphere. When the sun-spot activity is at its maximum, the attenuation in the atmosphere is very high andcommunication is very difficult to establish. It is then sometimes necessary to switchto higher frequencies for long distance communications. Since the Sun rotatesaround its own axis, there is also a 27-day sunspot cycle that can also influence theionization density in the ionosphere. The sunspot activity is regularly observed bytelescopes from Earth and satellites.

The lower range of radio frequencies is more affected by sunspots than UHF fre-quencies or microwave communications.

4.3 Manmade Sources of Electromagnetic Interference

Manmade sources of electromagnetic interference can be intentional or uninten-tional. Intentional interference is used when one party wants to disrupt the commu-nication capability of the other party by transmitting an interfering signal in thesame frequency band with a higher power than what is used by the second party.Most manmade interference however is unintentional. It comes from bad planningof mobile telephony, bad reuse of frequencies, intermodulation products, other ser-vices using the same frequency bands, and industrial sources (which may not beused for communications at all, but still interfere with useful communications). Allcommunications that are not intended for a certain use are considered interference,regardless of the fact that it is a useful communication for some other users. If inter-ference is in the same communication channel, it will increase the noise in the sys-tem. It is important to know all the potential sources of interference in order tocalculate the parameters of the communication link.

4.3.1 Commercial Radio and Telephone Communications

Commercial radio and telephone communications are the most widely used com-munication systems. Radio and TV broadcasting are several decades old, and can befound in most undeveloped countries. Cell phone communications are also becomemore prominent in these locations. Pager networks and private communicationsmostly exist in developed countries and are not so widely used.

4.3.1.1 Broadcast Systems

Broadcast systems include radio, TV, satellite, and any other transmission of audioor video signals to a broad audience.

Radio BroadcastingHistorically radio broadcasting can be divided into amplitude modulation (AM)and frequency modulation (FM). AM is generally used for larger distances and isnot as good in quality as FM radio stations. There is also digital radio, which has thehighest quality signal.

4.3 Manmade Sources of Electromagnetic Interference 69

AM radio can be divided into long wave (LW), medium wave (MW), and shortwave (SW).

• LW works in the frequency band from 148.5 kHz to 283.5 kHz. The channelsare 9 kHz apart.

• MW works in the frequency band of 526.5–1,705 kHz. It is most frequentlyused for AM radio. There are 117 carrier frequencies in 10 kHz intervals (out-side of United States 9 kHz). Each carrier frequency should not deviate morethan ± 20 Hz from the allocated frequency. Modulation frequencies rangefrom 50 Hz to 5 kHz; if it exceeds 5 kHz, the radio frequency bandwidth willexceed 10 kHz and therefore interfere with the adjacent channel. The classesof AM stations based on transmitting power are given in Table 4.8. There areseveral thousand AM radio stations in the United States alone.

• SW uses frequencies above the ones of MW radio stations (i.e., from 2.3 MHzto 26.1 MHz). The channels are separated by only 5 kHz. They usually do notbroadcast 24 hours a day, and they sometimes change frequency during theday to compensate for the deterioration of reception conditions. The range isnot as large as with MW radio. SW is divided into frequency bands as given inTable 4.9.


Table 4.9 SW RadioBands

NameFrequency(MHz)

120m 2.3–2.495

90m 3.2–3.4

75m 3.9–4.0

60m 4.75–5.06

49m 5.9–6.2

41m 7.1–7.35

31m 9.4–9.9

25m 11.6–12.1

21m 13.57–13.87

19m 15.1–15.8

16m 17.48–17.9

13m 21.45–21.85

11m 25.6–26.1

Table 4.8 MW Radio Stations

Class Power (kW) Frequency (kHz)

A 10–50 535–1,605

B 0.25–50 1,605–1,705

C 0.25–1 1,230, 1,240, 1,340, 1,400, 1,450, 1,490

D 0.25-50 535–1,605, 1,605–1,705

Sunspots, weather conditions, and whether the station operates at day or nightwill influence the signal quality and propagation. The power of the transmitters canbe from 1W (or less) to 500 kW.

FM radio stations use the frequency band from 88 to 108 MHz. They have amuch higher quality than AM radio stations. Unlike AM radio stations, which havea very large range (several hundred km), FM stations normally can only be heard upto approximately 100 km from the stations. If there is clear frequency (no othertransmitter in the vicinity transmitting at the same or very near frequency), and ifthe transmitter is placed high on a mountain, this distance may be even larger.

The frequency band of 20 MHz is divided into 100 carrier channels with 200kHz in width, which are placed 200 kHz apart. The frequency deviation should notexceed ±75 kHz, while the stability of the carrier should be ±2 kHz. The maximumpowers of the transmitter classes are given in Table 4.10.

There are also many digital radio technologies in the world today, both terres-trial and satellite. This technology is still evolving and not a single one has gainedacceptance. Most users are still listening to radio stations with their old and verycheap receivers.

TV BroadcastingTV broadcasting can be either analog or digital. More and more countries in theworld are switching to digital systems. Old TV receivers might still be used with aDVB-T receiver. Analog channels are divided as shown in Tables 4.11 to 4.13.

TV standards are not the same in all countries. The National Television SystemsCommittee (NTSC) standard is used in the United States, Canada, Central America,most of South America, and Japan. NTSC has 525 horizontal lines. Phase Alterna-tion each Line (PAL) is used in Western Europe and China; it has 625 horizontal


Table 4.10 FMRadio Stations

Class Power (kW)

A 6

B 25

B1 50

C3 25

C2 50

C1 100

D 100

Table 4.11 VHF-1 TV Channels

Channel Frequency (MHz)

2 54–60

3 60–66

4 66–72

5 76–82

6 82–88

lines. Sequential Color ‘avec’ Memory (SECAM) is used in France, Russia, andsome eastern European countries. It also has 625 horizontal lines.

The maximum effective radiated power (ERP) for VHF-1 transmitters is 100kW, for VHF-3 it is 316 kW, and for UHF transmitters it is 5 MW.

While some analog TV systems cease to operate, new digital systems emerge.This trend is present around the world.

High definition television is of much higher quality than analog television. TheTV channels used for digital TV are the whole VHF-1 and VHF-3 bands, and a partof the UHF (14–36, 38–51) band.

Satellite broadcasting operates in the L-band (1,452–1,492 MHz), S-band(2,310–2,360, 2,520–2,655), Ku-band (11.7–12.7 GHz), K-band (17.3–17.8 GHz,21.4–22 GHz), and Ka-band (40.5–42.5 GHz). The frequency band from 11.7 to12.2 GHz is used for the fixed satellite service.


Table 4.12 VHF-3 TV Channels

Channel Frequency (MHz)

7 174–180

8 180–186

9 186–192

10 192–198

11 198–204

12 204–210

13 210–216

Table 4.13 UHF TV Channels

ChannelFrequency(MHz) Channel

Frequency(MHz) Channel

Frequency(MHz) Channel

Frequency(MHz)

14 470–476 32 578–584 50 686–692 68 794–800

15 476–482 33 584–590 51 692–698 69 800–806

16 482–488 34 590–596 52 698–704 70 806–812

17 488–494 35 596–602 53 704–710 71 812–818

18 494–500 36 602–608 54 710–716 72 818–824

19 500–506 37 608–614 55 716–722 73 824–830

20 506–512 38 614–620 56 722–728 74 830–836

21 512–518 39 620–626 57 728–734 75 836–842

22 518–524 40 626–632 58 734–740 76 842–848

23 524–530 41 632–638 59 740–746 77 848–854

24 530–536 42 638–644 60 746–752 78 854–860

25 536–542 43 644–650 61 752–758 79 860–866

26 542–548 44 650–656 62 758–764 80 866–872

27 548–554 45 656–662 63 764–770 81 872–878

28 554–560 46 662–668 64 770–776 82 878–884

29 560–566 47 668–674 65 776–782 83 884–890

30 566–572 48 674–680 66 782–788 — —

31 572–578 49 680–686 67 788–794 — —

The maximum power flux density on the Earth’s surface should not exceed−137 dBΩ/m2 for frequencies between 2.5 GHz and 27 GHz in order to not interferewith LOS terrestrial communications.

4.3.1.2 Cell Phone and Pager Networks

Cellular phones are part of everyday life in a manner so great that some people havemore than one mobile phone. The base stations are everywhere around us. Whilebroadcast transmitters are relatively rare, mobile base stations are much more pres-ent, whether they are placed on highways or inside offices.

Cell Phone NetworksThe Global System for Mobile Communication (GSM) is the most widely used cellphone network in the world. It operates in the 450 MHz band, 900 MHz band,1800 MHz band, 850 MHz band, and 1900 MHz band. The first three are mostcommonly used in Europe, Asia, and Africa, and the latter two in North and SouthAmerica. In Japan, CDMA technology is used. Table 4.14 gives some characteristicsfor various GSM networks.

The power of the transmitters is defined by international regulations as well aslocal country regulation. Up to 500W per channel (transmitter) of effective radiatedpower (ERP) is allowed in urban areas. In most cases, only 100W per channel isused. Usually there are 21 channels per sector, with three sectors totaling 63 chan-nels per base station. At the maximum, with a omnidirectional antenna, 96 channelsare possible. Thus, 48 kW would be the total maximum power if all the channelswere active, but that would be very rare. The power density drops very rapidly inaccordance with the distance from the antenna.

Pager NetworksThe pager system is a simplex communication system, which can send short mes-sages to a subscriber. This message can be numeric, alphanumeric, or a voice mes-sage. It is actually a warning or notice for further communications. The receivers areusually very simple and cheap, but the transmission system used for large distancesis not. Transmitters are usually high in power (kW) for wide-area paging systems,but for local systems (e.g., the office), which work at 2.4 GHz, the power is in mW.The paging frequencies are given in Table 4.15.


Table 4.14 GSM Networks

GSM 400 900 1,800 850 1,900

Uplinkfrequency (MHz)

450.4–457.6460.4–467.6

890–915 1,710–1,785 824–849 1,850–1,920

Downlinkfrequency (MHz)

478.8–486488.8–496

925(935)–960 1,805–1,880 869–894 1,930–1,990

Frequency spectrum 7 MHz 35 (25) MHz 75 MHz 25 MHz 70 MHz

Duplex separation 10 MHz 45 MHz 95 MHz 45 MHz 80 MHz

Carrier spacing 200 kHz 200 kHz 200 kHz 200 kHz 200 kHz

The maximum power of the transmitters for 152–153-MHz bands is 1.4 kW,for 153–159 MHz it is 150W, for 454–455 MHz it is 3.5 kW, and for 459–460MHz it is 150W.

4.3.1.3 Private Networks

Private networks differ from public networks as they are open only to users of thenetwork. Private networks are mostly used for corporate networks, which are notconnected to global Internet networks for security reasons.

Private networks can be established through an Internet connection or broad-band satellite. The Internet can also be wireless on 2.4 or 5.5 GHz (ISM frequencyband). There is a lot of interference from other services working in this license-freeband (Bluetooth, microwave oven, ZigBee).

The Spaceway satellite operating in the Ka-band enables broadband serviceswith up to a 16-Mbps connectivity rate.

4.3.2 Military Radio and Telephone Communications

There is a vast range of military transmitters in every army in the world, includingcommunication in HF and microwave ranges for both terrestrial and satellite appli-cations. The complete list of frequencies used by the military would be too large tofit into this book. They can be found in tables of frequency allocation for the specificfrequency band of interest. The civil transmitter frequencies in the vicinity of mili-tary bases (both land and sea) should be carefully planned in order not to interferewith the military communications. Furthermore, communication signals from air-crafts can arrive from great distances. Military communications are often coded andencrypted.

Portable radios have large power compared to cellular phones. They arerobustly made to endure severe conditions such as changes in heat, humidity, mud,dust, and so forth. Radios can also be mounted on vehicles. They are usually higherin power and range than portable versions carried by individuals.

As for propagation, the same laws apply for military as well as civiliancommunications.

4.3.3 Commercial Radar Systems

Radar is a system that uses electromagnetic impulses to identify the position of anobject. It also determines the altitude, direction, and speed. The object of interestcan be an airplane, naval vessel, clouds, or terrain. The word “radar” is an abbrevi-


Table 4.15 Paging Frequencies

Region Frequency (MHz)

United States 35–36, 43–44, 152–159, 454–460, 929, 931

European Union 47.0–47.25, 440–470

Japan 280

Australia 148

ation of radio detecting and ranging. Radar transmits pulses, which are reflectedfrom the objects. From the time of the reflected wave, the position can be calculated.It is used by the army, air traffic, in metrology and astronomy, and by the police.

4.3.3.1 Air Traffic Control

Air traffic control is placed on commercial (and also military) airports for prevent-ing collisions of airplanes when taking off or landing.

Radar operates on frequencies in the L-band (1–2 GHz) for long distances, inthe S-band (2–4 GHz) for medium distances, and the X-band (8–12 GHz) andKa-band (24–40 GHz) for short distances.

The power density from radars should not exceed 5 mW/cm2. Table 4.16 givesair traffic radar frequencies in the United States and European Union.

4.3.3.2 Astronomy

Radar in astronomy works on the same principle as radar for air traffic control.However, in this case the objects of interest are placed in the solar system. Radar isalso used for weather control. The frequencies are given in Table 4.17.

In metrology, radars in the S-band (2–4 GHz), K-band (18–24 GHz), andW-band (75–110 GHz) are used. The power density from radar should not exceed 5mW/cm2 for this application as well.

4.3.4 Industrial Sources

Dielectric heaters, neon signs, X-ray and welding machines, air conditioning, medi-cal devices, fluorescent lights, and lasers can interfere with communication systems.

Most international standards for communication equipment in industrial sur-roundings have a limit for susceptibility of either 3 V/m or 10 V/m. The standards ofinterest are EN 50082-2 (Electromagnetic Compatibility—Generic Immunity Stan-dard—Part 2: Industrial Environment), EN 61000-6-2 (Electromagnetic Compati-bility—Generic Standards—Part 6-2: Immunity for Industrial Environments), andEN 50082-1 (Electromagnetic Compatibility—Generic Immunity Standard, Part 1:Residential, Commercial, and Light Industry, CENELEC).


Table 4.16 Air Traffic Control Radar Frequencies


United States 1,300–1,350, 2,700–2,900, 3,500–3,650, 9,000–9,200, 13,250–13,400

European Union 1,215–1,350, 2,700–3,100, 3,300–3,500, 5,250–5,725

Table 4.17 Weather Radar Frequencies


United States 5,600–5,650, 9,300–9,500

European Union 5,250–5,570, 5,650–5,850

4.3.5 Intentional Interference

An example of intentional interference is jamming. The purpose of intentional inter-ference is to disable the communication of the adversary at a minimum cost. On theother hand, the goal of the party trying to communicate is to develop a systemimmune to interference. A total immunity from interference is impossible. It has tobe assumed that the party trying to jam the communication knows the frequencybut not the spreading codes. The shape of the signal should be chosen in such a waythat it leaves the jammer with no other option except the broadband Gauss noise.

There are several ways of possible interference. Figure 4.11 shows the spectralpower densities of interference against communication systems.

The width of the spread spectrum frequency band is B. If one party uses fre-quency hopping (S1 and S2) inside this band B, the interfering Gaussian noise spec-trum can be done in three ways. The first is the low spectral noise density in thewhole frequency band B [Figure 4.11(a)]. In this way, there will be interference inthe entire system—but this will not pose a great problem. The second method isincreased noise in one part of the spectrum [Figure 4.11(b)]. Here the damage willbe great in some cases, but in other cases there may be no damage at all. The lastmethod [Figure 4.11(c)] is to have large power in just one small portion of band B,but at the same time hop the position of this band. The damage is devastating if theinterference coincides with signal S2.

The party that wants to protect its communication system from jammingshould use frequency hopping or the time hopping spread spectrum and an antennasystem with high directivity.

This chapter is the last of the introduction chapters. The next chapters will dealwith active and passive interference control.


B

B

f

f

A

A

S1

S1

Bf

AS1

B

B

f

f

A

A

S2

S2

Bf

A S2

(a)

(b)

(c)

Figure 4.11 (a–c) Various types of interfering spectral densities.


Freeman, R.L., Radio System Design for Telecommunication, New York: Wiley-IEEE, 2006.Lin, G., and Alvarado, M., “Reviewing EU EMC Generic Standards,” EE-Evaluation Engineer-ing, July 2000, pp. 50–57.MIL-STD-464 Electromagnetic Environmental Effects, Requirements for Systems, 18 March1997, http://www.tscm.com/MIL-STD-464.pdf.Morrison, R., Noise and Other Interfering Signals, New York: John Wiley & Sons, 1992.Kocharyan, V., and D. Tolman, “An Express Diagnostic Method for ESD Simulators and Stan-dardized ESD Test Stations,” Proc. 2003 IEEE International Symposium on ElectromagneticCompatibility, Vol. 2, August 18–22, 2003, pp. 708–712.Rakov, V. A., “Transient Response of a Tall Object to Lightning,” IEEE Transactions on Electro-magnetic Compatibility, Vol. 43, No. 4, November 2001, pp. 654–661.Riaziat, M. L., Introduction to High-Speed Electronics and Optoelectronics, New York: JohnWiley & Sons, 1996.Van der Laan, P. C. T., and A. P. J. van Deursen, “Reliable Protection of Electronics AgainstLightning: Some Practical Applications,” IEEE Transactions on Electromagnetic Compatibility,Vol. 40, No. 4, November 1998, pp. 513–520.Young, P. H., Electronic Communication Techniques, Columbus, OH: Merrill Publishing Co.,1985.


C H A P T E R 5

Filter Interference Control

5.1 Filters

A filter is an electronic element with two ports that separates one frequency bandfrom another. The input signal, or excitation, goes through the filter to the outputport whose signal is called the response. The application of filters ranges fromacoustics to atomic clocks. In telecommunications, bandpass filters are used in theaudio frequency range for speech processing.

Filters can be divided in several ways. One division is into passive and active fil-ters. A passive filter does not require an external power source in order to operate,while an active filter does. A passive filter is made of inductors and capacitors ortheir equivalent (microwave waveguides). Active filters are made of resistors, capac-itors, and amplifiers.

Filters can be analog or digital, but analog filters are longer in use. They workwith analog or continuous signals, and their response signal is a continuous signal.Digital filters use analog-to-digital converters (ADC) to process the analog signal.The output from the filter or response is represented in digital numbers. In order toobtain the analog signal, a digital-to-analog converter (DAC) is needed.

Filters can be divided regarding the function they perform. There are lowpass,highpass, bandpass, and bandstop filters. A lowpass filter passes all frequencies upto the cutoff frequency and stops all of the frequencies above it. A highpass filterstops all frequencies up to the cutoff frequency and passes all of the frequenciesabove it. A bandpass filter passes all frequencies between two frequencies of interestand stops all of the frequencies below and above them. A bandstop filter stops thefrequencies between two frequencies of interest and passes all of the other frequen-cies above and below them.

The transfer function of the filter T(jω) [sometimes also called H(jω)] is a func-tion represented with gain (amplitude) and phase characteristics. In this chapter,only the amplitude will be dealt with. It shows how the filter changes the input sig-nal at the output response depending on the frequency. This response can be showngraphically or mathematically (the Bode plots).

The transfer function of the filters depends on the type used and the requiredapplication. It is the response of the filter depending on whether the filter is alowpass, highpass, bandpass, or bandstop and whether the type of filter isButterworth, Chebyshev, Bessel, or another type.

79

5.1.1 Lowpass Filter

The response of an ideal lowpass filter is shown in Figure 5.1. It must pass all of thefrequencies from 0 Hz up to the cutoff frequency fc without attenuation; at the sametime, it must stop (attenuate) all the frequencies above the cutoff frequency toamplitude 0.

Although the ideal response is desired in many situations, it is impossible to cre-ate it in the real world. The response of a real lowpass filter will be considered topass all the frequencies up to the point where the amplitude (magnitude) drops by

2 2/ or down to 0.707 (Figure 5.2)—in other words, by 3 dB.

The 3-dB point determines the width of the communication channel. Thismeans that the frequencies that are close to the cutoff frequency fc will still be passed(although somewhat attenuated), even though they are not supposed to be. Theywill however have a smaller amplitude. How much smaller, and at which frequen-cies this amplitude will reach 0, will depend on the type of the filter and its quality.The filters determine the quality and therefore the price of the electronic equipment.With good filters, the communication channels can be packed more closely to eachother without interference between them. The goal in designing the filters is to havea curve as steep as possible from the cutoff frequency up to the amplitude of 0 (i.e.,the filter should pass as little of the frequency band as possible above the cutofffrequency).

Looking closely at Figure 5.2, it can be seen that not only are the frequenciespassed above the cutoff frequency, but also the attenuation already starts below thecutoff frequency, which is certainly not intended. That is why, in designing thelowpass filter, it will be desirable to improve this setback or to have the attenuationstart at the cutoff frequency and not before.

5.1.2 Highpass Filter

A highpass filter response is shown in Figure 5.3. It should stop (attenuate) all thefrequencies from 0 Hz up to the cutoff frequency fc, and at the same time pass (atten-uate) all the frequencies above the cutoff frequency to infinity. Again, this is theideal response, which cannot be achieved.

80 Filter Interference Control

A

fc f0

1

Figure 5.1 Ideal lowpass filter response.

The response of a real highpass filter will pass all of the frequencies higher thanthe cutoff frequency up to the infinity. The cutoff frequency starts at the point wherethe amplitude (voltage) is at 0.707 (−3 dB) of the input signal (Figure 5.4). If thepower is used rather than voltage, then 3 dB is 50% less in power or 0.5.

Figure 5.4 shows that the responses of lowpass and highpass filters are actuallymirrored. This means that the highpass filter and the bandstop filter have similarresponses.

5.1.3 Bandpass Filter

The response of an ideal bandpass filter is shown in Figure 5.5. It should pass all ofthe frequencies from the first cutoff frequency fc1 to the second cutoff frequency fc2

without attenuation. At the same time, it should stop (attenuate) all of the frequen-cies below the first cutoff frequency and above the second cutoff frequency to theamplitude of 0.

The ideal bandpass filter will pass the frequency band between the two cutofffrequencies (0.707 amplitude) as shown in Figure 5.6. This has to be kept in mind

5.1 Filters 81

A

fc f0

1

0.707

Figure 5.2 Real lowpass filter response.

A

fc0

1

f

Figure 5.3 Ideal highpass filter response.

when planning the channel spacing in a real communication system—otherwise, thechannels might interfere with each other.


fc

0.707

f

A

1

0

Figure 5.4 Real highpass filter response.

fc1 fc2 f

A

1

0

Figure 5.5 Ideal bandpass filter response.

fc1 fc2 f

A

1

0

0.707

Figure 5.6 Real bandpass filter response.

5.1.4 Bandstop Filter

The response of an ideal bandstop filter is shown in Figure 5.7. It should stop all thefrequencies from the first cutoff frequency fc1 to the second cutoff frequency fc2. Atthe same time it is supposed to pass (without attenuation) all the frequencies belowthe first cutoff frequency and above the second cutoff frequency.

The real response of the bandpass filter will attenuate some of the frequenciesbelow the first cutoff frequency (0.707 amplitude) and also some of the frequenciesabove the second cutoff frequency (Figure 5.8). This filter is used when only oneband is to be attenuated and everything else is to be passed.

5.1.5 Resonator

A resonator is the most basic filter, intended to pass only one frequency or resonateon only one frequency and filter out (attenuate) all other frequencies. The resonatorresponse is shown in Figure 5.9.

The quality of the resonator or selectivity is defined with the Q-factor as

Qf

f

f

f fc c

c c

= =−∆ 2 1

(5.1)

5.1 Filters 83

fc1 fc2 f

A

1

0

0.707

Figure 5.7 Ideal bandstop filter response.

fc1 fc2 f

A

1

0

0.707

Figure 5.8 Real bandstop filter response.

where frequencies fc2 and fc1 are those where the voltage drops to 0.707 of the inci-dent value (power is at 50%). The more selective the resonator (filter), the smaller(narrower) the ∆f, and the larger the Q.

The simplest resonance can be done with the serial and parallel resonant circuit(Figure 5.10), which consists of resistor R, inductance L, and capacitor C.

The total impedance of serial resonance is

Z R jX jX R j L jC

R j LCL C= + − = + − = + +⎛

⎝⎜⎞⎠⎟

ωω

ωω

1 1(5.2)

where ω = 2πf. This means that the impedance will have both a real and imaginarypart. The latter can be either inductive or capacitive, depending on the frequencyand values of L and C. The resonance will appear when ImZ = 0, that is, when XL

= XC. Then, most of the energy will be transferred from the generator to the load,which can be an antenna in the transmitter system. To have this,

ωω

ωω

LC

LC

− = ⇒ =10

1(5.3)

must be valid. The above expression will be true for serial resonant frequency fsr


f

A

1

0

0.707

fc1 fc fc2

Figure 5.9 Resonator response.

R

I

U

C

L

Figure 5.10 Serial RLC resonance.

ωπ

sr srLC

fLC

= ⇒ =1 1

2(5.4)

In serial resonance, the voltages on L and C will be equal, but of oppositedirections.

The serial resonance is not the only possible resonance. There is also a parallelresonance of R, L, and C. It is shown in Figure 5.11.

The total admittance of the above circuit is

( )Y G j B BC L= + − (5.5)

where BC (1/XC) and BL (1/XL) are capacitive and inductive susceptances, and G(1/R) is the conductance. The imaginary part must again equal zero, or ImZ = 0.This will be fulfilled again for

ωπ

pr prLC

fLC

= ⇒ =1 1

2(5.6)

5.2 Analog Filters

There are many analog filter types; the following are used the most: Butterworth,Chebyshev, Bessel, and elliptic. Analog filters can be either passive or active. Passivefilters use resistors, inductors, and capacitors, while active analog filters use resis-tors, capacitors, and operational amplifiers as mentioned before. Active filters havehigher Q-factors, whereas passive filters with inductors do not.

5.2.1 Butterworth Filter

Butterworth filters have flat attenuation in the passband region without any ripple.At the cutoff frequency, fc, the attenuation is 3 dB (50% power). The frequencyresponse of an Nth-order Butterworth lowpass filter is obtained by the transferfunction T(jω) as

5.2 Analog Filters 85

U

I ICILIR

ILC

L CR

Figure 5.11 Parallel RLC resonance.

( )T jf

f c

Nω =

+⎛⎝⎜

⎞⎠⎟

1

12

(5.7)

where fc is the cutoff frequency and N is the filter order. The response is flat both at0 and infinity. The N number is related to the total number of reactive elements(inductors or capacitors) in the lowpass or highpass filter. For a bandpass orbandstop filter, the number of required reactive elements is twice as high as forlowpass or highpass filters. Therefore, the filter order determines the steepness ofthe slope in the bandstop part. The higher the filter order, the steeper the slope. Theslope for the lowpass Butterworth filter is −20N dB/decade (or approximately 6NdB/octave) as shown in Figure 5.12.

The frequency response is usually shown in logarithmic scale, where the ratio of10 to 1 is called a decade (the ratio of 2 to 1 is called an octave).

That is why (5.7) is used for drawing Figure 5.12 in decibels according to

( ) ( )T j T jω ω= 20 10log (5.8)

5.2.2 Chebyshev Filters

Chebyshev filters, in comparison to Butterworth filters, have a ripple in thepassband region. They are more similar to the ideal filter except for the ripple.Above the cutoff frequency, Chebyshev filters have much higher attenuation thanButterworth filters. The transfer function is calculated from

( )T j

Tf

fNc

ω

ε

=+

⎛⎝⎜

⎞⎠⎟

1

1 2 2

(5.9)

where ε is a real constant whose value is less than 1; it determines the ripple of thefilter calculated from


−30

−25

−20

−15

−10

−5

0

T,dB

0 1. 0.4 0 7. 1 0. 1 3. 1 6. 1 9. 2 2. 2 5. 2 8.

n=3

n=2

n=1

f

Figure 5.12 Butterworth filter response function for different orders.

( )ε = −10 10 1 0 5. .r (5.10)

with r being the positive real number and TN(f/fc) the Nth order Chebyshev polyno-mial calculated from

Tf

f

Nf

f

f

f

NN

c

c c⎛⎝⎜

⎞⎠⎟ =

⎛⎝⎜

⎞⎠⎟

⎛

⎝⎜

⎞

⎠⎟ ≤−cos cos ,

cosh cos

1 1

h − ⎛⎝⎜

⎞⎠⎟

⎛

⎝⎜

⎞

⎠⎟ ≥

⎧

⎨⎪⎪

⎩⎪⎪

1 1f

f

f

fc c

(5.11)

The Chebyshev filter has three parameters, ε, fc, and N. Figure 5.13 shows theChebyshev filter response for r = 1, and a different order N in the logarithmic scale.

A ripple is visible below the cutoff frequency. For different values of r, differentslopes can be achieved.

5.2.3 Bessel Filters

Bessel filters are used for reducing nonlinear phase distortion (i.e., they have a flatdelay). The transition from passband to stopband is much slower than with otherfilters. It is the only one of the filters mentioned in this chapter where the phaseresponse is important. The transfer functions for N = 1, 2, and 3 are given as

( )

( )

( )

T jf

N

T jf f

N

T jf f f

ω

ω

ω

=+

=

=+ +

=

=+ + +

1

11

3

3 92

15

6 45 25

2

4 2

6 4 2

,

,

53, N =

(5.12)


−10

−9

−8

−7

−6

−5

−4

−3

−2

−1

0

1

0.1 0.4 0 7. 1 0. 1 3. 1 6. 1 9. 2 2. 2 5. 2 8.

n=3

n=2

n=1

f

T,dB

Figure 5.13 Chebyshev filter response for a different order and for r = 1.

where N is the order of the polynomial. The response functions (amplitude) areshown in Figure 5.14.

5.2.4 Elliptic Filters

Elliptic filters have ripples in both passbands and stopbands. They also have a veryfast transition between passbands and stopbands. They are also called Cauer filters,and are used in communications where multiple carriers exist. The transfer functionis

( )T j

Ff

fNc

ω

ε

=+

⎛⎝⎜

⎞⎠⎟

1

1 2 2

, (5.13)

where ε is a ripple factor and FN is the Jacobian elliptic function. The calculation ofthis filter type is not easy.

There are other types of filters such as Gaussian, Legendre, and Linkowitz-Rileyfilters, but the most frequently used ones are described above.

The filters mentioned above can be either passive or active. Passive filters use R,L, and C components, while active filters use operational amplifiers instead of the Lcomponent.

5.2.5 Passive Filters

Passive filters do not require external power. For low- or high-pass filters, RL or RCcombinations can be used. For bandpass filters RLC combinations are used.


−10

−9

−8

−7

−6

−5

−4

−3

−2

−1

0

1

0.1 0.4 0.7 1.0 1.3 1.6 1.9 2.2 2.5 2.8

n=3

n=2

n=1

f

T,dB

Figure 5.14 Bessel filter response for a different order of N.

5.2.5.1 Lowpass RL Filter

The lowpass RL filter is shown in Figure 5.15. It consists of a resistor and imped-ance in an L shape. This is a first-order (N = 1) filter, because it has only one reactiveelement.

The transfer function of this filter is given as

( )T jf

f c

ω =

+⎛⎝⎜

⎞⎠⎟

1

12

, (5.14)

where the cutoff frequency fc is obtained from

fR

LRLc = =

2π ω(5.15)

5.2.5.2 Lowpass RC Filter

The lowpass RC filter is shown in Figure 5.16. It consists of resistor and conductor,also in an L shape. It is a first-order (N = 1) filter as well.


( )T jf

f c

ω =

+⎛⎝⎜

⎞⎠⎟

1

12

, (5.16)


R

L

Figure 5.15 Lowpass RL filter.

R

C

Figure 5.16 Lowpass RC filter.


fRC RCc = =1

21

π ω(5.17)

5.2.5.3 Highpass RL Filter

The highpass RL filter is shown in Figure 5.17. It consists of a resistor and imped-ance in an L shape. It differs from the lowpass RL filter in the position of the ele-ments—R and L exchanged places.


( )T jf

fc

ω =

+⎛⎝⎜

⎞⎠⎟

1

12

(5.18)


fR

LRLc = =

2π ω(5.19)

5.2.5.4 Highpass RC Filter

The highpass RC filter is shown in Figure 5.18. It consists of a resistor and conduc-tor with a changed position or R and C compared to the lowpass RC filter.


R

L

Figure 5.17 Highpass RL filter.

C

R

Figure 5.18 Highpass RC filter.


( )T jf

fc

ω =

+⎛⎝⎜

⎞⎠⎟

1

12

(5.20)


fRC RCc = =1

21

π ω(5.21)

It can be seen that the cutoff frequencies for the RC lowpass and RC highpassare the same. The same applies for RL highpass and RL lowpass filters as well.

Bandpass or bandstop filters require two sets of L or C components (each for itscutoff frequency).

5.2.6 Active Filters

Instead of inductors, active filters use both operational amplifiers and R and C com-ponents. Their transfer function can approach ideal filters more closely than passivefilters, and there can be an amplification of the signal, which is compared only to theattenuation in passive filters.

Figure 5.19 shows the lowpass active filter of the first order.The cutoff frequency is given with

fR C R Cc = =1

21

1 1π ω(5.22)

The gain in the passband is equal to −R1/R2, and in the stopband part it drops by20 dB/decade.

Figure 5.20 shows the highpass active filter of the first order.The cutoff frequency is given with

fR C R Cc = =1

22

2 2π ω(5.23)

The gain in the passband is also equal to –R1/R2.

More complex filters can be achieved with higher orders (N > 1) (i.e., with moreC elements).

5.3 Digital Filters

There is usually an analog-to-digital converter (ADC), a microprocessor acting asdigital filter, and a digital-to-analog converter (DAC) in digital filters, as shown inFigure 5.21. They are used in modern communication systems.

5.3 Digital Filters 91

There are two main types of digital filters: finite impulse response (FIR) andinfinitive impulse response (IIR). FIR filters are sometimes called nonrecursive fil-ters and IIR filters are known as recursive filters.

The advantage of digital filters is that they can be programmed and stored in thememory of the processor. They can also be reprogrammed without the change ofhardware, whereas with analog filters this is not possible. Digital filters are alsomore stable than analog filters regarding temperature and time.


C

R1

R2

Figure 5.19 First-order lowpass active filter.

RC

2

R1

Figure 5.20 First-order highpass active filter.

ADC Digital filter DACx t( ) y t( )xn yn

Figure 5.21 Digital filters.

5.3.1 FIR Filters

The impulse response of FIR filters has a finite length. The response lasts N + 1 sam-ples for the Nth filter order and then becomes zero.

If a time dependable analog signal is used at the input of a filter x(t), it must beconverted into a digital signal. This is done by discretization or taking the samplesin time intervals of ∆t. The sampled value of x at discretization time ti = i∆t will be

( )x x ti i= (5.24)

The digital values from the analog to digital converter (ADC) will have thesequence x0, x1, x2, x3, ..., xn, where x0 is the sampled value at t = 0, x1 the sampledvalue at ∆t, x2 is the sampled value at 2 · ∆t, xn is the sampled value at n · ∆t and soforth.

The digital output from the filter will have the sequence of values y0, y1, y2, y3,..., yn. The exact values of y will depend on the values of x and the function of thedigital filter. The values of y are then fed to the digital-to-analog converter (DAC) toobtain analog values again.

The delay filter, yn = xn−1, can be realized by taking the output value at the timeof (n − 1) · ∆t, or

y x

y x

y x

y x

y xn n

0 1

1 0

2 1

3 2

1

=====

−

−

(5.25)

If the same filter would take the output values at intervals n · ∆t, it would just bean all-pass filter.

Usually, all values of x before t = 0 are considered to be zero. The filter sums thecurrent value xn and the previous value xn−1:

y x x

y x x

y x x

y x x

0 0 1

1 1 0

2 2 1

3 3 2

= += += += +

−

(5.26)

The simple lowpass filter can be made by calculating the arithmetic mean of thecurrent and the previous value, that is, yn = (xn + xn−1)/2, or

( )( )( )( )

y x x

y x x

y x x

y x x

0 0 1

1 1 0

2 2 1

3 3 2

2

2

2

2

= +

= +

= +

= +

−

(5.27)


The order of the digital FIR filter is the number of previous inputs needed for thecalculation of the current filter output. The filter mentioned above is of the firstorder, since only one previous input was used. Thus, depending on the order, theFIR digital filter can be written as

y a x

y a x a x

y a x a x

n n

n n n

n n

== += +

−

0

0 1 1

0 1

0 order

first order

n na x− −+1 2 2 second order

(5.28)

The transfer function of the filter describes the filter function, which depends oncurrent and previous values of input for FIR filters. For this purpose, the delay func-tion, z−1, must be introduced. It gives the previous value of the sequence or delay ofthe same.

If the delay function is applied to the input value xn, the output will be the previ-ous value, that is, xn−1, or

z x xn n−

−=11 (5.29)

If the input sequence is for example equal to x0 = 5, x1 = 3, x2 = 6, x3 = 2, thenz−1x0 = 0, z−1x1 = 5, z−1x2 = 3, z−1x3 = 6, and so forth. It is assumed that x−1 = 0.

The delay does not have to be restricted to the previous input only, so the fol-lowing applies:

( )z z x z x xn n n− − −

− −= =1 1 11 2 (5.30)

or

z z z− − −=1 1 2 (5.31)

giving

z x xn n−

−=22 (5.32)

If needed, more delay can be used.The transfer function of FIR filters is as follows:

( )y a a z a z xn n= + +− −0 1

12

2 (5.33)

The general diagram of a FIR digital filter with delay functions is shown inFigure 5.22.

5.3.2 IIR Filters

The impulse response has an infinite length of numbers. Usually it is best to designan analog filter (i.e., Butterworth, Chebyshev, or Bessel) and then convert it into adigital filter.


While for FIR or nonrecursive filters, the current output (yn) depends only onthe current input current (xn) and/or previous (xn−1, xn−2, ...) inputs, the IIR or recur-sive filter’s current output also depends on previous outputs (yn−1, yn−2, ...).

In the recursive digital filter the current output depends on the current input andthe previous output

y x yn n n= − −1 (5.34)

or

y x y

y x y

y x y

0 0 1

1 1 0

2 2 1

= −= −= −

−

(5.35)

where y−1 is usually taken to be 0. If the values of y−1 are used in the followingexpressions, the above becomes

y x y x

y x y x x

y x y x x x

0 0 1 0

1 1 0 1 0

2 2 1 2 1 0

= − == − = −= − = − −

−

(5.36)

It can be seen that the current output yn is equal to the difference of the currentinput and all previous inputs.


x n( ) y n( )

z−1

z−1

z−1

a2

a1

a0

Figure 5.22 FIR digital filter.

For example, in the fifth time interval, the IIR or recursive filter will have theexpression

y x y5 5 4= − (5.37)

In order to have the same function performed with the nonrecursive FIR filter,the function should be:

y x x x x x x5 5 4 3 2 1 0= − − − − − (5.38)

This means that the nonrecursive filter would require much more time, opera-tion, and memory to operate as the recursive filter.

The order of the recursive IIR digital filter is the largest number of previousinput or output values needed to calculate the current output. The lowest order ofthe IIR filter is the first order; if it were of the zero order, it would not be a recursivefilter!

The IIR digital filter functions depending on the order can be written as

b y b y a x a x

b y b y b yn n n n

n n n

0 1 1 0 1 1

0 1 1 2 2

+ = ++ +

− −

− −

1st order

= + +− −a x a x a xn n n0 1 1 2 2 2nd order(5.39)

The transfer function of IIR filters is similar to those of FIR filters. The samedelay function is valid for output values as well:

z y yn n−

−=11 (5.40)

If the second order filter is used, then we will have

z y y

z y y

z x x

z x x

n n

n n

n n

n n

−−

−−

−−

−−

=

=

=

=

11

22

11

22

(5.41)

and substituting the above expression in the second-order function will yield

( ) ( )y b b z b z x a a z a zn n0 11

22

0 11

22+ + = + +− − − − (5.42)

The above expression can be then written as the transfer function of the second-order IIR filter, showing the dependence of the current output on the current inputand delay coefficients:

( )( )y

a a z a z

b b z b zxn n=

+ +

+ +

− −

− −

0 11

22

0 11

22

(5.43)

A general diagram of an IIR digital filter with delay functions is shown in Figure5.23.


5.4 Microwave Filters

A microwave filter is a two-port network used for controlling the frequencyresponse in a microwave system by passing the frequencies in a passband and atten-uating the frequencies in the stop band. The types are the same as in other filtertypes: lowpass, highpass, bandpass, and bandstop.

The electric circuits are similar to those of analog filters. The only difference isthat the L and C elements are realized differently at high microwave frequenciesthan at lower frequencies. Microwave filters can be realized as lumped element,waveguide cavity, and dielectric.

5.4.1 Lumped-Element Filters

The inductor and capacitor at microwave frequencies (gigahertz) can be replacedwith transmission lines; the inductor can be made with a short-circuited stub,whereas the capacitor can be made with an open stub as shown in Figure 5.24. Thisis done with the Richard’s transformation.

Inductive reactance is given as

jX j L jL lL = =ω βtan (5.44)

where βl = ωl/vp = 2π. The capacitive susceptance is given as

jB j C jC lC = =ω βtan (5.45)

For the cutoff frequency, it must be tan βl = 1 or βl = π/4. With β = 2π/λ, it fol-lows that the stub length l should be l = λ/8, where λ is the wavelength of the line atthe cutoff frequency ωc = 2πfc.

5.4 Microwave Filters 97

x n( ) y n( )

z−1

z−1

z−1

a2

a1

a0

z−1

z−1

z−1

b2

b1

Figure 5.23 IIR digital filter.

In designing the microwave lumped filter, the first task is to convert the elec-tronic circuits with L and C elements (i.e., Chebyshev or Butterworth) into an opencircuit or short circuit using Richard’s transformation. The exact normalized proto-type element values for each filter type can be calculated or found in tables. Next,the serial short circuit stubs must be converted into parallel open stubs because onlythey can be realized in microstrip lines. This is done with the Kuroda transforma-tion. Before that, a unit element must be added to both ends of the filter (i.e., 50Ω, ifthis is the characteristic impedance of the generator). Unit elements do not influencethe filter since they are matched to the generator and load. The Kuroda transforma-tion is shown in Figure 5.25.

The transformation is achieved using

nZ

Z2 2

1

1= + (5.46)

In the end, the filter is realized in a microstrip line as shown in Figure 5.26. Itcan be done on a printed circuit board (PCB).

The order of the filter is equal to the number of stubs.

5.4.2 Waveguide Cavity Filters

The cavity notch filter was mentioned in Section 1.5. It is often used for FM radiostations.

Waveguide filters are made from resonators coupled together by parallel induc-tive irises, which make the resonators. Their circular openings have an inductivecharacter. They are shown in Figure 5.27.

The irises are spaced along the waveguide at λg/2, where λg is the guide wave-length of each resonator. The equivalent circuit is shown in Figure 5.28.


λ ω/8 at c

S.C.

λ ω/8 at c

O.C.

jXLZ L=0

Z /C= 10jBC

jXL

jBC

L

C

Figure 5.24 Richard’s transformation of inductor and capacitor.

This type of filter can be used for the realization of bandpass or bandstop filterswith a high Q factor. The size of the openings will determine the coupling strength,which will influence the bandwidth as well as the shape of the filter transfer function(Chebyshev or close to ideal filter). The openings can also be rectangular, or theinductive element can be created with a vertical screw (column) inserted inside the


Unit element Unit element

Z2

Z1 Z n/22

Z n/12

l

l l

l

Series stub

S.C.

O.C.

Shunt stub

Figure 5.25 Kuroda transformation.

Z01

Zs1

Z02Z0 Z0

Zs2 Zs3

Figure 5.26 Third-order microstrip lowpass filter.

l 1 l 2 l 3

a

b

Figure 5.27 Waveguide cavity filter.

waveguide. If it is inserted a little, it will behave as a capacitor (the electric field isstronger at smaller distances from the top to the bottom of the waveguide), andwhen it connects the top and bottom, it will behave as an inductor (the electric fieldis equal to zero and hence the magnetic field is at a maximum, which is a character-istic of inductors). This is shown in Figure 5.29.

When the screw is inserted only a little inside the waveguide, the electric fieldgets stronger and thus behaves like a capacitor [Figure 5.29(a)]. When the top andbottom of the waveguide are connected, the inserted screw behaves like an inductor,because the electric field is zero at this point [Figure 5.29(b)]. In this way, inserting ascrew in or out can change the resonant frequency in some small bandwidth. Byadding more resonators (screws) in a series or parallel, a different shape andbandpass or bandstop can be achieved.

5.4.3 Dielectric Resonator

If instead of air a material with higher permittivity (εr 20) is used, the dimension ofthe resonator can be much smaller—up to two orders of magnitude. Dielectric reso-nators usually come in the shape of a small disc or cube (Figure 5.30).

The electromagnetic field is concentrated inside the dielectric material, so thedimensions of the dielectric resonator are much smaller than the waveguide cavityresonator for the same frequency resonance (mode). Due to small tangent loss, the


l1 l2 l3

jX1 jX2 jX3 jX4Z0 Z0 Z0 Z0

Figure 5.28 Equivalent circuit of waveguide resonator.

b

a

“C” b

a

“L”

Figure 5.29 Capacitor and inductor in a waveguide.

z

a

x

y

d

ε >>1r

Figure 5.30 Dielectric resonator.

quality factor can be more than 10,000. Such resonators are used in the frequencyrange from 300 MHz to 300 GHz.

As with the waveguide cavity, the resonant modes are the same as in circularcavity waveguides (TE and TM modes) as shown in Tables 5.1 and 5.2.

The resonant frequency of the desired mode of propagation (TEmn or TMmn) ofthe dielectric resonator can be calculated from

fc k

ac

r

=⋅

⋅0

2π ε(5.47)

where c is the speed of light, the values of k0 are taken from Tables 5.1 and 5.2, anda is the radius of the dielectric.


Caviacchi, T. J., Digital Signal Processing, New York: John Wiley & Sons, 2000.Douglas, S. C., “Adaptive Filtering,” in Digital Signal Processing, V. K. Madisetti and D. B. Wil-liams, (eds.), Boca Raton, FL: CRC Press, 1999.Dunlop, J., and D. G. Smith, Telecommunication Engineering, 3rd ed., London, U.K.: Chapmanand Hall, 1994.Di Paolo, F., Network and Devices Using Planar Transmission Devices, Boca Raton, FL: CRCPress, 2000.Haykin, S., “Adaptive Systems for Signal Process,” in Advanced Signal Processing Handbook, S.Stergiopoulos, (ed.), Boca Raton, FL: CRC Press, 2001.“Introduction to Digital Filters,” http://www.dsptutor.freeuk.com/dfilt1.htm.Massara, R.E., et al., “Active Filters,” The Electrical Engineering Handbook, R. C. Dorf (ed.),Boca Raton, FL: CRC Press, 2000.Paul, H., and P. E. Young, Electronic Communication Techniques, New York: Merril PublishingCo., 1985.


Table 5.1 Values of (k0) for TEmn Modes

n m

0 1 2 3 4 5 6 7

1 3.832 1.841 3.054 4.201 5.317 6.416 7.501 8.578

2 7.016 5.331 6.706 8.015 9.282 10.520 11.735 12.932

3 10.173 8.536 9.969 11.346 12.682 13.987 — —

4 13.324 11.706 13.170 — — — — —

Table 5.2 Values of (k0) for TMmn Modes

n m

0 1 2 3 4 5 6 7

1 2.405 3.832 5.136 6.380 7.588 8.771 9.936 11.086

2 5.520 7.016 8.417 9.761 11.065 12.339 13.589 14.821

3 8.654 10.173 11.620 13.015 14.372 — — —

4 11.792 13.323 14.796 — — — — —

Pozar, D. M., Microwave Engineering, 2nd ed., New York: John Wiley & Sons, 1998.Rahman, J., et al., “Filters,” in Measurement, Instrumentation and Sensors Handbook, J. G.Webster, (ed.), Boca Raton, FL: CRC Press, 1999.Rosa, A. J., “Filters (Passive),”in Engineering Handbook, R. C. Dorf, (ed.), Boca Raton, FL: CRCPress, 2000.Whitaker, J. C., “Circuit Fundamentals,” in The Resource Handbook of Electronics, J. C.Whitaker, (ed.), Boca Raton, FL: CRC Press, 2001.


C H A P T E R 6

Modulation Techniques

6.1 Signal Processing and Detection

At the receiving end of a digital communication system, the decision whether the bitvalue is 0 or 1 must be made. This decision takes place after the demodulation pro-cess and sampling of the waveform. This should be done with as little errors as pos-sible. The errors may come from filtering and noise in the communication signal.The most common noise in the radio channel is the Gaussian or white noise, whichis present in every communication signal with the same spectral density from verylow frequencies up to 1012 Hz.

In a binary channel, the transmitted signal si(t) in the time interval (0, T) willhave the following form:

( ) ( )( )s t

s t t T

s t t Ti =≤ ≤≤ ≤

⎧⎨⎩

1

2

0 1

0 0

""

" "(6.1)

The received signal, r(t), will be degraded by the noise, n(t), and possibly by thepulse response, hc(t), as

( ) ( ) ( ) ( )r t s t h t n t i Mi c= ∗ + = 1 2, , , (6.2)

where n(t) is the mean white noise and * is the convolution operator. If the channelis ideal (i.e., there is no binary transmission distortion), hc(t) will not introduce thedegradation, and the receiving signal r(t) can be written as

( ) ( ) ( )r t s t n t i t Ti= + = ≤ ≤1 2 0, (6.3)

Figure 6.1 shows the demodulation and detection process at the receiver end.Demodulation determines the waveform, whereas detection is the procedure ofdetermining the meaning of the waveform (i.e., decision making).

The frequency downconverter translates the frequency to a lower frequency;then the receiving filter extracts the wanted frequencies and prepares the signal fordetection. The filtering in the channel usually leads to intersymbol interference (ISI).This is why the equalizing filter is placed after the receiving filter. The sampling ofthe waveform is done before the actual detection. The pulse in the baseband isdescribed as

( ) ( ) ( )z t a t n ti= + (6.4)

103

At the moment when t = T, the sample z(T) is taken. Its voltage is directlydependent on the energy of the received symbol and inversely proportional to thenoise. If the input noise is Gaussian and the receiving filter is linear, the sample willbe

( ) ( ) ( )z T a T n T ii= + =, , ,1 2 (6.5)

where ai(T) is the wanted part of the signal, and n(T) is the mean value of Gaussiannoise.

The next step will be detecting or decision making, depending on the digitalmeaning of the sample. The value of the random Gaussian noise n can be written as

( )p nn= − ⎛

⎝⎜⎞⎠⎟

⎡

⎣⎢

⎤

⎦⎥

1

2

12

2

σ π σexp (6.6)

where σ2 is the noise change. The two above expressions combined give the possibil-ities of waveforms for s1 and s2 as

( )p z sz a

11

0

21

2

12

= −−⎛

⎝⎜

⎞⎠⎟

⎡

⎣⎢⎢

⎤

⎦⎥⎥σ π σ

exp (6.7)

( )p z sz a

22

0

21

2

12

= −−⎛

⎝⎜

⎞⎠⎟

⎡

⎣⎢⎢

⎤

⎦⎥⎥σ π σ

exp (6.8)

The probability functions for s1 and s2 are shown in Figure 6.2.

104 Modulation Techniques

White noise

Frequencydown-converter Filter Equalization

filter

Bordercomparison

><γ

DemodulationDetection

Sampling

Conversion of waveform into sample Decision making

s t( )i r t( ) z T( )

z t( )

Figure 6.1 Demodulation and detection of digital signals.

p z s( / )1 p z s( / )2

V2

V1

γα1 α1z T( )

Figure 6.2 Probability functions for p(z/s1) and p(z/s2).

The curves represent the probability functions of z(T) for symbols s1 and s2. Theabcissa axis z(T) represents all the possible sampled values.

After the waveform is converted into the sample, the true shape of the waveformis no longer important. All of the waveforms that were converted into the samevalue of z(T) are equal as far as detection is concerned. Therefore, it is not the shapebut the received energy that is the key parameter influencing the detection. Since thesignal level z(T) depends on the energy of the bit received, the higher the value ofz(T), the less errors in decision making.

6.2 Modulation and Demodulation

Modulation is a process of changing the electrical signal, which carries the informa-tion for its transmission. It changes one or more parameters of an auxiliary signal,depending on the signal that carries the information. This auxiliary signal is calledthe transmission signal or carrier. The signal that carries the information is calledthe modulating signal. It controls the changes of the transmission signal. The resultof the modulation process is the modulated signal. Modulation is performed in anelectronic device called the modulator, which is located in the transmitter. Thereverse process, or demodulation, is the transformation of the received signal intothe starting shape, which takes place in the demodulator at the receiving side.

The modulation can be either analog or digital. Table 6.1 gives the most usedbasic types of modulation in communication systems.

Analog modulations include amplitude modulation (AM), frequency modula-tion (FM), and phase modulation. Digital modulation includes amplitude shift key-ing (ASK), frequency shift keying (FSK), phase shift keying (PSK), and pulse codemodulation (PCM). Quadrature amplitude modulation (QAM) can be both analogand digital. From the above mentioned basic types of modulation, several othercombinations have been developed. They will be mentioned in Sections 6.2.1.1through 6.2.1.3 and 6.2.2.1 through 6.2.2.5.

6.2.1 Analog Modulations

In analog modulations, the message or information in analog form is superimposedon a carrier, which usually has a sinusoidal form. The three quantities that can bechanged regarding the modulating signal are amplitude, frequency, and phase.

6.2 Modulation and Demodulation 105

Table 6.1 Modulations

Analog Digital

Amplitude modulation (AM) Amplitude shift keying (ASK)

Frequency modulation (FM) Frequency shift keying (FSK)

Phase modulation (PM) Phase shift keying (PSK)

Pulse code modulation (PCM)

Quadrature amplitude modulation (QAM)

6.2.1.1 Amplitude Modulation (AM)

Amplitude modulation is the oldest modulation method. It works by changing thestrength of the transmitted signal depending on the information being sent.

The function of the amplitude of the AM modulated signal is linear and dependson the modulating signal. It can be written as

( ) ( )[ ] ( )u t U u t tAM cm m c= + +cos ω ϕ (6.9)

where Ucm is the amplitude of the nonmodulated carrier, um is the modulating sig-nal, ωc is the frequency of the sinusoidal carrier signal, and ϕ is the phase of the car-rier signal. The principle of AM modulation is shown in Figure 6.3.

Since the phase, ϕ, of the carrier does not influence the modulation process, itwill be assumed that ϕ = 0 in further analysis.

If the modulation signal is also sinusoidal it can be written as

( )u t U tm m m= cos ω (6.10)

The modulated signal can then be written as


Ucm

uAM

0 t

0 t

um

0

Ucm

uc

t

Figure 6.3 Waveforms of carrier, modulating, and modulated signals of AM modulation.

( ) [ ]u t U U t tAM cm m m c= + cos cosω ω (6.11)

or as

( )u t UU

Ut tAM cm

m

cmm c= +

⎡

⎣⎢

⎤

⎦⎥1 cos cosω ω (6.12)

The amplitude of the modulated signal changes around the mean value of Ucm.The maximum amplitude of the modulated signal is Ucm + Um. The minimum ampli-tude is Ucm − Um, accordingly. The ratio of the modulating signal and nonmodulatedcarrier signal is called the modulation index ma.

mU

Uam

cm

= (6.13)

The modulation index is sometimes also called modulation depth. The expres-sion for the modulated signal can then be written as

( ) [ ]u t U m t tAM cm a m c= +1 cos cosω ω (6.14)

The above expression using the cosine product yields

( ) ( ) ( )u t U tm

tm

tAM cm ma

c ma

c m= + + + −⎡⎣⎢

⎤⎦⎥

cos cos cosω ω ω ω ω2 2

(6.15)

The modulation index should be ma ≤ 1. For ma > 1, correct demodulation is notpossible. To determine the modulation index, a modulation trapezoid can be used.The modulation signal is on the horizontal axis and the modulated signal on the ver-tical axis, as shown in Figure 6.4.

The modulation index is calculated as

mA BA Ba = −

+(6.16)

Demodulation of AM signals can be done with envelope detection or with syn-chronous detection.


uAM uAM uAM

0 0 0um um um

ma <1 ma >1ma =1

BA

Figure 6.4 Modulation index depending on the trapezoid.

Envelope detection is the simplest procedure of AM signal demodulation. Theenvelope amplitude of the modulated signal with modulation index ma ≤ 1 is pro-portional to the modulation signal. Normally, a peak-amplitude detector or recti-fier is used. Figure 6.5 shows the AM demodulation procedure.

The diode conducts when the input voltage is higher than the diode cut-in volt-age, which can range from 0.2V to 0.7V. The capacitor is used for filtering thedemodulated signal. It also increases the efficiency of the demodulator by increasingthe peak value of the carrier pulses while the diode is conducting. When the diode isnot conducting, the capacitor is holding its charge. Additionally, an amplifier mightbe added to the demodulator.

Demodulation with synchronous or coherent detection requires an additionalsignal whose frequency and phase match the carrier frequency. They also must be inphase, thus the name coherent. This type of detection is shown in Figure 6.6.

The additional signal ua(t) (whose frequency matches the carrier frequency) willhave the following form:

( )u t U ta a c= cos ω (6.17)

Mixing this additional signal with the modulated signal gives

( ) ( )

( ) ( )

u t u t k

U tm

tm

t

AM a AM

cm ca

c ma

c m

⋅ =

⋅ + + + −⎡⎣cos cos cosω ω ω ω ω

2 2⎢⎤⎦⎥

⋅cos ωc t

(6.18)

Further, it follows that


AMsignal

R C

Demodulatedsignal

Diode

Figure 6.5 AM demodulation with envelope detection.

u t( )AM [ ]u t u t( )x ( )AM a [ ( )x ( )]u t u tAM a LPF

u t( )a

LPF

Figure 6.6 Synchronous detection.

( ) ( ) ( ) ( )u t u t k U

m t

tm t

AM a AM cm

a m

ca c m⋅ = ⋅

+ +

++1

2

1

22

2

cos

coscos

ω

ωω ω

( )+

−

⎡

⎣⎢

⎤

⎦⎥

⎧

⎨⎪

⎩⎪

⎫

⎬⎪

⎭⎪cos 2ω ωc m t

(6.19)

With the use of a lowpass filter, the components around 2ωc are filtered out:

( ) ( )[ ] ( )u t u t k U m tAM a LPF AM cm a m⋅ = ⋅ +12

1 cos ω (6.20)

6.2.1.2 Frequency Modulation (FM)

With frequency modulation, the frequency of the carrier is changed, unlike analogmodulation where the amplitude is changed. The most widely known use of FM isin radio station broadcasting.

The modulation of frequency happens when the frequency of the carrier ischanged according to the modulation signal. The frequency of the modulated signalwill be

( ) ( )ω ωFM c f mt k u t= + ⋅ (6.21)

If the modulating signal has the cosine form:

( )u t U tm m m= cos ω (6.22)

then the modulated frequency can be written as

( )ω ω ωFM c f m mt k U t= + ⋅ cos (6.23)

Factor kf determines the largest frequency change at a certain amplitude of themodulating signal. The largest frequency deviation of the carrier frequency is

∆∆

fk

UFMFM f

m= =ω

π π2 2(6.24)

The carrier frequency of the modulated signal will be

( )f t f f tFM c FM m= + ∆ cos ω (6.25)

The waveform of frequency modulation is shown in Figure 6.7.The waveform of the FM signal is determined with

( )u t U t U t mFM cm cFM

mm cm c f mtcos sin cos sinω

ω

ωω ω ω+

⎡

⎣⎢

⎤

⎦⎥ = +

∆ [ ]t (6.26)

where the modulation index mf is calculated from

mf

f

k U k U

ffFM

m

FM

m

f m

m

f m

m

= = = =∆ ∆ω

ω ω π2(6.27)


The modulation index is therefore the ratio of the frequency deviation and mod-ulating frequency. It can be higher or lower than 1. The modulation index dependson both the frequency and amplitude of the modulating signal. With AM, only theamplitude determines the modulation index.

Demodulation of the FM signal is performed with slope detection as shown inFigure 6.8.

The slope detector consists of an FM to AM converter and an AM envelopedetector (described in Section 6.2.1.1). The first part of the detector is actually alowpass RC filter whose cutoff frequency is determined from

fRCc = 1

2π(6.28)

and is chosen to be the carrier frequency of the FM signal. The signal is nextprocessed with the envelope detection circuit, as if it were an AM modulated signal.


Ucm

uFM

0 t

0 t

um

0

Ucm

uc

t

Figure 6.7 Frequency modulation waveforms.

6.2.1.3 Phase Modulation (PM)

Phase modulation, along with frequency modulation, is a form of angle modula-tion. Here, the phase of a carrier signal is changed depending on the modulating sig-nal. The phase modulation is closely related to the frequency modulation becausethe frequency cannot be changed without varying the phase. Thus, with phase mod-ulation there is always parasitic frequency modulation and vice versa.

The change of the carrier phase is determined with the following expression:

( ) ( ) ( )ϕ ϕ ϕ ϕFM p mt k u t t= + ⋅ = +0 0 ∆ (6.29)

Since the relative phase of the carrier signal, ϕ0, does not influence the modu-lated signal, it can be 0. The angle of sinusoidal function is called the phase of mod-ulated signal and is:

( ) ( )ΦPM c p mt t k u t= + ⋅ω (6.30)

The phase modulated signal can be described as

( ) ( )[ ]u t U t k u tPM m c p m= +cos ω (6.31)

and if the modulating signal has a sinusoidal waveform,

( )u t U tm m m= sin ω (6.32)

the phase of the modulated signal becomes

( ) ( ) ( )Φ ∆ΦPM c p m m c PM mt t k U t t t= + ⋅ = +ω ω ω ωsin sin (6.33)

where factor kp determines the largest phase change at some amplitude of the modu-lating signal. The largest phase shift of the modulated signal is called the phase devi-ation or ∆ΦPM. This phase deviation is also the modulation index, or

m k Up PM p m= =∆Φ (6.34)

The same as with frequency modulation, mp can be higher or lower than 1.Figure 6.9 shows the waveforms of the carrier signal, modulating signal, and

modulated signal for phase modulation.


Demodulatedsignal

AMenvelopedetector

FMsignal R

C

Figure 6.8 FM demodulation with slope detection.

As can be seen, the phase deviation changes with the angle of the modulatingsignal. It can be positive or negative.

Demodulation of PM is performed using frequency coherent demodulation (i.e.,using a reference signal with a fixed phase reference). The circuit used is the phasedetector, whose principle is beyond the scope of this book.

PM has a better demodulated S/N ratio than FM, but since coherent demodula-tion was not as simple as envelope detection in the past, FM was spread more thanPM, especially in broadcasting. Although today this is not a problem anymore, FMstill stays more in use than PM.

6.2.2 Digital Modulation

The difference between analog and digital modulations is in the modulating signal,which is analog in analog modulations and digital in digital modulations. The digi-tal modulating signal will be either 0 or 1. The three main types of modulations areamplitude shift keying (ASK), frequency shift keying (FSK), and phase shift keying(PSK). In this chapter, pulse code modulation and quadrature amplitudemodulation will also be covered.


Figure 6.9 Phase modulation waveforms.

6.2.2.1 Amplitude Shift Keying (ASK)

Amplitude shift keying (ASK) is a modulation method where the amplitude of theanalog signal carrier, usually sinusoidal, is changed according to the digital modu-lating signal. This type of modulation is also called on-off keying (OOK), where thesignal exists when the digital modulating signal is equal to 1 and there is no signalwhen the digital modulating signal is equal to 0. ASK is used in optical communica-tions. The principle of ASK modulation is shown in Figure 6.10.

The modulated signal is obtained by modulating the carrier signal having fre-quency fc with the modulating signal having frequency fm:

( )u t U t t tASK cm c m m= ⋅ + − +⎡⎣⎢

⎤⎦⎥

cos cos cosωπ

ωπ

ω12

2 23

3 (6.35)

The ideal ASK signal has an infinite spectrum. It is therefore necessary to shapethe modulating pulses. The disadvantage of the ASK modulation is that the ratioS/N does not apply for 1 and 0 of the modulating signal. There is also a large differ-


Ucm

uASK

0

um

0

Ucm

uc

0

1

t

t

t

Figure 6.10 ASK waveforms.

ence in the power consumption between these two states. Another disadvantage isthat the loss of link is read as 0.

Since the ASK modulated signal has a very definitive envelope, an envelopedetector can be used for the first step in the demodulation process. Further process-ing is usually required because the shaping of modulating pulses has to be donebefore transmission in order to limit the frequency bandwidth.

6.2.2.2 Frequency Shift Keying (FSK)

Frequency shift keying is a modulation method where the frequency of the analogsignal carrier, usually sinusoidal, is changed discretely according to the digital mod-ulating signal. The simplest FSK modulation is BFSK or binary FSK modulationwith two carrier frequencies. Another derivative of FSK is minimum shift keying(MSK) where the modulation index is smallest or 0.5. One type of MSK is called theGaussian MSK, which is used in mobile telephone standards. The principle of FSKmodulation is shown in Figure 6.11. Usually a higher level of the modulating signalis associated with a higher frequency.


Ucm

uFSK

0 t

0 t

um

0

Ucm

uc

t

1

Figure 6.11 FSK waveforms.

The modulation index is equal to the ratio of frequency deviation ∆f and modu-lation frequency fm:

mf

fFm

= ∆(6.36)

where frequency deviation is defined as

∆ff f

=−1 0

2(6.37)

and current frequencies of FSK signal being f0 = fc − ∆f and f1 = fc + ∆f , with fc beingthe carrier frequency.

FSK modulation is a nonlinear procedure whose frequency spectrum has manycomponents. It can be obtained with two ASK signals, with carrier frequencies f1

and f2. The ASK modulation is a linear procedure with a much simpler frequencyspectrum. If FSK is obtained with two ASK signals, envelope detection can be usedfor demodulation. In other cases, synchronous or asynchronous demodulation isnecessary. Synchronous demodulation of FSK signals is shown in Figure 6.12.

As mentioned before, the demodulator requires two local oscillators which gen-erate the carrier frequency and must be synchronized. There are two low pass filters,tuned to f1 and f2. At the end, a decision is made as to which of the two signals is theright one. This type of demodulator actually has two receiving channels. Asyn-chronous demodulation of the FSK signal is shown in Figure 6.13.

Asynchronous demodulation uses the advantage of the fact that the FSK modu-lation can be achieved with two ASK signals. The FSK signal is separated with bandpass filters 1 and 2. The filter outputs can be demodulated like ASK signals withenvelope detectors. At the end, the decision is made as to which of the two signals iscorrect.


FSKsignal

FSKsignal

LFP

LFP

Demodulatedsignal

Decision

f2

f1

Figure 6.12 Synchronous demodulation of FSK signals.

6.2.2.3 Phase Shift Keying (PSK)

Phase shift keying is a modulation method where the phase of the analog signal car-rier, usually sinusoidal, is changed according to the digital modulating signal. WithPSK the relative phase of the modulated signal can have two or more differentphases from the previously defined set of phases.

The waveforms of PSK are shown in Figure 6.14.


FSKsignal

FSKsignal

BPF 2

BPF 1 Envelopedetector

Envelopedetector

Decision Demodulatedsignal

Figure 6.13 Asynchronous demodulation of FSK signals.

Ucm

uPSK

0 t

1

t

um

0

Ucm

uc

t

0

Figure 6.14 PSK waveforms.

The PSK modulation can also be obtained from two ASK signals, just like theFSK modulation. The difference is that with PSK these two signals must be in aquadrature relation. The PSK signal will be

( ) ( ) [ ]u t U t U t tPSK cm c m cm m c m c= + = ⋅ − ⋅cos cos cos sin sinω ϕ ϕ ω ϕ ω (6.38)

where ϕm is the modulating phase obtained from

( )ϕ

πm

n c

Mn M c=

+= − =

20 1 2 1 0 1, , , , , ; , (6.39)

For M = 2, and c = 0, the modulating phases can be ϕm = 0, π, and for c = 1,another set of modulating phases will be ϕm = π/2, (3π)/2. This modulation is calledBPSK or binary phase shift keying. Usually phase 0π or 0° is dedicated to state 1 andπ or 180° to binary state 0. Their relation is shown in Figure 6.15.

BPSK modulation is very resilient to interference, but its spectral efficiency isnot very high. It is possible to use a set of phases much higher than M = 2. For M = 4,there will be four different phases. This type of PSK modulation is called QPSK. Thepossible set of phases will be for c = 0, ϕm = 0, π/2, π, 3π/2, and for c = 1, ϕm = ±π/4,±, 3π/4. With QPSK, there will be two bits necessary for each state instead of one bitfor BPSK. This requires more memory and increases the possibility of error com-pared to BPSK. The spectral efficiency of QPSK is increased compared to BPSK,which means that the amount of information that can be carried in a communica-tion channel is doubled. QPSK phases are shown in Figure 6.16. Each state isassigned binary digits according to the Gray code where the neighboring states dif-fer only by one digit.

The QPSK signal can be described with

( ) ( ) ( )u t I t t Q t tQPSK c c= −cos sinω ω (6.40)

The QPSK signal can be achieved by combining two BPSK signals.There are other types of PSK modulations with a higher number of phases (8,

16, ...). However, by increasing the number of phases, the possibility of error alsoincreases, since the phases are coming closer to each other. This can be compensated


180° 0°I

“0” “1”

Q

Figure 6.15 BPSK phases in I-Q plane.

for by increasing the power; however, in some cases this is not possible because ofinternational power levels according to standard regulations.

PSK modulated signals cannot be demodulated using envelope detection.Demodulation is possible only with synchronous demodulation, which means thatthe receiver will have to include a referent signal, which will provide the signal onthe carrier frequency.

The quadrature demodulator shown in Figure 6.17 is used for demodulatingPSK signals (i.e., for obtaining I and Q components).

6.2.2.4 Pulse Code Modulation (PCM)

Pulse code modulation is a process where the amplitude of an analog signal is sam-pled and quantized into a binary code.

The principle is given in Figure 6.18.PCM requires analog-to-digital conversion (ADC). The sampling rate of an

analog signal must be at least twice the frequency of an analog signal, that is

f fs c≥ 2 (6.41)


00 10

I

1101

Q

Figure 6.16 QPSK phases in I-Q plane.

FSKsignal

FSKsignal

LFP

LFP

Demodulatedsignal

Decision

f2

f1

Figure 6.17 Quadrature demodulator.

The level of quantization will determine how closely the analog signal is sam-pled. More levels will result in a closer resemblance of the sampled and quantizedsignal to the analog signal, but will require more processor memory.

The demodulator works in a similar way. It takes the numbers (bits) and assignsthem the analog voltage accordingly. The receiver will require an analog-to-digitalconversion (DAC) circuit to perform this operation.

6.2.2.5 Quadrature Amplitude Modulation (QAM)

The quadrature amplitude modulation is a process where the amplitude of two car-riers is changed. The two carriers are usually sinusoidal and have a phase differenceof 90°. QAM can be either analog or digital.

The analog QAM works similar to AM. The difference is that QAM uses twocarriers instead of one, which both have the same frequency, but their phase differ-ence is 90°. They are modulated with two different modulating signals and thencombined before transmission. The QAM signal has the following form:

( ) ( )u u t t u t tQAM m c m c− +1 2cos sinω ω (6.42)

Modulation changes the amplitude of carrier signals, but not the phase differ-ence between them. Analog QAM is demodulated using synchronous detection.

Digital QAM is used more than analog QAM, so it is enough to call it justQAM. Digital QAM has two modulating signals, I(t) and Q(t), whose relation is asfollows

( ) ( )u I t t Q t tQAM c ccos sinω ω− (6.43)


000001010011100101110111

01234567

uPCM

0

1

um

0

uc

t

t

Figure 6.18 PCM modulation.

PSK is actually a special case of QAM. The difference is that QPSK has noamplitude modulation while QAM has. QPSK has a constant amplitude. Dependingon the quantization level, QAM can be 4 QAM, 16 QAM, 64 QAM, and so forth.Figure 6.19 shows the 8-QAM and 8-PSK states. The more states there are, thehigher the possibility of error.

6.3 Control of System Drift

If there is a difference between the carrier frequency in the transmitter and the car-rier frequency in the receiver (which is necessary for the synchronous demodulator),a frequency offset will occur.

∆f f fcr ct= − (6.44)

International standards require that the frequency offset be kept under a certainlevel. For example, for the HIPERLAN/2 transmitter carrier frequency fc, the ratio∆f/fc must be less than 0.002%. If the same demand were required for the receiver,∆f/fc would need to be 250 kHz. This means that the majority of the power from thetransmitted subcarrier will be received in the neighboring channel, which will leadto a large bit error rate (BER).

The frequency offset must be measured in the receiver system and corrected.This can be done with a voltage controlled oscillator and phase-locked loop.


ETSI EN 300 910 V.8.5.1. (2000-11), Digital Cellular Telecommunications System, (Phase 2+);Radio transmission and reception (GSM 05.05 version 8.5.1 Release 1999)Feher, K., Wireless Digital Communications, Upper Saddle River, NJ: Prentice-Hall, 1995.Modlic, B., and I. Modlic, Modulacije i modulatori, Skolska knjiga Zagreb, 1995.Sklar, B., Digital Communications—Fundamentals and Applications, 2nd ed., Upper SaddleRiver, NJ: Prentice-Hall, 2001.


Q Q

II

8-QAM 8-PSK

Figure 6.19 8-QAM and 8-PSK modulations.

Van Hoesel, L. F. W., et al., “Frequency Offset Correction in a Software Defined Hiperlan/2Demodulator Using Preamble Section A,” Proceedings of the Third International Symposium onMobile Multimedia Systems & Applications, Delft, the Netherlands, December 6, 2002,pp.51–62.Xiong, F., Digital Modulation Techniques, Norwood, MA: Artech House, 2000.

6.3 Control of System Drift 121

C H A P T E R 7

Electromagnetic Field Coupling to Wire

7.1 Field-to-Wire Coupling

Every wire that acts as a conductor carries charge. When a wire is placed in the elec-tromagnetic field, it will be under its influence. This influence will depend on thetype of wire, the intensity of the field, and the quality of the shield. If a conductor isused for carrying information, a foreign electromagnetic field may cause errors inthe receiving system. Any cable can also act as an antenna (i.e., it can radiate orreceive unwanted interfering signals).

The flow of the current largely depends on the frequency. As the frequencyincreases, the current tends to flow more closely to the surface due to the skin effect.The resistance R of the wire depends on the length, l, cross-section, S, and resistivityof the material, ρ:

RS

= ρ1

(7.1)

The resistivity of some materials is given in Table 7.1.

7.1.1 Skin Effect

At high frequencies the resistance is not the same as it is on DC current. The surfaceresistivity of the wire dependant on the frequency can be calculated from

R fs = = =π µρδσ

ρ

δ

1(7.2)

where f is the frequency, µ is the permeability of the wire, and ρ is the resistivity

(ρσ

=1

, with σ being the conductivity and δ being the skin depth). The higher the fre-

quency, the smaller the skin depth—and the resistivity increases. This is shown inFigure 7.1.

The current flows only in the dashed area. At DC, the wire cross-section is usedfor the current flow. At higher frequencies, the current tends to stay closer to thesurface. At RF frequencies, the current is concentrated only at the surface and thepenetration depth is very small. This characteristic is used for building shields fromelectric fields.

The AC resistance can be written as

123

RS ef

= ρ1

(7.3)

where the effective area, Sef, is obtained by

( )S r r r ref = − ′ = − −π π π π δ2 2 2 2(7.4)

Further analysis gives

( ) ( )S r r r ref = − + − = −π δ δ π δ δ2 2 2 22 2 (7.5)

The resistance of a conductor at radio frequencies will then be

( )[ ]R

r=

−ρ

π δ δ

1

2 2(7.6)

124 Electromagnetic Field Coupling to Wire

Table 7.1 Resistivity of DifferentMaterials at 20°

Conductor Resistivity (n m)

Aluminum 28.2

Copper 17.2

Gold 24.4

Iron 97.1

Platinum 106

Lead 220

Silver 15.9

Zinc 58

Semiconductor Resistivity (m m)

Carbon 0.015

Germanium 460

Silicon 250,000

Insulator Resistivity (T m)

Glass 0.01–100

Rubber 10–10,000

Wood 0.01

DC

r

HF

δ

r´

Figure 7.1 Skin effect.

The resistance of a copper wire (1m long, 1.5 mm in diameter) at different fre-quencies is shown in Figure 7.2. It can be seen that the resistance rises quite fast withthe frequency. At 200 MHz all of the current will flow in about 5 outer micrometersof the cable.

As frequencies increase, cables have difficulty carrying the signals properly.Additionally, it is difficult to prevent them from leaking. Regarding EMC problems,nonmetallic conductors—which include wireless, fiber-optic, microwave, or laserlinks—are better for carrying signals.

Coupling paths between transmitters and receivers can be either radiated orconductive and include antenna-to-antenna coupling, cable-to-cable coupling,antenna-to-wire (cable) coupling, or cable-to-antenna coupling as shown in Figure7.3. Crosstalk or cable-to-cable coupling depends on frequency and bandwidth.

7.1.2 Unshielded Twisted Pair (UTP)

When current flows through the wire, electromagnetic radiation is inevitable. Whentwo cables (forward and return) are needed, it is possible to twist them into a

7.1 Field-to-Wire Coupling 125

0

0 1.

0 2.

0 3.

0 4.

0 5.

0 6.

0 7.

0 8.

0.9

0 1 2 5 10 20 50 100 200f (MHz)

Resi

stan

ce(o

hms)

Figure 7.2 Resistance of 1-m-long copper wire versus frequency.

T R

Radiatedcoupling

Transmitterequipment

ReceiverequipmentSignal line

Conductedcoupling

Figure 7.3 Coupling paths.

twisted pair for the cancellation of electromagnetic interference. This old techniqueis called UTP or unshielded twisted pair. Here, the signals flowing in a pair haveopposite directions and the fields tend to cancel each other out. This solution cansometimes effectively cancel crosstalk between neighboring pairs. This method hasbeen used for telephone lines for many years now.

An unshielded twisted pair usually comes in two colors (Figure 7.4). It is usedfor the Internet, telephone cables, and video. There are usually between 4 to 25 pairsinside a sheath. While the UTP cable has no shield, there are other designs of wireswith shielding, which will be covered in Section 7.4.3.

Whether the conductors are used inside or outside a certain electronic devicewill determine the type of wire used. If used inside a product with a good shield, thechoice of wire is not so important in regards to interference, although signal perfor-mance might be of interest. It would be best not to have internal cables at all, andinstead have PCB traces. This simplifies the shielding and reduces its cost.

If conductors are used outside of an electronic device, a shield of any sort isdesirable regardless of whether the communication is analog or digital.

7.1.3 Ferrite Filter

Filtering the signals with ferrite filters (Figure 7.5) can sometimes help protectagainst interference. A ferrite filter can, if properly used, suppress interference. It ispossible to use an in-line filter or onboard suppression circuits, but they are usuallymore expensive solutions. Ferrites have a concentrated homogenous magneticstructure with high permeability. Their characteristics do not change with time andtemperature, and their application depends on the frequency of use. Prior to theusage of a ferrite filter, the cable impedance has to be known. It is usually 50Ω, butit can also vary from a few ohms to several hundreds of ohms. Ferrite impedancedepends on dimensions (length, outer and inner diameter). The most importantthing to have in mind is that the ferrite diameter should be as close to the wirediameter as possible.


Sheath

4 pairs UTPConductor

Isolation

Figure 7.4 Unshielded twisted pair (UTP).

The insertion loss (a measure of the filter effectiveness at a certain frequency),which is usually described as the ratio of voltage with and without a filter, can becalculated from

Insertion loss =+

20 10logZ Z

ZC F

C

(7.7)

where ZC is the circuit impedance and ZF is the ferrite impedance. If, for instance,circuit impedance is 50Ω and ferrite impedance at 40 MHz is 100Ω, the insertionloss will be 9.5 dB. If circuit impedance is 75Ω instead of 50Ω, the insertion loss willbe 7.4 dB. This means that the insertion loss will be higher when the circuit imped-ance is lower.

The effectiveness of the ferrite can be enhanced if cable or wire is passedthrough the ferrite more than once. The ferrite impedance, ZF, increases geometri-cally with the number of loops (Figure 7.6). For two loops, the ferrite impedance isfour times bigger.

The disadvantage of multiple loops is that the frequency band of ferrite becomesnarrower. If, for the previous example, there are two loops instead of one, ferrite

7.1 Field-to-Wire Coupling 127

Ferrite filter

Connector

Cable

Figure 7.5 Ferrite filter.

Figure 7.6 Multiloop ferrite.

impedance ZF will be 400Ω at 40 MHz. The insertion loss in this case will be 19.1dB and for three loops it will be 25.6 dB.

Ferrite placement is also important. It is best to place the ferrite close to the endof cable where it leaves the electronic equipment. If the cable connects two pieces ofthe electronic equipment, two ferrites might be necessary.

Large ferrites generally have higher impedance, but since their size increases thetotal weight and space, this has to be kept in mind when choosing the ferrite.

Filtering only attenuates the interfering signals—it does not remove them. Forsome cases, cable shielding is necessary.

The coupling to wires can be either capacitive (electric field) or inductive (mag-netic field).

7.2 Electric Field Coupling to Wires

An electric field can be coupled to wire by stray capacitance. Figure 7.7 shows howan electric field is coupled into a wire (circuit) carrying the useful signal. This elec-tric field can come from several sources such as another parallel cable or circuit, orfrom the electric field of an antenna.

An interfering electric field is coupled through stray capacitance in an equiva-lent circuit, as shown at the bottom of Figure 7.7. The signal source impedance, RS,and load impedance, RL, are the same. The useful signal is characterized with VS,and the source of noise, VN, is coupled via stray capacitance, CN.

Stray capacitance occurs when two conductors are close to each other and thereis no shield or grounding present. It usually occurs between parallel traces on a PCboard. Bad planning can lead to lower stability, greater noise, and reduced fre-quency response. The stray capacitance is proportional to the area of interlap, S,and inversely proportional to the distance between two circuits, d, as:

CSdr= ε ε0 (7.8)

Increasing the distance or minimizing the overlap will minimize the capacitanceand thus the coupling of noise into the signal. The coupling also depends on noiselevel, frequency, and load impedance. Coupled voltage, VC, will be equal to

V VR

RC

C NL

LN

=+ 1

ω

(7.9)

Capacitive coupling will be smaller for lower values of noise voltage, frequency,and circuit impedance.

The voltage induced on a wire from the electric field using Faraday’s law is:

V E dlddt

B dsi C S= ⋅ = − ⋅∫ ∫∫

(7.10)


where

E is the electric field intensity vector and

B is the magnetic flux density vector.The above expression can be simplified to

V Ehl

i = ⎛⎝⎜

⎞⎠⎟

22

sinβ

(7.11)

where h and l from Figure 7.7 define the loop area. If the phase constant β = 2π/λ isintroduced in the above expression, the equation for induced voltage is obtained:

V Ehl

i = ⎛⎝⎜

⎞⎠⎟

2 sinπ

λ(7.12)

For low frequencies, where wire length l is short relative to the wavelength (l <λ/2), sinx = x, in which case the above expression becomes

V lhEi = 2π λ (7.13)

Above a frequency at which l = λ/2, (7.13) can be simplified:

V hEi = π (7.14)

7.2 Electric Field Coupling to Wires 129

RS

VS

RL

VNCN

RS

RL

l

h

VS

E-field

Figure 7.7 Capacitive coupling.

This is a high-frequency asymptote, overinduced voltage at all frequenciesabove the one where l = λ/2.

If, for example, we take a case with l = 1m and h = 0.5 cm, the normalized elec-tric field-to-wire coupling, Vi /E will depend on the frequency as shown in Figure7.8.

The dashed line shows the approximation for low and high frequencies dis-cussed above. The maximums of the coupling will appear when

( )f

n c

ln=

+=

2 1

20 1 2, , , , (7.15)

and the minimums will appear when

fncl

n= =, , , ,0 1 2 (7.16)

where c is the speed of light.Electric fields can be coupled using common mode coupling or differential

mode coupling. With common mode coupling the currents in cables run in the samedirection, while with differential mode coupling the currents run in opposite direc-tions. Both examples are shown in Figure 7.9.


−60

−50

−40

−30

−20

−10

0

1 10 100 1000f (MHz)

VE/ i

Figure 7.8 Electromagnetic field-to-wire coupling versus frequency.

i c id

i c id

Figure 7.9 Common mode coupling and differential mode coupling.

7.3 Magnetic Field Coupling to Wires

If current is flowing through a wire it will create a magnetic field around the wire.The magnetic field creates an electric field perpendicular to the magnetic field. Thiselectric field can cause a current flow in the wire in its vicinity. This characteristic ofmutual inductance is the basis of how transformers actually operate.

Magnetic coupling of interference is unwanted inductive coupling from oneloop to another. Typical noise sources are motors, transformers, relays, and soforth.

Inductive coupling is the result of a magnetic field in the area enclosed by thesignal circuit loop. The magnetic field is generated from the current flowing in theadjacent noise circuit as shown in Figure 7.10.

The induced voltage VN in the signal circuit is calculated from:

V fBSN = 2 cos ϕ (7.17)

7.3 Magnetic Field Coupling to Wires 131

VS

VS

RS

RS

RL

RL

R

R

M

VN

VN

IN

IN

Figure 7.10 Inductive coupling.

where f is the frequency of the noise signal, B is the magnetic flux density, S is thearea of the signal circuit loop, and ϕ is the angle between the flux density, B, and thearea, S.

The induced voltage according to the equivalent circuit shown at the bottom ofFigure 7.10 is

V fMIN N= 2 (7.18)

where IN is the current value in the noise circuit.Mutual inductance, M, is proportional to the area of the receiver circuit loop,

frequency of the source, and current level of the source, and is inversely propor-tional to the distance between the signal and loop circuit. Thus, the coupling can beminimized by separating these two circuits. The coupling can also be minimized bylowering the frequency and current of the noise circuit or twisting the wires of thenoise source. Twisting the wires reduces the circuit loop area, S.

Finally, magnetic shielding of both the noise circuit as well as the signal circuitcan further lower the magnetic (inductive) coupling.

The ratio of noise voltage to signal voltage is lowest when the circuit impedanceis highest. This condition is exactly the opposite for the case of minimum electricfield coupling for the lowest circuit impedance. There is an optimum circuit imped-ance where the overall coupling will be the smallest.

7.4 Cable Shielding

Cable shielding is a procedure of protecting the wires (conductors), which are carry-ing information or are used as power lines, with an outer protective layer. Theshield protects the wire from the electric field coming from the noise source and atthe same time protects the surrounding electronic equipment or other cables fromthe wire’s own electromagnetic radiation.

In Section 7.1.2 the twisted pair was mentioned as a method for protection ofthe wire from unwanted electromagnetic coupling. In some cases this is not suffi-cient. An additional metal shield provides better noise suppression.

The parasite, or stray capacitance, CN, can be reduced by applying the capaci-tive shielding. The coaxial cable is a good example of this method. The idea is toprovide another path for the induced current rather than the wire carrying the sig-nal. The shield is placed between the capacitively coupled conductors and con-nected to the ground only at the source end. If the shield is connected to the groundat both ends, a large ground current may exist, so this is generally notrecommended.

If a cable shield is thick (several skin depths), and has no holes in it, it will have ashielding effectiveness (SE) above 200 dB. Shielding effectiveness will be discussedin depth in Chapter 8. The conducted currents flowing on the surface of the shieldwill not penetrate the shield and interfere with the wires inside, which carry usefulsignals. This type of shield does not require grounding at all.

If, on the other hand, the shield is thin (less than skin deep) and leaky (i.e., withholes), the absorption loss will be small. There will only be the reflection loss of theshield; the holes might further lessen the reflection loss. The only protection this


shield can give to the cable would occur if it were grounded. In this case it must beconnected to the closest metal reference to form a Faraday cage. If the interferingnoise field is uniform, it does not matter which end of the shield is grounded. Inother cases, it is better to ground the end closer to the interfering noise source.Grounding the shield at both ends generally is not a good idea because this will cre-ate a new ground loop, resulting in an even greater surface current. Since the shieldhas holes, the greater surface current would induce high voltages in internal wires.Thus, if a bad cable shield is used, grounding only one end is desirable.

7.4.1 Tri-Axial Cable

Shielding cables differentiates shielding and the screening. Screening is actuallyshielding more wire pairs and not just one pair. Usually more screening and shield-ing will raise the cable cost.

The cable screen should cover the entire length of cable with 360° coverage. Thescreen (shield) carries both the return signal and external interference on oppositesides of the cable (due to the skin-depth effect), which is possible for solid copperscreens. Flexible screened cables cannot keep the two currents separated, so thereturn currents leak out and the interfering currents leak in. The solution for thisproblem is a tri-axial cable shown in Figure 7.11.A tri-axial cable is quite similar to the coaxial cable, except with the addition of onemore shield. One center conductor is surrounded by two shield layers insulatedfrom each other. Usually the outer shield is grounded and the inner shield is used forthe return signal. The shielding properties of such a cable are better than those for asimple coaxial cable. The outer shield lowers the ground loop interference and elim-inates the radiated noise or crosstalk.

7.4.2 Cable Termination

The shield, as mentioned before, must be terminated (or matched). Cable screensshould always be connected to their enclosure shield, and should be terminated in360° to the skin of the screened enclosure they are going into. The choice of the con-nector at the end of the cable is essential.

The most common connection is the pigtail connection. It is the connectionwhere the screen is brought down to a single wire and extended through a connectorpin to the ground point. It is easy to assemble. At high frequencies this connection

7.4 Cable Shielding 133

Copperconductor

Foilshield

Coppershield

Dielectric insulator Plastic coating

Figure 7.11 Tri-axial cable.

becomes inappropriate because of its inductance, which is serially connected to thecable screen. The inductance will introduce a voltage when interference currentsflow along the screen to the ground. This voltage can be coupled from the screeninto the inner conductors. The same goes for the emission of radiation from theinner conductors to the ambient area. The impedance of such a connection increaseswith the frequency and may completely degrade the use of a good cable screen.

The pigtail connection should be avoided at frequencies above 10 MHz if possi-ble, or kept very short. A much better solution is crimping and employing backshellin combination with stress relief bolts as shown in Figure 7.12.

Connector backshells can be potted and molded. Even the best cable shield willbe useless if the bonding to the connector is done poorly. Some sort of enclosureshielding at the end of the cable is always necessary, especially at higher frequencies.

7.4.3 Shielded Twisted Pair Cables

The previously mentioned unshielded twisted pair (UTP), which is the most simpleand cheapest cable available, is sometimes not appropriate because of its lack ofshielding. There are other types of twisted pairs with screening and shielding. Theshielding can be applied to individual pairs or several pairs in a cable. The first con-sidered is the screened unshielded twisted pair (S/UTP).

The S/UTP cable is a screened UTP cable, which means that it has a single shieldbeneath the sheath for all of the twisted pairs together as shown in Figure 7.13.Since the shield is a metal foil, it is sometimes called the foiled twisted pair (FTP).

The next type of shielded twisted pair is the shielded twisted pair (STP), whichhas metal shielding over each individual pair of wires instead of just one shield forall the twisted pairs together like S/UTP. It provides better protection from electro-magnetic interference and prevents crosstalk between the pairs as well. Thecross-section of STP is shown in Figure 7.14.

The last shielded cable considered is the screened shielded twisted pair (S/STP),also known as the screened fully shielded twisted pair (S/FTP) cable. This cable hasshielding of the pairs as well as a shield beneath the sheath. It has the best protectionfrom interference of all the cables mentioned in this section. The cross-section isshown in Figure 7.15.


Cable

Stress relief bolts

Backshell Connector

Figure 7.12 Bonding cable to connector.

7.4 Cable Shielding 135

Pair

Screen

Sheath

Conductor

Insulator

Figure 7.13 Screened unshielded twisted pair (S/UTP) cable.

Pair shield

Sheath

Conductor

Insulator

Figure 7.14 Shielded twisted pair (STP).


“Crimping, Interconnecting Cables, Harnesses, and Wiring,” NASA-STD-8739.4, February1998, http://snebulos.mit.edu/projects/reference/nasa-generic/nasa-std-8739-4.pdf.EMC and Compliance Yearbook 2003, CD-ROM, Nutwood UK Ltd, Eddystone Court, De LankLane, St Breward, Bodmin, Cornwall, PL30 4NQ, United Kingdom.Javor, K., “On Field-to-Wire Coupling Versus Conducted Injection Techniques,” Proceedings ofIEEE International Symposium on Electromagnetic Compatibility, August 18–22, 1997, Austin,TX, pp. 479–487.Martin, L., and A. Kamiens, “Magnetic Shielding Theory and Practice,” ITEM 2001, pp. 1–3.May, J., “Filtering Out Interference Signals with Cable Ferrites,” Compliance Engineering Maga-zine, November 2002.Trout, D., “Investigation of the Bulk Current Injection Technique by Comparison to InducedCurrents from Radiated Electromagnetic Fields,” 1996 IEEE EMC Symposium, Santa Clara, CA,1996, pp. 412–417.Worshevsky, A., and R. Patlaty, “Low Frequency Common Mode Voltages in Electrical CableRuns,” Proceedings of IEEE 6th International Symposium on Electromagnetic Compatibility andElectromagnetic Ecology, June 21–24, 2005, Saint Petersburg, Russia, pp. 224–225.


Pair shield

Screen

Sheath

Conductor

Insulator

Figure 7.15 Screened shielded twisted pair (S/STP).

C H A P T E R 8

Electromagnetic Field-to-ApertureCoupling

8.1 Field-to-Aperture Coupling

The aperture is a hole or an opening in an enclosure of an electronic device throughwhich electromagnetic fields may enter or leak out. Since immunity of an electronicdevice is defined as its resistance to ambient electromagnetic fields, any holes orapertures in the shielding may compromise the operation of such a device. At thesame time, other electronic devices in the vicinity of the above mentioned electronicdevice, which has holes or apertures in its shield, will be in danger from its emis-sions. The example of this is shown in Figure 8.1.

Electronic device emissions are not a problem if they are kept below levels pre-scribed by international standards. Most of the emissions will stay inside the shieldand diminish with multiple reflections. Some emissions may leak out from the shieldand reach other electronic equipment in the vicinity. A shield aperture can be poten-tially dangerous to other electronic equipment as shown in Figure 8.1(a). Similarly,the aperture in the shield can leak in the electromagnetic radiation. The shield pro-vides good protection from most of the ambient radiation, but some might still pen-etrate the shield, as shown in Figure 8.1(b). Some of the radiation that enters theshield will be lessened through multiple reflections (and absorptions) from theshield, but not all.

The best shield will be the one without any holes or apertures; no electromag-netic fields can enter the shield and interfere with the electronic equipment. Also,this kind of equipment will not produce any ambient fields, provided that the shieldis designed in such a way as to protect from both the electric and magnetic fields.This type of shield would be beneficial even though it is impossible to achieve totalshielding against electric or magnetic fields; however, such a shield is impractical.This is because there has to be some sort of power cord going in or out of the shield.Also, apertures for ventilation, sensors, antennas, and connectors to other equip-ment all must be taken into consideration. These are just some of the reasons why aperfect shield is impossible to achieve.

Since the apertures in the shield cannot be avoided, there are still methods avail-able to keep the influences of these apertures on the shield effectiveness at aminimum.

137

8.1.1 Shielding Effectiveness (SE)

Apertures in a metal shield can be considered as half-wave resonant slot antennas.This means that the geometrical dimensions of the apertures will determine whichfrequencies can travel through the slot or aperture and which cannot. If there is onlyone aperture in the shield, the shielding effectiveness (SE) can be obtained from

SEd

= ⎛⎝⎜

⎞⎠⎟

20210logλ

(8.1)

where λ is the wavelength and d is the maximum dimension of the aperture (diago-nal). Thus, for a desired SE, there is a maximum aperture, with the largest d allowedfor a given frequency:

( ) ( )d mm

f MHz= 150

(8.2)

Figure 8.2 shows the maximum allowable aperture diameter d for SE = 60 dB.For 1 MHz this value is 150 mm, and it becomes smaller, reaching only 0.15 mm for1 GHz.

8.1.2 Multiple Apertures

It is much better to have several smaller apertures than one large one, if possible.Induced currents (from the magnetic field) in the shield will flow as long as there isno obstruction in their path. Currents flowing in a shield coming to an aperture willcreate the magnetic fields. There will be a voltage difference on the aperture sides,

138 Electromagnetic Field-to-Aperture Coupling

Emission Immunity

(a) (b)

Figure 8.1 (a) Emission and (b) immunity depend on the shield apertures.

0

50

100

150

1 10 100 1000f (MHz)

Aperture dimensions for SE = 60 dB

d(m

m)

Figure 8.2 Aperture dimensions for SE = 60 dB.

which will result in an electric field. It is desirable that the apertures interfere withthose currents as little as possible, as can be seen from Figure 8.3.

As can be seen in Figure 8.3, smaller apertures will not stop the currents asmuch as a large one will. The resonant frequency of the smaller aperture will bemuch higher than that of the larger aperture.

Since every aperture in the shield lessens its effectiveness, the shielding effective-ness of multiple apertures is degraded compared to a single aperture by 20logn,where n is the number of apertures. That means that two apertures will degrade theshield by 6 dB, four apertures by 12 dB, and so forth. This works only up to thepoint where the wavelength becomes comparable with the size of the small aperturearrays, or in the case when the apertures are not close to each other compared to thewavelength. If the apertures are placed at a distance more than half of a wavelengthapart, they can be considered individual apertures. At a frequency of 1 GHz, half ofa wavelength will be 15 cm.

The smaller the aperture, the less the electromagnetic fields penetrate inside theshield. This is shown in the Figure 8.4. The electromagnetic field will not penetratevery deep if the aperture dimension is small compared to the wavelength. The effec-tiveness of the shield at the distance l from the shield, depending on the diameter dof the aperture, will be

201

1

401

2

601

5

dB if

dB if

dB if

d

d

d

≈

≈

≈

where λ > d. The rule of thumb is that d/ ≤ 30. For the frequency f = 1 GHz or λ = 30cm, d can be 1 cm at most. According to the above rule, the shielding effectiveness of60 dB will be achieved at the distance of l = 5 cm from the aperture.

8.1 Field-to-Aperture Coupling 139

One large aperture Several smaller apertures

Figure 8.3 Multiple aperture currents.

Figure 8.4 Penetration of the electromagnetic field through the apertures.

The above rules are only an approximation. In reality they will depend on thethickness of the shield, the type of the material used, and on the distance of the innercables or wires to the shield aperture. It is recommended to have an additionalshielding effectiveness of 20 dB.

8.1.3 Waveguides Below Cutoff

When the aperture dimension becomes too small, waveguides below the cutoff fre-quency should be used (Figure 8.5). There can be just one or multiple waveguidesforming the honeycomb. They are often used for ventilation.

Placing waveguides inside the aperture holes can reduce the emission of thewaves through the aperture. Multiple reflections inside the waveguide’s walls willreduce wave strength. The size of the aperture can be much larger when usingwaveguides below cutoff than when not using them.

The characteristics of the waveguide are determined by its geometrical dimen-sions: gap (g) and height (h).

A waveguide will allow all the waves to pass when its internal diagonal (g) ishalf of the wavelength. Thus, the cutoff frequency of the waveguide is determined as

fgc = 150000,

(8.3)

where fc is in megahertz and g in millimeters. Below its cutoff frequency, the wave-guide does not leak very much, and it will provide sufficient shielding for f fc/2.

The attenuation (SE) of the waveguide dependent on the frequency is given by

[ ]SE h

f

f

gc=

−⎛⎝⎜

⎞⎠⎟

272

12

. dB (8.4)

where both h and g are given in mm. Figure 8.6 shows the shielding effectiveness(SE) depending on the frequency for h/g = 3.

For f fc/2, according to the above graph, the SE will be about 70 dB, which issufficient for most purposes. A smaller g results in higher cutoff frequency, while alarger height h increases the value of shielding effectiveness (SE).


gh

Figure 8.5 Waveguides below cutoff.

If multiple waveguides similar to multiple apertures are used, there will be areduction in shielding by 10logn, where n is the number of apertures. Thus, 10 aper-tures will have 10-dB less attenuation than a single waveguide. If conductors areplaced inside the waveguides below cutoff, they will not be equally effective, and theshielding effectiveness will be greatly reduced, so this should be avoided.

8.2 Reflection and Transmission

The shielding theory is based on two mechanisms: reflection and transmission(absorption) losses. When an electromagnetic wave in free space—with electric andmagnetic fields perpendicular to each other (TEM mode of propagation)—hits ametal wall, one part will be reflected depending on the angle of incidence, and theother part will travel through the metal wall with attenuation (Figure 8.7).

The attenuation of the electromagnetic wave will be exponential, depending onthe skin depth, δ, and the distance from the border of the two mediums (free spaceand metal), d. The absorption loss through the metal shield at distance d can be cal-culated from

( )S eAd= 20 10log δ (8.5)

or

( )[ ]S dA = 8686. δ dB (8.6)

where δ is the skin depth at which the field intensity drops to the value of 1/e:

δ ωµ µ σ= 2 0 r (8.7)

8.2 Reflection and Transmission 141

0

10

20

30

40

50

60

70

80

90

0,00,2 0,4 0,6 0,8 1,0

f f/ c

SE

Figure 8.6 SE of waveguide below cutoff versus frequency for h/g = 3.

with f being the frequency, µr being the permeability of the material, and σ being theconductivity of the material.

Absorption loss SA increases with frequency. The necessary shield depth, d, isgetting smaller with the rise of frequency, which means that the shield will be moreeffective on a higher frequency than on a lower frequency. Figure 8.8 shows the alu-minum shield depth, d, for SA of 100 dB and 60 dB versus frequency.

Figure 8.8 shows that at higher frequencies a good shield can be achieved with avery thin aluminum foil. The results will be similar for any other metal.

A part of the electromagnetic field will not be absorbed but reflected. The freespace has an impedance of


Metal wall

H

E

P P

H e* −d/δ

E e* −d/δ

Figure 8.7 Absorption loss in the shield.

0,0000001

0,000001

0,00001

0,0001

0,001

0,01

0,1

1

10

1,E+00 1,E+01 1,E+02 1,E+03 1,E+04 1,E+05 1,E+06 1,E+07 1,E+08 1,E+09

Sa = 60 dB

Sa = 100 dB

d(m

)

f (Hz)

Figure 8.8 Necessary aluminum shield depth versus frequency.

Z0 120 377= = =µ

επ Ω (8.8)

The metal shield has a smaller impedance than free space:

Zs = ωµ

σ(8.9)

Since the two impedances are different, there will be a reflection. The transmis-sion coefficient rt1 at the border of the air and the shield is given as

rZ

Z Zts

s1

0

2=

+(8.10)

The electromagnetic wave must exit the shield again into free space (air) asshown in Figure 8.9, and the transmission coefficient at this second border of thetwo mediums will be:

rZ

Z Zts

20

0

2=

+(8.11)

The total transmission rttot is given as a product of rt1 and rt2:

( )r

Z Z

Z Zttot

s

s

=+

4 0

0

2(8.12)

Total reflection and transmission is equal to the incident wave:

r rt r+ = 1 (8.13)

Since a metal shield has a much smaller impedance than free space, the reflec-tion loss, SR, is given as


Air

Metal

Er

Ei

Et

EttEtr

Air

d

Figure 8.9 Reflection and transmission of an electromagnetic wave at a metal shield.

SZ

Z

ZR

s r

= =0 0

04 4 ωµ µ σ

(8.14)

or in [dB],

SZ

R

r

= 204

100

0

logωµ µ σ

(8.15)

The reflection loss decreases with the frequency. Figure 8.10 shows the reflec-tion loss of the aluminum shield depending on the frequency. The total shield losswill be the combined reflection and absorption loss:

[ ] [ ]S S S S SA R A R= = +dB dB (8.16)

Total shield loss for aluminum foil 0.1 mm in depth is shown in Figure 8.11.Total shield loss for aluminum foil that is 0.1 mm in depth is shown in Figure

8.11. The figure shows that the reflection loss is dominant on lower frequencies, upto 10 MHz; at 100 MHz their contribution is about the same, and on higher fre-quencies absorption loss becomes dominant. At even higher frequencies, the shieldthickness has almost no influence at all. Total shielding loss is the combination ofboth contributions (full line); it stays above 100 dB on all frequencies. Other metalsgive similar results.

If the shield consists of several laminate layers, the total reflection and absorp-tion losses will be a sum of reflections between each layer and the attenuation inevery layer.

The above discussion is valid for far field conditions where E/H = Z0 = 377ohms. In the near field, the ratio of the electric and magnetic field is different—itchanges depending on the distance from the electromagnetic source. Therefore, theshield effectiveness should be considered separately for electric and magnetic fields.


70

80

90

100

110

120

130

140

150

160

170

1,E+00 1,E+02 1,E+04 1,E+06 1,E+08 1,E+10

f (Hz)

S(d

B)R

Figure 8.10 Aluminum shield reflection loss.

8.2.1 Electric Field

Shielding against the electric field is usually made with the Faraday cage (Figure8.12). The Faraday cage can be a sphere, rectangle, or any other shape. Whenplaced inside an electric field, it will not absorb the field, but rather produce electricpotential of different polarity along the edge of the cage. This will create an oppositeelectric field, which will result in no electric field inside the cage. The earlier discus-sion proved that the thickness of the cage (shield) is not very important.

If the shield has an aperture, the electric field will penetrate inside the shield,and with a wire in the vicinity of the shield, there will be an induced voltage alongthe wire inside the shield as is shown in Figure 8.13. This figure shows that the elec-tric field intensity falls with the distance from the aperture. It is therefore advisableto place unshielded wires carrying information as far away from the apertures aspossible, or to place the apertures away from the critical areas.


0

50

100

150

200

250

300

350

400

450

1,E+00 1,E+02 1,E+04 1,E+06 1,E+08 1,E+10

f (Hz)

S

S

SA

SR

Figure 8.11 Reflection and absorption loss for an aluminum shield versus frequency.

Chargedmetal wall

Faraday cageE

E = 0

Figure 8.12 Faraday cage.

8.2.2 Magnetic Field

The Faraday cage does not work for the magnetic field. The magnetic field (tangen-tial to the shield) will penetrate the shield that works for the electric field. The atten-uation of the magnetic field can be done with material that has a permeability muchlarger than 1 (µ > 1) as is shown in Figure 8.14. The magnetic field stays in theshield; there will be no magnetic field inside the shield if the permeability of thematerial is great enough. If the shield has an aperture, the magnetic field will pene-trate the shield the same way an electric field penetrates the Faraday shield.

The magnetic shield can also be made with a thin conducting material of smallpermeability for AC. The alternating magnetic field will create the eddy currents,which flow along the shield. These currents create a magnetic field of the opposite


Shield withaperture

E

E 0≠

Chargedmetal wall

Wire

Chargedmetal wall

E

Figure 8.13 Shield with an aperture.

WireB > 0

B = 0

Magnetic shieldwith 1µ >>

H

Figure 8.14 Magnetic shield.

direction, which will cancel out the outer magnetic field. For higher frequencies, thiseffect will be stronger. Such an aluminum shield will be sufficient protection againstthe magnetic field at 50/60 Hz for power lines. It is harder to make the shield forlow-frequency magnetic fields than it is for high frequency magnetic fields.

The magnetic shield should have as few apertures/holes as possible to sustainshield efficiency. All the apertures (doors, windows, ventilation, cable openings,and so forth) will compromise the shield and allow the tangential magnetic fields toenter the shield. If there should be a cable or wire in the vicinity of the aperture,there can be an induced current, which will be carried further inside the shield andinterfere with other equipment (Figure 8.15).

8.3 Equipment Shielding

As can be concluded from the above discussion, it is easier to make a shield againstthe electric field that it is for the magnetic field. Even a very thin metal sheet willprovide good shielding effectiveness, especially at high frequencies.

Generally it is advisable to keep the wires and equipment as far as possible fromthe shield walls and the apertures in it. However, today the industry is constantlytrying to make electronic devices smaller, and big shields are not practical. If theshield has parallel walls, there will be standing waves between them and thus reso-nances. It would be best if the shield would be of an irregular shape, which isimpractical. It is better to use the rectangular than the cubic shape for the shield inorder to lower the number of resonances.

Another problem arises when apertures are to be placed in the shield.

8.3.1 Gasketing

Gaskets are elements that are placed inside the apertures to ensure continuity of theshield (Figure 8.16). This can also include doors, windows, or other apertures. Gas-

8.3 Equipment Shielding 147

Wire

H

Shield

Figure 8.15 Magnetic field entering through aperture in the shield.

kets should be able to withstand environmental conditions such as temperature,salt, moisture, and heat for a long time. After a certain period of time they should bereplaced. If the gasket is made of the same material as the shield, theoretically thecurrents in the shield could flow without interruption. This is hard to achieve due tomechanical constraints.

Gasket types include spring fingers, metal meshes, and conductively wrappedpolymers. Spring fingers (Figure 8.17) are usually made of beryllium copper. Theyare placed on frequently used doors and must be pressed tightly to achieve a goodimpedance match with metal walls. As the frequency rises, the finger size should getsmaller.

Metal meshes consist of elastomer with impregnated metal particles in them.Such a gasket can also function as an environmental seal. Conductively wrappedpolymers are a combination of polymer foam or tube and an outer conductive coat-ing. They are flexible and do not require high contact pressure, which makes themvulnerable to environmental conditions.

Gaskets should be painted with conductive paint only. They have a wide area ofuse, from wireless communications (also in mobile phones) to facilities used for test-ing immunity and emission. Gasket attenuation of an electric field is between 40 dBand 60 dB. The frequency of use is from 10 kHz up to 20 GHz.

8.3.2 PCB Protection

Complete PCB shielding will have a six-sided metal box around it and shielded con-nectors and filters for power and signal cables going in and out. Another possibilityis to have a five-sided metal box attached to the ground plane in several places tocreate a Faraday cage. There will still be the problem of apertures in the groundplane, the connections between ground plane and metal shield, and in the shielditself (ventilation).


Gasket

Shield

Figure 8.16 Gasket in the shield.

Figure 8.17 Spring fingers.

Generally it is better to have as many tracks or striplines as possible instead ofwires and cables. The striplines can be filtered with feed-through filters, or at leastwith ferrite beads or capacitors. Unfiltered signals should not be close to the filteredones. The cables entering the PCB should also at least have ferrite beads if they arenot filtered. Figure 8.18 shows the PCB shielding.

Besides shielding, the design of PCB can improve electromagnetic interferencesuppression. Every cable is a possible antenna. The PCB traces (striplines) do notradiate as much as cables or wires because of much smaller dimensions. However,they might still radiate, especially when combined with cables going out of theshield. With a proper design, this effect can be minimized.

The goal in designing PCB traces is to have a return signal trace close to the onegoing in the opposite direction. In this way, the electromagnetic fields of both traceswill cancel each other out.

Figure 8.19 shows a typical PCB connected to the power cable (this can be a sig-nal cable as well). A PCB with a power cable connected can be seen in Figure8.19(a). The PCB trace line in combination with the power cable can form a struc-ture, which resembles a dipole [Figure 8.19(b)]. The currents flow in the same direc-tion and the radiation is strong and increases with the frequency. This can beprevented with the mains filter (or ferrite beads) on the power cable. On the otherhand, another PCB trace going in the opposite direction will form a structure similarto the transmission line [Figure 8.19(c)]. Here, the currents will cancel each otherout and the total radiation will be small.

The efficiency of the antenna is higher when the antenna is large; cables repre-sent the largest possible antennas, so the currents in the cables are the greatest possi-ble sources of interference. Therefore, the PCB design should be used fortransmission line structures and not dipole structures.

8.3.3 Magnetic Shield

Magnetic shielding is needed in protecting computer hard disks, speakers in audioengineering, power sources, and so forth. Shielding from the magnetic field can be


Signal filters

Ground plane

Ventilation

ShieldMains filter

Figure 8.18 PCB shielding.

done with magnetic materials of high permeability. Materials for magnetic shield-ing differ in saturation (in Gauss) and permeability (Table 8.1).

Other characteristics of magnetic materials include the loss factor, Curie tem-perature, density, and resistivity.

Amumetal is used for high attenuation in a small space. It is available in thethickness from 50 µm to 3 µm. For smaller attenuations, a more economic materiallike UCLS can be used. S1 and L8 are used for ferrite toroids in the EMIsuppression.

The attenuation of a circular shield can be calculated from

Sr

ro

t

= −⎛

⎝⎜

⎞

⎠⎟

µ

41

2

2(8.17)

where µ is the material permeability, and ro and ri are the outer and inner shieldradiuses.


Power cable

Transmissionline

Dipole

I

I

I

I

I

Z

ZI

(b)

(a)

(c)

Figure 8.19 Dipole and transmission line structures in the PCB. (a) PCB structure, (b) dipole, and(c) transmission line.

Table 8.1 Magnetic Material Properties

Material Saturation Permeability r

Amumetal 8,000 400,000

Amunickel 15,000 150,000

ULCS 22,000 4,000

L8 2,550 1,500

J70 2,500 620

M7 2205 160

S1 1625 120


Gooch, J. W., and J. K. Daher, Electromagnetic Shielding and Corrosion Protection for AerospaceVehicles, New York: Springer, 2007.Kaires, R. G., “Stopping Electromagnetic Interference at the Printed Circuit Board,” Conformity,November 2003, pp. 12–21.Moongilan, D., and E. Mitchell, “EMI Gasket Shielding Effectiveness Evaluation Method UsingTransmission Theory,” Proc. IEEE International Symposium on Electromagnetic Compatibility,August 18–22, 2008, pp. 1–6.Pothapragada, P., “Selecting Material for Shielding Enclosures,” Conformity, November 2003,pp. 37–39.Raza, I., “Faraday Cage Enclosures and Reduction of Microprocessor Emissions,” ComplianceEngineering, 2001.Strauss, I., “Shielding Review,” Conformity, April 2004, pp. 24–32.Vasquez, H., et al., “Simple Device for Electromagnetic Interference Shielding Effectiveness Mea-surement,” IEEE EMC Society Newsletter, Winter 2009, pp. 62–68.


C H A P T E R 9

Electrical Grounding and BondingThe electronic equipment’s safety should be dealt with prior to the electromagneticinterference problem. This is why grounding and bonding are important in a propercommunication system design.

For every signal current sent to the load there will be a return current path (orseveral of them). Knowing where return currents flow can help minimize interfer-ence and improve safety. Grounding and bonding are methods of connecting equip-ment or cables to each other or to the Earth in order to ensure safety and currentflow.

Grounding is a procedure in which conductive equipment is connected to theEarth for safety reasons. The conductor that connects equipment to the Earth iscalled the grounding electrode. If there is an unintentional connection between theequipment and the ungrounded conductor, there will be a ground fault. In this case,a ground-fault current may exist from the ground fault to the electrical supplysource and not to the Earth. Not every conductor through which the current flowscan be used for grounding. Grounding consists of the following electrically inter-connected subsystems: the Earth electrode subsytem, the fault protection system,the lightning protection subsystem, and the signal reference subsystem.

All conductive objects or equipment that are not grounded or electrically iso-lated from the ground by nonconductors or gaskets should be bonded. An untreatedisolated conductive object, if charged, can cause static electricity discharge.

Bonding is a process of making a low impedance path for the flow of electriccurrent between two metallic objects. The two conductive surfaces are electricallyconnected to each other to prevent electric potential between metal surfaces, whichcan cause interference or sparking. There are various bonding methods, which willbe discussed in Section 9.4.

To provide a mechanically strong and low impedance path for the current flowand to achieve grounding, different metallic objects must be connected (bonded)together. The bonds should be made in such a way that the junction itself does notdetermine the electric and mechanic properties of the junction. The bond propertiesshould primarily be determined by the metallic objects (members) that are to beconnected.

Grounding and bonding cables should have low resistance and the ability towithstand environmental conditions and the passage of time. Contact with themetal surface must be kept at all times, regardless of paint loss, corrosion, and sur-face contamination. If the resistance between bonded objects and the groundingconductor (i.e., 25Ω) is specified, the bonding can be easily tested with simpleinstruments.

153

9.1 Grounding for Safety

Grounding is a process of connecting metal parts to the Earth for protecting person-nel and facilities, as well as for lightning and electrostatic discharge protection. Aperson can touch the metal enclosure and experience a shock. Electronic deviceswith AC currents should have grounding incase a current flows between the powersource and enclosure, which is usually the biggest conductor or metal plateavailable.

Lightning protection includes low impedance conductors (low resistance andlow inductive reactance), which are used to prevent arcing between nearby metallicobjects. Grounding must also be performed for protection against electrostatic dis-charge (ESD) from people, furniture, or other objects that can be charged. Thegrounding should have a path back to the ground for the discharge current.Grounding can also be used for signal reference. To have an unchanging referencevoltage, stray currents must be kept away from the reference ground. As long as thesignal reference wire is connected only to one place of another ground, there will beno stray currents. If, however, the signal reference wire is tied to some other groundat two or more points, there will be noise currents causing interference.

Short, wide wires are better for grounding than long, thin wires because oflower inductance. Round wires should be avoided for grounding, because they havethe highest inductance. If the wire is grounded at more than one point, so calledground loops will be created with interference voltages between them. A connectionto the Earth can be through capacitive coupling, accidental contact, and intentionalcontact. The ground is a direct path of low impedance between the Earth and differ-ent communication or electronic equipment.

The fault protection subsystem ensures that personnel are protected from shockhazard. In addition, equipment should be protected from damage resulting fromfaults in the electric system. This is usually done with a green wire placed inside thedevice, which represents the Earth. The fault protection wire should be separatedfrom the signal reference ground, except at the Earth electrode subsystem.

9.1.1 Shock Control

Human resistance is between 1 kΩ and 10 kΩ, depending on the person’s moistureand wetness levels. This means that voltages up to 50-V AC cannot harm humans.However, higher voltages might cause damage to or even kill a person. This is whyelectronic equipment is grounded with a metal shield.

Personnel operating communication equipment should follow safety measureswhen working, repairing, installing, and operating dangerous equipment. Theymust make sure that high-voltage devices have been grounded. Figure 9.1 shows atypical case of a shock hazard. Figure 9.1(a) shows an example of no ground protec-tion. When operating normally, the current will flow only through the intentionalresistance R on the return path. If there is an accidental short circuit to the casing ofthe electronic equipment, the casing will become a shock hazard. If a person touchesthe frame, the current will flow through the connection of the frame and the equip-ment (RS) and then through the person (RP) touching the casing to the ground. Acurrent of 75 mA through the body can be fatal. This can be avoided by adding a

154 Electrical Grounding and Bonding

grounding wire to the equipment casing [Figure 9.1(b)]. The person touching thecasing will be protected from electric shock.

If there is a fault in the grounding, further protection can be ensured by using fil-ters for electromagnetic interference. Capacitors (0.1 µF for 50/60 Hz) or filters canbe placed from hot and neutral wires to the ground wire. If there is a ground fault inthe electronic equipment, but there is additional protection with EMI filters orcapacitors, the person touching the casing will not experience currents exceeding 5mA through his or her body, which is safe.

9.1.2 Fault Protection

Fault protection includes a path with low resistance between the location of thefault and the power source. The low resistance path in the building is provided bythe green wires. If there is contact between the ground wire and energized conduct-ing objects, the fuses will blow and protect the power source from further damage.The danger from the fault (shock) depends on its duration. That is why fast fusesmust be used. The longer the time of fault exposure, the higher the temperature,which can cause a fire.

Faults can occur either as a direct short or as an arc. The cause of direct shortcan be: rodents, water, moisture combined with dirt on insulator surfaces, overload,deterioration from age, and damage from improper installation. The currents in theground can result in the casing having a higher potential than the ground. Theenergy from the fault can result in high temperatures, which can damage the equip-ment or the personnel, or result in a fire.

For single phase AC power distribution (Figure 9.2), the ground conductor(green wire) must be one of the four wires. The other three wires are the two phasehot black and red wires and the neutral white wire. The ground wire will carry cur-rent only if there is a fault. The hot wires are connected to the high sides of the distri-bution transformer secondary. The neutral white wire is grounded at the servicedisconnecting means. The ground green wire is grounded at the supply side of thefirst service disconnect to the Earth electrode and also to the ground terminal at thedistribution transformer. All metal parts should be connected to the green groundwire.

The three phase system is similar to the single phase system. There are threephase conductors, one neutral, and one ground. The ground (safety) wire must be

9.1 Grounding for Safety 155

Accidental shortElectronic device

No ground protection

Ground

220/110 V

0 V

R

R

R

S

IS

ISP

Electronic device

Ground protection

Ground

220/110 V

0 V

R

R

R

S

P

(b)(a)

Figure 9.1 (a, b) Shock hazard.

connected to the Earth electrode both at the supply side of the first service discon-nect of the facility, as well as at the distribution transformer. The neutral wire mustbe grounded at both locations as well. More explanations on ground connectionscan be found in the literature at the end of this chapter.

9.2 Grounding for Voltage Reference Control

Signal loss and hardware malfunction in the communication link between two elec-tronic devices that use the Earth ground as a voltage reference can occur due tounwanted noise in the ground loop. The cause can lie in bad grounding, a lightningstrike, electrostatic discharge, or ground faults.

Signal reference has a common reference for all of the equipment forminimization of currents between the equipment and elimination of the noise volt-ages on signal paths. It can also be a bus or a conductor for an internal circuit refer-ence of the electronic device.

Signal circuits are referenced to the ground for establishing signal return pathsbetween a load and a source, providing fault protection, and controlling electro-static discharge. Grounding at low frequencies depends on the surface, length, resis-tance, inductance, and capacitance of the conductor. At higher (RF) frequencies, theconductor must be considered a transmission line.

The signal reference in equipment can be a single reference plane or a grid, ormore precisely a floating ground, single point ground, or multipoint ground, whichis actually an equipotential plane.

9.2.1 Floating Ground

The floating ground shown in Figure 9.3 is used as signal reference for a number ofelectronic devices in a facility. The floating ground is isolated from the building orfacility ground and conductive objects. The noise currents will not be conductivelycoupled to the signal circuits.

How effective the floating ground will be depends on its isolation from the con-ductors in the vicinity. In large facilities it is hard to achieve and maintain a com-plete floating ground. The problem lies in the electric static, which can occur inisolated signal circuits, especially if placed near voltage power lines. Electric static


Distributiontransformer

Disconnectingmeans Hot

Neutral

115 V

115 V

Ground

230 V

Hot

Figure 9.2 Single phase AC power ground connection.

can also cause sparks or shock. Most electronic equipment is referenced to the Earthground, so the power faults in the signal system can cause electronic devices to riseto dangerous voltage levels relative to other conductive objects in the facility. Inaddition, in the case of lightning, since the conductors are not coupled together thewhole system can have an increase in voltage resulting in the breakdown of insula-tion and arcing. Therefore, the floating ground solution for the signal referencevoltage is not the best possible solution.

9.2.2 Single Point Ground

For signal reference, a better solution than the floating ground is the single pointground (Figure 9.4). In this design, the signal paths are referenced to a single point,which in turn is connected to the ground.

Ideally, each of the electronic devices as well as circuits inside the devices shouldhave separate ground conductors. This requires a very large number of long wires,which is impractical. The above solution is intended for frequencies up to 300 kHz.

The noise coupled in the single point ground is not conductively coupled intothe signal circuitry over signal ground wires. However, at higher frequencies, singlepoint grounds become transmission lines, where the ground is the other side of theline. In addition, every part of the equipment bonded to the transmission line isactually a tuned stub (see Section 3.1). At different frequencies, the single point

9.2 Grounding for Voltage Reference Control 157

Electronic devices

Fault protectionFault protection

Signal reference

Ground

Figure 9.3 Floating ground.

Electronic devices


Signal reference

Ground

Figure 9.4 Single point ground.

ground will have a different impedance appearing either as an inductor or capacitorand therefore not as a ground. Large facilities also require long wires, which can actas antennas. There is also a problem of stray capacitance between the wires. Withall of the above in mind, the single point ground is not recommended forcommunication systems.

9.2.3 Multipoint Ground

A multipoint ground (Figure 9.5) has many conductive paths from the ground tovarious electronic devices. Inside each electronic device, circuits are multiply con-nected to the ground.

Multipoint grounding is effective for high frequency signal circuits. Coaxialcables can be easily interfaced since their outer conductor or shield does not have tobe floated relative to the equipment casing. If the length of conductors is longer thanλ/8 for the highest frequency of use, a multipoint ground will require an additionalequipotential ground plane for effective ground.

Care must be taken to prevent 50/60-Hz power currents, as well as any low fre-quency currents of high amplitudes, flowing through the ground system to be con-ductively coupled to the signal circuits and introduce noise voltages.

9.2.4 Equipotential Plane

Grounding does not always reduce all interference. In some cases it can increaseinterference by providing conductive coupling paths as well as inductive loops—orit can radiate. This can be countered by placing an equipotential plane on the floorbeneath electronic devices being grounded. The equipotential plane is a large con-ducting material with negligible impedance. The equipotential plane can also beplaced above the electronic devices if it is impossible to install it below them. Why isan equipotential plane better than a grounding wire? The characteristic impedanceis a function of L C/ , so if the capacity C increases, the characteristic impedance

decreases. The capacity of a large metallic sheet is much larger than that of a wire. Inaddition, the inductance L is decreased with width, which decreases the characteris-tic impedance even more. An equipotential plane with large dimensions has a very


Electronic devices


Figure 9.5 Multipoint ground.

low characteristic impedance for a wide frequency range; it represents a referenceplane for all the electronic devices bonded to it.

Compared to other grounds, the equipotential plane represents a safer protec-tion for personnel, since there is no need for long grounding wires. If placed belowan antenna, the equipotential plane protects the antenna from radiating cables orwires that might be placed below it.

The equipotential plane can be built with a copper grid placed in a concretefloor, an aluminum (or copper) screen placed under the carpet/floor tile, or on theceiling. The equipotential plane must be bonded to the Earth electrode at severalpoints.

9.3 Bonding for Current Control

Bonding is a procedure that permanently joins two metallic parts to form a lowimpedance connection, which will ensure electrical continuity as well as capacity toconduct any current. Bonding also ensures equal potential between separate con-nections to the ground.

In every communication system there are many interconnections betweenmetallic parts in order to provide lightning and fault protection, reference signals,and power. These connections must be made in such a way as not to change the elec-tric path’s properties. The connection or junction must be strong and durable, havelow impedance, and be resilient to corrosion.

Bonding is used for prevention of static accumulation, lightning, shock, faultcurrent return path, and minimization of RF potentials on casings. Improper bondscan cause load voltage drops or heat, which can result in a fire. In signal paths, badbonds can cause noise and lower the signal level. Bonding can also influence shieldeffectiveness.

Bonding must be done carefully for interference reduction using a lowpass π fil-ter for a power line as shown in Figure 9.6. The interfering high frequency current I1

will reach the ground through the ZB and should not reach the load, ZL. If the casingis not bonded properly to the ground reference plane, the bond impedance, ZB,could be relatively large compared to the reactance, XC, at the interfering frequency,and the interference current, I2, could reach the load and thus lower the filterefficiency.

9.3 Bonding for Current Control 159

U Z

Z

L

B

I1

I2

L

C/2 C/2

Figure 9.6 Improper bonding for a lowpass filter.

9.3.1 Bonding Classes

Table 9.1 gives bonding classes. The class A bond is used for bonding antennainstallations. Radiating elements have to be installed with a ground plane of negligi-ble impedance for operating frequencies of the antenna, but should not interferewith the antenna radiation pattern. The RF currents must have a low impedancepath of a small length. For the coaxial antenna there must be continuity betweenouter conductors or the shield and ground plane of antennas. The class C bondreduces power and voltage losses. This bond type requires low impedance and lowvoltages in joints for assuring adequate power to the user. The class H bond protectsagainst fire and shock to the personnel. It is applied to electronic devices required tocarry the fault current. Bonding resistance for this class must be 0.1Ω or lower. Theclass L bond is used for lightning protection and must be able to endure very highcurrents (200 kA) and magnetic forces. Low inductance of the bond is required. Theclass R bond is applied in cases of RF noise, which is present in a wide frequencyrange. The R bond requires low RF impedance at high frequencies. The bondingresistance must be 5 mΩ or lower. Low inductance is also required. This bond typewill be discussed in more detail below. The class S bond protects against electro-static discharge. The bonding resistance must be 1Ω or lower. All isolated conduc-tors with dimensions greater than 7.5 cm must be bonded with the S bond becausethey can be charged.

9.3.2 Strap Bond for Class R

Isolated metallic elements whose linear dimensions are close to half of the wave-length (λ/2) associated with the operating frequency can act as an antenna andreceive RF signals as well as produce enough voltage to cause discharge to otherelectronic devices or circuits. The strap bond discussed in this section is an indirectbonding type. The inductance L of a thin metal strap is given with

[ ]L ll

w tw t

lH=

+⎛⎝⎜

⎞⎠⎟

+ + +⎛⎝⎜

⎞⎠⎟

⎡⎣⎢

⎤⎦⎥

00022

05 02235. ln . . µ (9.1)

where l is the length, w is the width, and t is the thickness of the strap in centimeters.If the strap is round, the inductance is given as

[ ]L ll

dx H= ⎛

⎝⎜⎞⎠⎟

−⎡⎣⎢

⎤⎦⎥

00024

. ln µ (9.2)


Table 9.1 Bonding Classes

Class Application

A Antenna installation

C Current path return

H Shock hazard

L Lightning protection

R RF potential

S Static charge

where d is the diameter of the strap in cm. For low frequencies x = 0.75 and for highfrequencies x = 1. The inductive reactance of the strap is

X L fLL = =ω π2 (9.3)

where f is the frequency in hertz and L is the inductance in H.If, for example, the round strap has the following characteristics: resistance (l =

20 cm) R = 32.2 · 10−6Ω, length 20 cm, diameter 1 cm2, thickness 0.1 cm, and width6 cm, its inductance will be according to (9.2): L = 0.1453 µH.

The capacitance of the bond can be found from

CSd

= ε (9.4)

where ε is the dielectric constant, d is the distance between mating surfaces, and S isthe area of mating surfaces. The capacitive reactance is

XfCC = 1

2π(9.5)

where f is the frequency and C is the capacitance.If the square area of the bond is 200 cm2 and the bond is covered with a 0.02 cm

layer of nonconductive paint, the bond capacitance will be according to (9.5): C =8,852 pF.

The bond impedance depends on the frequency. The capacitive reactance is highat low frequencies and decreases as the frequency increases. The inductive reactanceincreases with frequency. At a resonant frequency, the impedance will reach itsmaximum, which can be more than a thousand Ωs. The resonant frequency is foundfrom

fLC

r = 1

2π(9.6)

The resonant frequency for the above example is 4.44 MHz. The inductive andcapacitive reactance will be XL = 4.05Ω and XC = 4.05Ω. The impedance at the reso-nance is found from

ZXRr =

2

(9.7)

where X is the inductive or capacitive reactance at the resonant frequency and R isthe resistance of the strap. For the above example, Zr will be 5.1 · 105Ω.Theobtained value is much higher than the required 5 mΩ. That is why bonding strapsmust be checked for their resonance.

9.3 Bonding for Current Control 161

9.3.3 Resistance Requirements

The most important requirement for bonding is the low resistance path between thetwo parts that are to be joined. For the antistatic discharge, even the relatively highresistance of 50 kΩ is sufficient. This value, however, is not sufficiently low enoughfor lightning protection or fault currents. If bonding is used for voltage reference,the necessary resistance will depend on the estimated voltage and current levels.

If the surfaces of the two parts to be joined together are properly cleaned, andthe pressure used for bonding the mating surfaces is continuous, the bonding resis-tance achieved can be as low as 1 mΩ. Any attempts to achieve lower bonding resis-tance than intrinsic resistance of the conductors are unnecessary.

The surfaces have to be cleaned to prevent corrosion. This issue will be dealtwith in Section 9.5. All the bond resistances to the ground equipotential planeshould have the same resistances, which will ensure minimum voltage drops andlower the noise in the system. Low bond resistance at DC does not necessarily meanthat it will stay low at high frequencies. The bond resistance at high frequencies willdepend on path resonances, transmission line effects, stray capacitance, and con-ductor inductance.

9.4 Types of Electrical Bonds

Bonding two metal parts regardless of whether they are intended for lightning pro-tection, Earth electrodes, or mating of the equipment front panels to the equipmentracks can be done with different direct or indirect electrical bonds. The best bond isachieved by welding and brazing. Silver soldering also makes a very good bond.Round wires used as jumpers do not make very good bonds because of high induc-tance. A metal strap has a much lower impedance than a round wire and is a bettersolution. The strap length to width ratio should equal 5 to 1, while the strap widthto thickness should be 10 to 1 or more. If the bond length is equal or above 0.1λ forthe corresponding frequency, the bond will not be effective.

Whatever bonding method is applied, it is important to obtain electric continu-ity and keep the DC resistance and RF resistance as low as possible. Direct bonding(welding, brazing) is the best type and is used if the two members have no relativemovement and the bond will be permanent. If the two members must be separated,indirect bonding (bolting, clamping, straps, or other auxiliary conductors) can alsobe used. Figure 9.7 shows the direct bond of two members, by both butt joint andlap joint. The current flowing between the two members will depend on the resis-


Lap jointButt joint

I

Member 2Member 2

Member 1 Member 1

I

Figure 9.7 Butt and lap joint.

tance of the two conductors and on bond resistance. The bond resistance increasesthe total resistance of the path, and it must be much smaller than the conductors’resistances so that the path resistance depends primarily on the conductorresistances.

Both the direct and indirect bonding methods will determine the bonding resis-tance, which will depend on the type of metal used, surface cleanness, contact pres-sure at the surfaces, and cross-section area of the mating surfaces.

9.4.1 Welding and Brazing

Welding is the best bonding method in view of the electrical properties of the bond.Heat over 2,000°C cleans the metal surfaces from any contamination. The bondresistance is close to 0 due to a very short bond length compared to member lengths.The bond strength is equal to if not stronger than the strength of the members.Intensive heat prevents moisture penetration into the bond, so there is almost nocorrosion. The longevity of the bond depends on the duration of its members.Although welding is expensive, it should be utilized for permanent bonds.

Brazing (Figure 9.8), including silver soldering, is also a metal flow process sim-ilar to welding for permanent bonding. The temperature used in brazing is above800°C, which is above the melting point of the brazing filler metal, but below themelting point of the bond members. The filler metal is used to make contactbetween the two metal members. The brazed bond resistance is close to 0, but sincethe filler metal is different from the bond members, corrosion is more probable thanin welding.

9.4.2 Bolting

In some cases permanent bonds are not desirable. This includes moving equipmentfrom one place to another (e.g., for repairing) or disconnecting the connections.Less permanent bonds are also easier to achieve and are more flexible. One of themost used semipermanent bonds is the bolting connection (Figure 9.9) with bolts,screws, or similar fasteners, which should be able to sustain shock and vibrations.

The bolts provide the necessary pressure between contact surfaces. The primarypurpose of the bolts is not to be conductive; they do not even have to be metallic.The necessary pressure between the mating surfaces, which should be over 8MN/m2, will determine the number of bolts. It is better to use more bolts for largeconnecting surfaces, or even to use rigid backing plates or clamps.

9.4 Types of Electrical Bonds 163

Member 1 Member 2

Brazing filler metal

Figure 9.8 Brazing.

9.4.3 Conductive Adhesive

Conductive adhesive is a direct low resistance bond without the use of heat. Con-ductive adhesive is made of two-component silver-filled epoxy resin, which pro-vides good electrical conductivity. It is used in places where heat might damage theequipment or cause a fire. In combination with bolts, the conductive adhesive low-ers the danger of corrosion, at the same time maintaining high mechanical strength.The problem with conductive adhesive is that it is not easy to disassemble.

9.5 Galvanic (Dissimilar Metal) Corrosion Control

Corrosion is the deterioration of conductive material due to environmental influ-ences. Almost all environments are corrosive. This is especially true for industrialareas. Corrosion raises the required low resistance connection up to the point whereit becomes unusable. Corrosion is actually a chemical process in metals (Figure9.10).

The metal surface will form an anode and a cathode due to impurities in contactthrough the metal. If there is an electrolyte or conducting fluid in the environmentabove the surface of the metal, the circuit will allow the currents to flow from theanode into the cathode and thus make corrosion possible. This will result in oxida-tion (i.e., the transfer of electrons from the metal into the environment/oxidizingagent). The oxidation will cause an electromotive force (EMF) between the metaland oxidation agent. Metal in contact with an oxidizing solution will cause a fixedpotential difference compared to any other metal in the same condition. This set of


Member 1Member 2

BoltBonding area

Figure 9.9 Bolting.

Metal

Anode I

I

Electrolyte Cathode

Figure 9.10 Corrosion.

potentials, or EMF series, depends on the temperature and ion concentration in thesolution, and is given for some metals in Table 9.2.

Table 9.2 shows the relative tendencies of materials to corrode; higher valuesrepresent more of a chance for corrosion, meaning that aluminum is more likely tocorrode than gold. The higher the potential is between two metals, the more dissimi-lar the metals and the more chance for corrosion. The metal with the higher voltagewill be the anode and the metal with the lower voltage will be the cathode. The cur-rent flowing between the two metals will result in metal loss. This phenomenon iscalled galvanic corrosion. Metals in a bond should be of the same material or as sim-ilar as possible (close electrode potentials) to prevent galvanic corrosion.

Corrosion can exist only if current can flow from anode to cathode. If there iswater on the surface, the impurities of the water will support corrosion. Therefore,the metal parts have to be painted (coated), since paint prevents moisture fromreaching the metal and thus provides the electrolytic path for the current betweenthe anode and the cathode.

If bonding is to occur between different materials, the member representing theanode (higher voltage) should always be larger than the one representing the cath-ode (lower voltage) as shown in Figure 9.11. If, for example, a copper strap isbonded to an iron plate, the iron will not corrode much because of the large anodicarea. In reverse, an iron strap in contact with a copper plate will corrode very fastdue to a small anodic area.

It is desirable to cover both the anodic and the cathodic member with paint.Never should only the anode be painted, since this will speed up the corrosion—thesmall breaks in the paint actually become a small anodic area.

Before bonding, the metal members should be cleaned with a wire brush, steelwool, or other means to achieve a bright metal finish. Sometimes chemical cleaning

9.5 Galvanic (Dissimilar Metal) Corrosion Control 165

Cathode

Moisture

Anode

Figure 9.11 Dissimilar junction.

Table 9.2 Standard EMFSeries

MetalElectrodePotential (V)

Aluminum 2.37

Iron 0.440

Tin 0.136

Lead 0.126

Copper −0.337

Silver −0.799

Gold −1.5

will be necessary. It is a good idea to use washers for bonding as well, because theycan be easily replaced in case of corrosion. At the end, a protective coating should beapplied for anticorrosion, and the bonds should be tested periodically to ensure thattheir performance is satisfactory.


Military Handbook, Grounding, Bonding and Shielding For Electronic Equipment and Facilities,MIL-HDBK-419A, December 29, 1987.MIL-STD-1542B(USAF), Electromagnetic Compatibility and Grounding Requirements for SpaceSystem Facilities, November 15, 1991.MIL-STD-188-124B, Grounding, Bonding and Shielding for Common Long Haul/Tactical Com-munication Systems Including Ground Based Communications—Electronics Facilities andEquipments, February 1, 1992.MIL-STD-1310G, Standard Practice for Shipboard Bonding, Grounding, and Other Techniquesfor Electromagnetic Compatibility and Safety, June 28, 1996.NASA-STD-P023, Electrical Bonding for NASA Launch Vehicles, Spacecraft, Payloads, andFlight Equipment, August 17, 2001.NSTS 37330, Space Shuttle Bonding, Electrical, and Lightning Specifications, December 2, 1999.


C H A P T E R 1 0

Emissions and Susceptibility—Radiatedand Conducted

10.1 Control of Emissions and Susceptibility—Radiated andConducted

Electromagnetic interference (EMI) can be regarded as a sort of environment pollu-tion with consequences similar to those of poison chemicals, automobile gasexhaust, and so forth. The electromagnetic spectrum is a natural source that hasbeen devastated to a large extent in the last hundred years. The spectrum is full,which is why new technologies operate on increasingly higher frequencies. Modernlife depends on the systems using the electromagnetic spectrum, and its protection isa priority. Uncontrolled electromagnetic radiation can lead to equipment damage,loss of money, injuries, and even death. Electromagnetic interference introducesunwanted voltages and currents into the equipment—the victim. This can lead toaudio noise in radio receivers, as well as snow or picture loss on TV receivers. Whenthe victims are communication links or computer systems operating industry facili-ties, the damage is even greater. Interference can find its way to the victim in twoways: through cables (conducting) and by electromagnetic radiation.

10.1.1 Sources of Electromagnetic Interference

Every electric or electronic device that changes voltage or current can be an EMIsource. Electric devices can introduce interference by electromagnetic radiation orthrough the cables. Electric shavers and dish washers can cause interference on TVsets, not only directly via radiation but by power cables as well (Figure 10.1).

Generally, a faster change of voltage or current will result in a wider interfer-ence spectrum. Similarly, a higher voltage or current level will cause a higher con-ducted or radiated interference level. Therefore, electric machines generating highvoltages and currents with a fast rise will be strong EMI sources. Table 10.1 showsthe EMI sources regarding usage.

EMI sources can also be divided into continuous and transient. Radar, althoughan impulse system, has a wide but stable RF spectrum and is considered a continu-ous EMI source. The radiation from continuous EMI sources can best be analyzedwith the spectrum analyzer. On the other hand, lightning or nuclear electromag-netic pulse occurs unpredictably and is called transient radiation. Transient signalsare very short and have a wide spectrum. They are analyzed much easier in the time

167

domain. It is very hard to monitor the fast transient spectrum. Table 10.2 showssome continuous and transient EMI sources.

An EMI problem exists only when the EMI source can exchange electromag-netic energy with its victim. The general EMI problem is shown in Figure 10.2. Thecoupling between the EMI source and EMI receiver (victim) can be either radiatedor conducted.

The energy exchange between the EMI source and victim equipment can hap-pen through the metal guides of the victim. If the primary coupling is radiated, andthe EMI receiver has an input cable (which is not connected to the EMI source), thecurrents in the cable can appear and go directly to the victim, as shown in Figure10.3, even if the victim is protected from radiated interference with a shield.

168 Emissions and Susceptibility—Radiated and Conducted

Figure 10.1 EMI from an electric shaver and dish washer to the TV receiver.

Table 10.1 EMI Sources Regarding Usage

Manmade Sources

Communication Power LinesVehicleSystems

Engines andTools

Industrial/Commercial

Customers ESD

Broadcasting Generators Automobiles Compressors Dielectric heaters Microwaves

Radar Converters Ignitions Saws Air conditioners Ovens

Walkie-talkies Amplifiers Mobiledevices

RF heaters Fluorescent lights PCs

Amateur radio Transmissions Ultrasoundcleaners

Lasers Blenders

ELF/VLFnavigation

Line noise Weldingmachines

Neon signs Vacuumcleaners

Mobile devices Cranes Medical devices Hair dryers

Remote controls Refrigerators

Electricshavers

Natural Sources

Static Noise Atmosphere Solar Noise Lightning Space Radio Noise

Similarly, if the primary coupling is conducted, and the victim has a good filterconnected to the cable entrance, the currents in the cable can still radiate EMenergy, which can be coupled into the victim if it does not have an appropriateshielding (Figure 10.4).

These two examples show that both conducted and radiated interference (directand indirect) must be considered to prevent EMI energy going from the EMI sourceto the victim equipment. It is, therefore, necessary to shield and filter the equipmentsimultaneously.

10.1 Control of Emissions and Susceptibility—Radiated and Conducted 169

Radiated

EMI receiver(victim)

Conducted

CouplingTransmitter Receiver

EMI source

Figure 10.2 EMI problem.

Table 10.2 Sources of Continuous and Transient EMI

Continuous EMI Sources—Frequency Domain Analysis

Transient EMI Sources—Time Domain Analysis

Broadcasting Lightning

Ship radar (pulsed output) Nuclear electromagnetic pulse

Electric machine noise Power line sparks

Fixed and mobile communications Relays and switches

PCs, printers Welding machines

Solar and space radio noise Human electrostatic discharge (ESD)

I

EMI source

Radiated EMI

The cable works as unwantedantenna to other equipmentRF EMI current into the cable

Conducted RF currents

Victim inputconnector

Shield protectsradiated coupling

Figure 10.3 Conducted EMI from radiated EMI.

Some examples of intentional and unintentional EMI receivers are in Table10.3.

EMC is divided according to Figure 10.5. The main activities are conductedemission, conducted susceptibility, radiated emission, and radiated susceptibility.Other activities include transients [electrostatic discharge (ESD), nuclear electro-magnetic impulse, and lightning]. Electroexplosive devices and nonionizing effectsof electromagnetic radiation fall into general activities of electromagnetic compati-bility and are beyond the scope of this book.

Emission and susceptibility are the two most common tests of electromagneticinterference (EMI) that a device or piece of equipment should undergo, whether itoperates on low or RF frequencies. Emission is the unintentional or undesired exit-ing of potentially interfering electromagnetic energy from electrical or electronicsources (devices, modules, equipment, and systems). Emission can also be inten-tional, such as from a transmitter, although it is not intended to cause interferenceto other devices or equipment. Emission can be conducted (carried along cables) orradiated (via propagation). Conducted emission (CE) is the potential EMI that isgenerated inside the equipment and is carried out of the equipment over I/O lines,control leads, or power mains. Radiated emission (RE) is the potential EMI that


I

Good filter preventsconducted EMI

Strong conducted EMI

EMI source

RF current source

Radiation from the cablereaches PCB

Circuits directly exposed toradiated EMI

Weak or no shield

Victim

PCB

Figure 10.4 Radiated EMI from conducted EMI.

Table 10.3 EMI Receivers

Intentional Receivers Unintentional Receivers

Radio receivers Airplane control systems

TV receivers Military systems, guided missiles

Mobile phone receivers Ship electronic systems

Microwave relay systems Computer equipment

Air system receivers Signalization systems

Navigation Pacemakers

Radar Explosives

radiates from escape-coupling paths such as cables, leaky apertures, or inadequatelyshielded housings.

Susceptibility is the characteristic of electronic equipment that permits undesir-able responses when subjected to electromagnetic energy. It is sometimes also calledimmunity. There are two types of susceptibility: conducted and radiated. Con-ducted susceptibility (CS) is EMI that couples from the outside of a piece of equip-ment to the inside over conductors (I/O cables, control and signal leads, or powermains). Radiated susceptibility (RS) is the undesired potential EMI that is radiatedinto a piece of equipment or system from a hostile outside electromagnetic source.

10.1.2 Test Requirements for Emission and Susceptibility

Table 10.4 indicates the military and commercial tests that are required for the testsample. Regardless of the EMC test standard, the product must be set up in a con-trolled environment. This includes providing the test sample with standardizedpower to compare results from one lab to another. Commercial equipment is some-times used in a military environment. In some cases it is possible to compare mili-tary and commercial standards, and in other cases it is not. The reason is thedifferent frequency bands specified for different tests, as well as injection methodsused.

Commercial standards such as EN 50081, EN 50082, IEC 60533, and IEC 945can be compared with military standard Mil-Std-461 to some extent.

For conducted emission both military and commercial standards use the LineImpedance Stabilization Network (LISN) for injection of interference. The military


Nuclear EMP50 kV/m RS (10/400 ms)100 A

Cable interferencelightning10 kV transientsharmonic distortions

Human ESD15 kV transientsdirect and indirect

Radiated emission

10 Hz–40 GHzE field 14 kHz–40 GHzH field 10 Hz–30 MHzlevel to 110 dB V/m−

10 Hz–40 GHz1–200 V/m (CW)

Cables20 Hz–100 MHzantenna100 MHz–40 GHzlevels to 120 dB V/m−

20 Hz–400 MHz20 dBm VA−

Radiated sensitivity Conducted sensitivity Conducted sensitivity

Radiated EMI Transients Conducted EMI

Electroexplosivedevice safety

Nonionizing effects ofelectromagnetic radiation

EMCactivities

Figure 10.5 EMC activities.

standard (Mil-Std-461E) measures the current in the frequency range from 30 Hz to10 kHz. In the frequency range from 10 kHz to 10 MHz, the voltage is measured.However, the commercial standards (IEC 60533 and EN 50081-2) measure onlyvoltages between 150 kHz and 30 MHz.

For radiated emission, in the harmonized commercial standards, only the elec-tric field is measured. Commercial standards require an open test site environment(OATS), while military standards require a shielded room. The cage effect is respon-sible for a deviation to open site measurement results of maximum of 6 dB. Themeasuring distance in the commercial standards is 30m, and sometimes 10m—onlyin some cases is 3m allowed. However, IEC 60533 has a distance of 3m in all cases.Mil-Std 461 has a measuring distance of only 1m, which can sometimes be a prob-lem because the antenna is placed in near field. Most commercial standards have ameasuring range from 30 MHz to 1 GHz, with the exception of IEC 60533 having astarting frequency of 150 kHz. Mil-Std 461 has a frequency range from 10 KHz to18 GHz.

For conducted susceptibility commercial and military standards are not reallycomparable because of different injection methods.

For radiated susceptibility the frequency range of commercial standards is from30 MHz to 1 GHz (80 MHz to 2 GHz in IEC 60533), while military standards coverthe frequency range from 10 kHz to 40 GHz. While military standards usenonmodulated test signals, commercial tests use a modulation of 80% with 1 kHz.

In some cases like with electromagnetic pulse (EMP) where equipment shouldwithstand 50 kV/m, only military tests exist.


Table 10.4 Commercial and Military Tests of Emissions and Susceptibility—Radiated and Conducted

Requirement Commercial Military

CE Power Line (30 Hz to 10 kHz + Harmonics) Yes Yes

CE Power Line (fluctuations) Yes No

CE Power Line (10 kHz/150 kHz to 10 MHz/30 MHz) Yes Yes

CE Antenna (10 kHz to 40 GHz) No Yes

CS Power Line (30 Hz to 150 kHz) Yes Yes

CS Structure CM (60 Hz to 100 kHz) No Yes

CS Bulk Cable (10 kHz/150 kHz to 200 MHz/230 MHz) Yes Yes

CS Bulk Cable (impulse) Yes Yes

CE Cables P/S (damped sine 100 kHz to 100 MHz) No Yes

RE Magnetic Field (30 Hz to 100 kHz) No Yes

RE Electric Field (10 kHz/30 MHz to 18 GHz/40 GHz) Yes Yes

RE Antenna (10 kHz to 40 GHz) No Yes

RS Magnetic Field (30 Hz to 100 kHz) No Yes

RS Electric Field (10 kHz/26 MHz to 40 GHz/1 GHz) Yes Yes

RS Transient EM Field (impulse) No Yes

RS ESD (up to 8 kV) Yes NoLegend: CE: conducted emission; CS: conducted susceptibility; RE: radiated emission; RS: radiatedsusceptibility.

10.1.3 Standard Organizations

Today’s products must conform to regulations and standards of both private and gov-ernment regulatory agencies. Their number is increasing, and updates are constantlybeing made. There are several international standard agencies like ISO, IEEE, IEC, ITU,and so forth. Each country has its own laws and standard organizations. Accreditedtesting laboratories issue certifications for products.

A standard is a document established by a consensus and approved by a recog-nized body, which provides (for common and repeated use) rules, guidelines, orcharacteristics for activities or their results aimed at the achievement of the opti-mum degree of order in a given context. Standards cover several disciplines, dealingwith all technical, economic, and social aspects of human activity and covering allbasic disciplines such as language, mathematics, physics, and so forth. Standardsare developed by technical committees. There are several interested parties such as:producers, users, laboratories, public authorities, and consumers. Standards arebased on actual experience and lead to material results in practice (products—bothgoods and services, test methods, and so forth). They are a compromise between thestate of the art and the economic constraints of the time. Standards are documentsthat are recognized as valid nationally, regionally, or internationally; they arereviewed periodically and evolve with technological and social progress. Standardsare available to everyone, and can be consulted and purchased without restriction.Generally, standards are not mandatory, but voluntary. In certain cases their imple-mentation may be obligatory (e.g., safety requirements, electrical installations, pub-lic contracts, and so forth). Standards are used more and more by jurisprudence. Forthe user, the standard is a factor for production rationalization, which makes it pos-sible to master technical characteristics of products, satisfy the customer, validatethe manufacturing methods, increase productivity, and give operators andinstallation technicians a feeling of security. There are four major types ofstandards:

1. Fundamental standards concerning terminology, metrology, conventions,signs and symbols, and so forth;

2. Test methods and analysis standards, which measure characteristics;3. Standards defining product characteristics (product standard), specification

standards (service activities standard), and standards for performancethresholds to be reached (fitness for use, interface and interchange ability,health, safety, environmental protection, standard contracts, documenta-tion accompanying products or services);

4. Organization standards dealing with the description of functions of thecompany and their mutual relationships, as well as the modeling of activities(quality management and assurance, maintenance, value analysis, logistics,quality management, project or systems management, productionmanagement).

A national standard is programmed and studied under the authority of thenational standards body, which publishes it. It is therefore protected, as early as atthe draft standard stage, by a copyright belonging to the national body. Interna-tional standards are protected by a copyright of the international standards body


(ISO, IEC). The exploitation right of this copyright is automatically transferred tothe national standard bodies that are members of ISO or IEC, for the purpose of cre-ating national standards. The national standards body is obliged to take all appro-priate measures to protect the intellectual property of the ISO and IEC on nationalterritory.

The International Organization for Standardization (ISO), founded in 1947, isa worldwide federation of national standards bodies currently comprised of over125 members—one per country. The mission of the ISO is to encourage the devel-opment of standardization and related activities in the world in order to facilitateinternational exchanges of goods. Its work concerns all of the fields of standardiza-tion except for electrical and electronic engineering standards, which fall within thescope of IEC. ISO counts over 2,800 technical work bodies (technical committees,subcommittees, working groups, and ad hoc groups). To date, ISO has publishedover 16,000 international standards.

The International Electrotechnical Commission (IEC) was founded in 1906,and is responsible for international standardization in the fields of electricity, elec-tronics, and related technologies. It deals with all electrotechnologies including elec-tronics, magnetism and electromagnetism, electroaccoustics, telecommunication,energy production and distribution, as well as associated general disciplines such asterminology and symbols, measurement and performance, dependability, designand development, safety, and environment. The IEC currently has over 50 members(national committees), one for each country, which are required to be fully repre-sentative of all electrotechnical interests in the country concerned. National com-mittees are largely supported by the industry and are recognized by their respectivegovernments. The IEC has published over 10,000 standards. Both the ISO and IEChave their central offices in Geneva, Switzerland, and operate according to similarrules. The incorporation of ISO and/or IEC standards into national collections isvoluntary—it can be complete or partial.

The birth of the International Telecommunication Union (ITU) can be tracedback to 1865. A specialized agency of the United Nations since 1947, ITU member-ship currently includes some 180 member states and over 400 sector members. ITUinternational recommendations are developed in the fields of both telecommunica-tions and radiocommunications. ITU headquarters are located in Geneva,Switzerland.

The Institute of Electrical and Electronics Engineers (IEEE) is the world’s largesttechnical professional society. It was founded in 1884; today it has over 380,000members in more than 150 countries and has created about 2,000 standards. TheIEEE Standards Association (IEEE-SA) is the newly founded organization underwhich all IEEE Standards Activities and programs will be carried out.

In the United States there are:

• ANSI—The American National Standards Institute was founded in1918. It is the official U.S. representative to the International Organizationfor Standardization (ISO) and, via the U.S. National Committee, the Interna-tional Electrotechnical Commission (IEC). ANSI is also a member of the Inter-national Accreditation Forum (IAF).


• FCC—The Federal Communications Commission is an independent UnitedStates’ government agency. The FCC was established by the CommunicationsAct of 1934 and is in charge of regulating interstate and international commu-nications via radio, television, wire, satellite, and cable. The FCC’s jurisdic-tion covers the 50 U.S. states, the District of Columbia, and U.S. possessions.

• NIST—The National Institute of Standards and Technology, founded in1901, is a nonregulatory federal agency within the U.S. Commerce Depart-ment, which develops and promotes measurement, standards, and technologyto enhance productivity, enable trade, and improve the quality of life.

In the European Union there also several standard organizations:

• CEN—The Comité Européen de Normalisation (European Committee forStandardization) was founded in 1961. It draws up European standards andconsists of 27 European standards’ institutes. The CEN has witnessed strongdevelopment with the construction of the European Union. Its headquartersare located in Brussels, Belgium. A technical board is in charge of coordina-tion, planning, and programming of the work conducted within the workbodies (technical committees, subcommittees, working groups); the secretari-ats of which are decentralized in the different EU member states. CEN, whichcounts over 250 technical committees, has published several thousand docu-ments.

• CENELEC—Comité Européen de Normalisation Électrotechnique (Euro-pean Committee for Electrotechnical Standardization) was founded in 1959and is located in Brussels, Belgium. CENELEC fulfils the same functions asCEN within the electrotechnical sector.

• ETSI—The European Telecommunications Standards Institute, developsEuropean standards in the telecommunications field (ETS, European TelecomStandard). Its headquarters are at Sophia Antipolis, France. ETSI has 400members (administrations, operators, research bodies, industrialists, users)representing over 30 countries (EU, Eastern Europe).

• ECMA—The European Association for Standardizing Information and Com-munication Systems is an international, Europe-based industry associationfounded in 1961 and dedicated to the standardization of information andcommunication systems. ECMA standards and technical reports are madeavailable to all interested persons or organizations, free of charge and copy-right, and can be obtained in printed form.

• EBU—The European Broadcasting Union was created in 1950, initially withthe aim of solving technical and legal problems and then to develop news andprogram exchanges. The result is that today the EBU assists its members in allareas of broadcasting, briefs them on developments in the audio-visual sector,provides advice and defends their interests with international bodies. Head-quartered in Geneva, Switzerland, the EBU is the world’s largest professionalassociation of national broadcasters. Following a merger with the EBU onJanuary 1, 1993, the International Radio and Television Organization(OIRT)—the former association of Socialist Bloc Broadcasters—expanded


the EBU to 75 active members from 56 countries in and around Europe, and45 associate members around the world.

• CEPT—The Conference Européen des Administrations des Postes et desTélécommunications [European Post, Telephone, and Telegraph Agencies(PTT)] recommends communication specifications to the International Tele-communication Union Standardization Sector (ITU-T).

In South America there are:

• COPANT—The Pan American Standards Commission is a civil, nonprofitassociation with complete operational autonomy. The basic objectives ofCOPANT are to promote the development of technical standardization andrelated activities in its member countries with the aim of promoting the indus-trial, scientific, and technological development for the benefit of an exchangeof goods and provision of services, while facilitating cooperation in the intel-lectual, scientific, and social fields. The commission coordinates the activitiesof all institutes of standardization in Latin American countries and developsall types of product standards, standardized test methods, terminology, andrelated matters. COPANT headquarters are in Buenos Aires, Argentina.

• MERCOSUR—The Common Market of the South (Portuguese acronymMERCOSUL), is a common market made up of the economies of Argentina,Brazil, Paraguay, and Uruguay. Its principal objectives are to improve theeconomies of its member countries by making them more efficient and com-petitive, and by enlarging their markets and accelerating their economic devel-opment by means of a more efficient use of available resources Furtherobjectives are to preserve the environment, improve communications, coordi-nate macroeconomic policies, and harmonize the different sectors of SouthAmerican economies. MERCOSUR’s permanent headquarters are in the cityof Montevideo, Uruguay.

Each national standards body manages its own collection of standards and hasaccess to the collections of other institutes. The collections can be either free infor-mation tools or services for identifying standards or announcing new standards.This can include catalogs, newsletters, Web servers, or chargeable services provid-ing access to the normative texts in different forms (subscription, hardcopy form,CD-ROM).

National members of the ISO and IEC maintain links to related national organi-zations and, when applicable, to national standards–related networks. Informationabout standards can also be found in the ISO/IEC Directory of International Stan-dardizing Bodies. Normally, information on standardization and certification sys-tems, the identification of information concerning standards, and products andservices is free. The publications of the standards bodies (standards, handbooks,hardcopy catalogues) are chargeable, and each body has its own tariffs.

Every country has its national body for issuing its own standards or they useinternational ones.


10.2 Commercial Requirements

The United States, European Union, Japan, Canada, Australia, and other countrieshave set up requirements for the emission and susceptibility of commercial equip-ment. These requirements are included in the standards given below. The FCC (Fed-eral Communications Commission) requires manufacturers of most types ofelectronic products to test the emission specifications. The requirements are speci-fied in the Code of Federal Regulations. Certain types of equipment require specialtesting by FCC.

The IEC 60533 is a standard applicable for electromagnetic compatibility ofelectrical and electronic installations in merchant ships. Electrical installations ofships with electric and/or electronic systems need to operate under a wide range ofenvironmental conditions. The control of undesired electromagnetic emissionensures that No other device on board is influenced by the equipment under test.

On the other hand, the equipment needs to function without degradation in anormal electromagnetic environment (immunity). Special risks (e.g., lightningstrikes), transients from the operation of circuit breakers, and electromagnetic radi-ation from radio transmitters are also covered.

This standard also gives guidelines and recommendations on the measures toachieve EMC in electrical and electronic installations of the following equipmentgroups:

• Group A: radio communication and navigation equipment;• Group B: power generation and conversion equipment;• Group C: equipment operating with pulsed power;• Group D: switchgear and control systems;• Group E: intercommunication and signal processing equipment;• Group F: nonelectrical items and equipment;• Group G: integrated systems.

The IEC 945 (now IEC 60945) is an interference standard for navigation equip-ment installed in a ship’s environment. IEC 945 was originally produced to providetest methods and, where appropriate, limit values for electronic navigational aids.

The two European EN standards are generic standards for equipment installedin an industrial environment and do not deal with product standards. EN 50081 isan emission standard and EN 80082 an immunity standard. EN 50081 and EN50082 provide limits for emission and immunity of electromagnetic disturbancesfrom electrical and electronic apparatus (for which there are No dedicated prod-uct-family standards) intended for use in the industrial environment.

“EMC Testing and Measurement Techniques Section 3: Radiated, Radio Fre-quency, EM Field Immunity Test” (IEC 61000-4-3) has been used for many years asthe basic test standard for radiated electromagnetic field immunity testing in orderto fulfill one of many EU requirements for the CE mark. The IEC 61000-4-3 stan-dard is usually used together with a product standard that will specify this and othertest standards, detailing the requirements the product must meet. The aim of thisstandard is to establish a common reference for immunity to radio frequency (RF)

10.2 Commercial Requirements 177

radiation caused by any source. Electronic products need to be designed and testedto have immunity from these sources.

“EMC Testing and Measurement Techniques Section 6: Immunity to Con-ducted Disturbances by Radio Frequency Fields” (IEC 61000-4) relates to the con-ducted immunity requirements of electrical and electronic equipment toelectromagnetic disturbances from intended radio-frequency (RF) transmitters inthe frequency range of 9 kHz to 80 MHz. Equipment without at least one conduct-ing cable (such as a mains supply, signal line, or earth connection), which could cou-ple the equipment to the disturbing RF fields, is excluded.

The objective of this standard is to establish a common reference for evaluatingthe functional immunity of electrical and electronic equipment when subjected toconducted disturbances induced by radio-frequency fields. The test method docu-mented in this part of IEC 61000 describes a consistent method to assess the immu-nity of a piece of equipment or system against a defined phenomenon.

10.3 Military Requirements

The military environment is different from the commercial in the areas of radartransmissions, communications in a wide frequency range, electromagnetic pulses,and inner deck situations because of high equipment density.

Until the publication of 461E, MIL-STD-461 documented the test limits andlevels while MIL-STD-462 specified the test methods and procedures that were tobe used for conducting the tests. The E version of this standard combined both stan-dards into one document. Previous versions were A, B, C, and D. Table 10.5 showsthe tests of Mil-Std 461E.

Mil-Std 461 deals with two basic areas of electromagnetic effects: conductedand radiated. Each area is represented in two different modes—emission and sus-ceptibility—which include conducted emissions, conducted susceptibility, radiatedemissions, and radiated susceptibility.

Different devices and components such as ships, weapons, aircraft, ground andsupport equipment, and electrical and electronic systems that are used in a militaryor aerospace application are subject to meeting the requirements established inMil-Std 461. Each branch of the armed services—Army, Navy, Air Force, andNASA—have identified specific requirements that are applicable to the specificneeds and applications. Not all tests are required for every application.

Mil Std 461 uses shielded enclosures for testing. The shielded enclosure shouldbe large enough to hold the equipment being tested and be equipped to handle therequirements needed to perform simulation tests. Mil-Std 461E is free of charge andcan be downloaded for free on the Internet.

10.3.1 Specific Conducted Emissions Requirements Mil-Std 461E

10.3.1.1 CE101—Power Leads (30 Hz to 10 kHz)

This level of low frequency testing is most applicable to the following platforms:submarines, Army, and Navy aircraft.


10.3.1.2 CE102—Power Leads (10 kHz to 10 MHz)

This is a similar test to CE101, but for higher frequencies. Additionally, this test hasa much wider application and is required on the following platforms, systems, andsubsystems: submarines; Army and Navy aircraft; air force aircraft; space systems;Army, Navy, and Air Force ground systems; and equipment surface ships.

10.3.1.3 CE106—Antenna Terminals (10 kHz to 40 GHz)

The CE106 testing is applicable to most antenna terminals, receivers, transmitters,and amplifiers, with the exception of equipment designed with the antenna perma-nently mounted to the equipment undergoing testing. CE106 has been recentlymodified to include amplifiers and is widely applicable and required on: subma-rines; Army and Navy aircraft; space systems; Army, Navy, and Air Force groundsystems; and equipment surface ships.

10.3.2 Specific Conducted Susceptibility Requirements Mil-Std 461E

10.3.2.1 CS101—Power Leads (30 Hz to 150 kHz)

The CS101 test is applicable to equipment and subsystems of AC and DC powerleads, excluding returns. If EUT is operating under DC power, testing is requiredonly between 30 Hz and 150 kHz. If EUT is operating under AC power, testing is

10.3 Military Requirements 179

Table 10.5 Tests of Mil-Std 461E

MIL-STD-461E

Specific Conducted Emissions Requirements

CE101 Power Leads (30 Hz to 10 kHz)

CE102 Power Leads (10 kHz to 10 MHz)

CE106 Antenna Terminals (10 kHz to 40 GHz)Specific Conducted Susceptibility Requirements

CS101 Power Leads (30 Hz to 150 kHz)

CS103 Antenna Port-Intermodulation (15 kHz to 10 GHz)

CS104 Antenna Port Rejection of Undesired Signals (30 Hz to 20 GHz)

CS105 Antenna Port-Cross Modulation (30 Hz to 20 GHz)

CS109 Structure Current-Spike (60 Hz to 100 kHz)

CS114 Bulk Cable Injection (10 kHz to 200 MHz)

CS115 Bulk Cable Injection-Impulse Excitation

CS116 Damped Sinusoidal Transients (10 kHz to 100 MHz)Radiated Emissions Requirements Mil-Std 461E

RE101 Magnetic Field (30 Hz to 100 kHz)

RE102 Electric Field (10 kHz to 18 GHz)

RE103 Antenna Spurious and Harmonic Outputs (10 kHz to 40 GHz)Radiated Susceptibility Requirements Mil-Std 461E

RS101 Magnetic Field (30 kHz to 100 kHz)

RS103 Electric Field (2 MHz to 40 GHz)

required to start at the second harmonic of the power frequency, and extends up to150 kHz. These requirements are applicable on the following platforms: subma-rines; surface ships; space equipment and systems; Army, Navy, and Air Force air-craft; army, navy, and air force ground systems.

10.3.2.2 CS103—Antenna Port-Intermodulation (15 kHz to 10 GHz)

The CS103 test is required for communications receivers, RF amplifiers, transceiv-ers, radar receivers, acoustic receivers, and electronic ware receivers. The aim of thistest is to control the response of the antenna connect receiving subsystems toin-band signals resulting from potential intermodulation products of two signalsoutside the intentional passband of the subsystems produced by the nonlinearity inthe subsystem. The EUT should not exhibit any modulation beyond specified toler-ances. The CS103 test is applicable to the following platforms: submarines; surfaceships; space equipment and systems; Army, Navy, and Air Force aircraft; and Army,Navy, and Air Force ground systems.

10.3.2.3 CS104—Antenna Port Rejection of Undesired Signals (30 Hz to 20 GHz)

The CS104 test controls the response of antenna connected receiving devices or sub-systems to signals outside the intentional passband produced by nonlinearity. Theapplications are similar to CS103, and are required for communications receivers,RF amplifiers, transceivers, radar receivers, acoustic receivers, and electronic warereceivers. CS104 can be applied to the following platforms: submarines; surfaceships; space equipment and systems; Army, Navy, and Air Force aircraft; Army,Navy, and Air Force ground systems.

10.3.2.4 CS105—Antenna Port-Cross Modulation (30 Hz to 20 GHz)

The CS105 test controls the response of antenna connected receiving subsystems tomodulation being transferred from an out-of-band signal to an in-band signal. Thiscan be caused by a strong out-of-band signal near the operating frequency of thereceiver that modulates the gain in the front end of the receiver and adds amplitudevarying in formation to the desired signals. The applications are similar to CS103and CS104 and are required for communications receivers, RF amplifiers, trans-ceivers, radar receivers, acoustic receivers, and electronic ware receivers. CS105 isapplicable to the following platforms: submarines; surface ships; space equipmentand systems; Army, Navy, and Air Force aircraft; Army, Navy, and Air Forceground systems.

10.3.2.5 CS109—Structure Current-Spike (60 Hz to 100 kHz)

CS109 has limited applications and is intended to simulate a spike in voltage,according to which the EUT must continue to operate without malfunction, degra-dation of performance, or deviation, even beyond the accepted tolerance range.Most applications are within the area of submarines.


10.3.2.6 CS114—Bulk Cable Injection (10 kHz to 200 MHz)

CS114 is widely applied to all interconnecting cables, including power cables.According to CS114, the EUT should not malfunction when subjected to a bulkinjection probe drive level. CS114 is a requirement on the following platforms: sub-marines; surface ships; space equipment and systems; Army, Navy, and Air Forceaircraft; and Army, Navy, and Air Force ground systems.

10.3.2.7 CS115—Bulk Cable Injection-Impulse Excitation

The CS115 test serves to protect equipment from fast rise and fall time transientsthat may be present due to platform switching and external transient environments,such as an electromagnetic pulse (EMP). The test will verify the ability of the EUT towithstand the impulse signals that are coupled onto the EUT associated cabling.This test replaces the old chattering relay test, referenced in Mil Std 461 C-RS 106.CS115 is applicable on the following platforms: submarines; surface ships; spaceequipment and systems; Army, Navy, and Air Force aircraft; and Army, Navy, andAir Force ground systems.

10.3.2.8 CS116—Damped Sinusoidal Transients (10 kHz to 100 MHz)

The concept of CS116 is to simulate electrical current and voltage waveforms occur-ring in platforms from natural resonances. CS116 is applicable to all electricalcables interfacing with the EUT and individually on each power lead. The testingshould verify the EUT’s ability to withstand damped sinusoidal transients coupledonto the EUT associated cables and power leads. Power returns and neutrals neednot be tested individually. The CS116 test is applicable to the following platforms:submarines; surface ships; space equipment and systems; Army, Navy, and AirForce aircraft; and Army, Navy, and Air Force ground systems.

10.3.3 Radiated Emissions Requirements Mil-Std 461E

10.3.3.1 RE101—Magnetic Field (30 Hz to 100 kHz)

The RE101 testing requirement is intended to control magnetic fields for applica-tions in which the equipment presents an installation potentially sensitive to mag-netic induction to lower frequencies. This test verifies that the magnetic fieldemissions from the EUT and its associated electrical interfaces do not exceed speci-fied requirements. A common example for this test is a tuned receiver that operateswithin the frequency range of the test parameters. The applications for this test are:submarines, surface ships, and Army and Navy aircraft.

10.3.3.2 RE102—Electric Field 10 kHz to 18 GHz

The RE102 test is one of the most widely required tests for electrical and electronicequipment. Its aim is to protect sensitive receivers from interference coupledthrough antennas associated with a receiver, and to verify that the electric fieldemissions from the EUT and its associated cabling do not exceed specified limits.The requirements vary depending on platform and application. The platforms


required to meet these test parameters are: submarines; surface ships; space equip-ment and systems; Army, Navy, and Air Force aircraft; and Army, Navy, and AirForce ground systems.

10.3.3.3 RE103—Antenna Spurious and Harmonic Outputs 10 kHz to 40 GHz

The RE103 test is used to confirm that radiated spurious and harmonic emissionsfrom transmitters do not exceed the specified limit requirements. RE103 has differ-ent starting frequencies depending on the actual operating frequency of the trans-mitters. The platforms required to meet this test are: submarines; surface ships;space equipment and systems; Army, Navy, and Air Force aircraft; and Army,Navy, and Air Force ground systems.

10.3.4 Radiated Susceptibility Requirements Mil-Std 461E

10.3.4.1 RS101—Magnetic Field (30 kHz to 100 kHz)

RS101 is primarily intended to ensure the performance of equipment potentiallysensitive to low frequency magnetic fields. It is applicable to subsystems enclosuresand electrical cable interfaces. It is not applicable to electromagnetic coupling viaantennas. This test is required for the following platforms: submarines and Armyand Navy aircraft.

10.3.4.2 RS103—Electric Field (2 MHz to 40 GHz)

The main purpose of RS103 is to ensure that the EUT will continue to operate with-out degradation in the presence of electromagnetic fields generated by antennatransmissions both on board and outside of the tested platform. According toRS103, the EUT should not exhibit any malfunction, degradation of performance,or deviation from the specified requirements. The requirements are applicable toequipment, subsystems enclosures, and all interconnecting cables. Most require-ments are referenced up to 18 GHz, with an optional 40-GHz range. The fieldstrength can vary depending on the specific requirements. This is a widely used testand is required on the following platforms: submarines; surface ships; space equip-ment and systems; Army, Navy, and Air Force aircraft; and Army, Navy, and AirForce ground systems.


Abrams, S., and C. R. Brown, “A Primer on Regulations and Standards,” Compliance Engineer-ing, 1998.Altay, B., and S. S. Seker, “Application Tables for MIL-STD 461D Emission Tests,” Proc. IEEEInternational Symposium on Electromagnetic Compatibility, August 18–22, 1997, pp. 500–503.Björklöf, D., “EMC Standards and Their Application,” Compliance Engineering, 1999.“DoD Interface Standard, Requirements for the Control of Electromagnetic Interference Emis-sions and Susceptibility,” MIL-STD-461E EMC, 1999.Dorey, P., “Gap Analysis of Military Standards for CE Marking,” Proc. EMCUK 2008, October14–15, 2008, pp. 1–5.


Klok, H. A., “Risk Analysis by the Use of Commercial Equipment in a Military Environment,”IEEE EMC Society Newsletter, Winter 2001.Smith, J., “EMI Testing for IEC 61000-4-3 Edition 3,” Microwave Journal, Vol. 50, No. 6, June2007.


C H A P T E R 1 1

Measurement FacilitiesFacilites used for measurement of susceptibility and emission, or interferenceinclude: full anechoic and semianechoic chambers, open area test sites (OATS),reverbation chambers, and transmission line structures; the two following are themost known: TEM and GTEM cells. OATS is the oldest and still the most acceptedfacility. Full and semianechoic chambers are mostly used for testing larger equip-ment and are generally the most expensive types of facilities. TEM and GTEM cellsare mostly used for smaller equipment. The reverbation chamber can be almost anysize.

11.1 Full Anechoic and Semianechoic Chambers

The anechoic chamber is a facility used for testing with an electromagnetic fieldabsorbing wall, thus creating an electromagnetic-field-free environment. In acous-tics, anechoic rooms absorb sound; in RF, walls absorb electromagnetic radiation.The outer structure is the Faraday cage, which means that the interior of the room isquiet regarding RF radiation (i.e., there is no surrounding electromagnetic interfer-ence). Any radiation created inside the chamber cannot escape. For susceptibilitytesting, the floor must absorb the radiation (hence the name Full anechoic cham-ber), while for emission testing, the floor can be conductive (semianechoic chamber,see Figure 11.1). If the floor is removed, the chamber can be used for both types ofmeasurements.

Almost all chambers have ferrite absorber tiles, which can be used with pyrami-dal absorbers impregnated with carbon for the attenuation of radio waves. In earlyanechoic chambers only pyramidal absorbers were used for attenuation reflections,making the absorbers (approximately 1m long) effective only at frequencies from100 MHz and above. Ferrite tiles are used for frequencies of 25 MHz and higher.

For frequencies above 1 GHz, smaller absorbers (0.5m) are used in combinationwith ferrite tiles. In this way, the useful chamber range can rise up to 18 GHz. Fur-thermore, since the antennas in this frequency range have high directivity, the reflec-tions are localized, and only a part of the wall has to be covered with the absorbers.

The commercial chambers should have an area with uniformity of the field lessor equal to 6 dB and attenuation of 4 dB. In Table 11.1 some types of anechoicchambers are presented.

Anechoic chambers can have various dimensions. For testing susceptibility andprecompliance testing, the room must be large enough to allow for a distance of 3m(1m for military style applications) between the antenna and the device under test(DUT). There should also be additional 1m between the antenna, DUT, and the

185

room wall. The absorbing material must be placed on all six of the room walls. (Fortesting the emission, the floor can be movable.) Large chambers simulate 3-m or10-m Open Area Test Sites (OATS), and must be high enough to place the antennaat 4-m height. Currently, 5m chambers (which are much cheaper), instead of the10-m ones, are being experimented with.

When building the chamber, cable filtering should be considered. If possible,optical cables should be used as well as other nonmetal interfaces. An access boardwith RF connectors should be as close as possible to the measurement equipment.The cables should be as short as possible. Thus, the necessary amplifiers for obtain-ing sufficient electromagnetic field levels will not have to be high in power, sincethey are very expensive. The cables should have as little attenuation as possible for agiven frequency range.

The door must be large enough for the biggest equipment, and should be sealedwith gaskets and copper fingers to prevent leakage of electromagnetic fields in orout.

In anechoic chambers, absorbers attenuate radio waves. It is desired that theincident wave continues traveling (i.e., to “see” the impedance, which is close to theone of free space). Such impedance has to be created, even though the metal wallimpedance practically presents a short circuit. There are three methods for making

186 Measurement Facilities

Figure 11.1 Full anechoic and semianechoic chamber.

Table 11.1 Commercial Anechoic Chamber Size

Type l × w × h (m) Standard Testing Price (USD)*

Smallest (26 MHz–1 GHz) 7 × 3 × 3 IEC 61000-4-3 RF susceptibility, emission $100,000

Small (26 MHz–18 GHz) 8 × 4 × 4 IEC 61000-4-3GR-1089 mm

RF susceptibility, emission $120,000

3-m replica OATS(26 MHz–18 GHz)

9 × 6 × 5.5 IEC 61000-4-3ANSI C63.4GR-1089EN 50147

RF susceptibility, emission,EUT up to 2m

$300,000

5-m replica OATS(26 MHz–18 GHz)

11 × 7 × 5.5 Experimental(3m chamber)

RF susceptibility, emission,large EUT

$360,000

10-m replica OATS-a(26 MHz–18 GHz)

18 × 13 × 8 IEC 61000-4-3ANSI C63.4GR-1089EN 50147

RF susceptibility, emission,large EUT

$1,100,000

* Turning tables, cameras, and raising the floor can increase the price for an additional $15,000 USD.

the incident wave continue to travel: pyramidal absorbers, ferrite tiles, andSalisbury paper.

First, only one frequency will be observed. At λ/4 from the wall, the impedancewill be infinite (open end). The generator will “see” the short-circuited λ/4 trans-former as an open end. At this point, the reflected wave will be shifted by one periodand added to the incident wave. If the Salisbury paper (with free space impedance of377Ω) is placed at a distance of λ/4 from the metal wall, the metal wall will disap-pear for this narrow frequency range (Figure 11.2).

For a plane wave, this effect cannot be distinguished from the wave propagatingin free space. If the susceptibility testing had to be performed only at this frequency,the construction of the anechoic chamber would not be a problem. However, thechambers must be designed for a frequency range of 80 MHz to 1 GHz or more.

A possible solution is shown in Figure 11.3, in which several Salisbury paperswith different surface resistances are placed at λ/4 distance from the metal wall.Placing papers in such an order results in a reflection coefficient less than 0.1 at thefrequency range of 2.5 to 1 around λ.

Is it possible to cover the airplane with paper of a 377Ω impedance, thus mak-ing it invisible to the radar? The answer is no. The total impedance would still be377Ω in parallel to whatever impedance is below the Salisbury paper.

11.1.1 Absorbers

Pyramidal absorbers are just an expanded application of Salisbury papers. Manysmall reflections are created when the electromagnetic wave travels through the pyr-amid. Pyramids must be at least λ/2 long for the lowest frequency of interest (λ iseven better). This is shown in Figure 11.4.

11.1 Full Anechoic and Semianechoic Chambers 187

Salisbury paper377Ω

Metal wall

λ/4

Figure 11.2 Salisbury paper of impedance 377Ω placed at λ/4 from the metal results in the walldisappearing for the incident wave with a wavelength of λ.

Pyramidal absorbers (Figure 11.5) in anechoic chambers are applied for suscep-tibility and emission testing as well as antenna calibration. Pyramidal absorbers aremade of dense flexible foam and impregnated with carbon for obtaining the desiredelectrical characteristics. They are wideband and used in closed spaces. The pyrami-dal design enables multiple reflections of electromagnetic waves, while the carbonattenuates them through scattering and dissipation. The radio wave attenuation isaround 45 dB.


Metal wall3 Salisbury paper

λ/4 λ/4 λ/4

R = 1565Ω R = 250ΩR = 625Ω

Figure 11.3 Several Salisbury papers for a larger frequency range of lower reflection.

Metal wall

λ/2 for lowest frequency

µ = 1, = 2 1− jεr r

Figure 11.4 Pyramidal absorbers as a practical application of Salisbury papers.

Pyramidal absorbers are used in the frequency range of 80 MHz to 18 GHz;attenuation for a typical model is shown in Figure 11.6.

11.1.2 Ferrite Tiles

Ferrite tiles (Figure 11.7) can be used in combination with pyramidal absorbers (orrecently even alone) for covering walls of anechoic rooms for attenuating radiowaves. Ferrite tiles are resistant to fire, moisture, and chemicals. Compared withpyramidal absorbers (> 1m), they are much smaller (6 cm). The ferrite tile imped-ance must be 377Ω (i.e., the ratio of permeability and permittivity). However, thiswill not stop the reflection of the wave. The ferrite must be of a complex impedance(i.e., with losses for absorbing the electromagnetic wave energy). The typical attenu-ation for a ferrite tile of 1-cm width at 100 MHz is 11 dB, amounting to 22 dB intotal (tile attenuates both the incident and reflecting wave).

11.1 Full Anechoic and Semianechoic Chambers 189

Base 60 cm

Figure 11.5 Pyramidal absorber.

5

10

15

20

25

30

35

40

0,01 0,10 1,00 10,00 100,00

Attenuation (dB)

f (GHz)

Figure 11.6 Attenuation of a pyramidal absorber.

Figure 11.8 shows the basic principles of an electromagnetic absorber. Whenthe electromagnetic wave travels through free space and reaches a medium of differ-ent characteristics, the wave will partially reflect and partially absorb. The reflectedwave is more important. The ferrite tile thickness is chosen in such a way that therelative phase of the reflected and transmitted wave cancel each other out and create


Ferrite

Dielectric

Metal wall

Figure 11.7 Ferrite tile.

y

Reflectedwave

Incidentwave

Material 1 Material 2

Transmittedwave

Metal

HrHt

E rE t

z=0

yz

x

Hi

E i

Figure 11.8 Incident, reflected, and transmitted waves.

a resonant state. It is a function that depends on the electric characteristics of ferritematerials such as relative permeability (µr) and permittivity (εr), which determinethe reflection coefficient, impedance, and return loss according to the followingexpression and Table 11.2:

( )Zj d

fr

rr r= ⋅ ⎛

⎝⎜⎞⎠⎟

⎡

⎣⎢⎤

⎦⎥µ

ε

π

λµ εtanh

2 Ω (11.1)

Figure 11.9 shows the attenuation of ferrite tiles depending on frequency. It isclear that ferrite tiles are best to use at frequencies from 10 MHz to several hundredmegahertz.

11.2 Open Area Test Site (OATS)

OATS is the oldest and still the most accepted test facility for acceptance of results.Compared to anechoic chambers, it is much cheaper to build. OATS should beplaced close to the production site, but in a quiet RF surrounding. These two differ-ent requirements are often opposite. Furthermore, when selecting a location, inter-national standards need to be checked. It is desirable to use numerical modelingmethods before building.

11.2 Open Area Test Site (OATS) 191

10 100 1000f (MHz)

50

40

30

20

10

0

A(d

B)

Figure 11.9 Ferrite tile attenuation depending on frequency.

Table 11.2 Magnetic Characteristics

Permeability µr 2,100

Curie temperature Tc > 95°C

Resistance ρ 5 · 106 Ωcm

Specific density 5.2 g/cm3

Linear coefficient 9 · 10−6/ºC

Figure 11.10 shows the schematics of OATS. Distance F depends on the testcondition. The antenna should be able to move vertically in order to find the stron-gest signal (due to reflections from the ground). The ellipse dimension can be 3, 10,or 30m. The space above the ellipse should be free, without reflecting surfaces. Incase of precipitation, a dielectric roof is allowed. The emission from the equipmentunder test (EUT) has to be measured with an appropriate receiver and antenna.

Figure 11.11 shows the position of EUT and antenna. The antenna should beable to move vertically 1m to 4m. The turntable must be able to turn 180°. The dis-tance between EUT and the antenna can be 3m or 10m. Ferrites are placed on thecable going to the spectrum analyzer or test receiver.

Perfect OATS should have an infinite ground without metal objects in the vicin-ity (fence, power cables, and so forth). To be able to use OATS in all weather condi-tions, it is desirable to build a protection roof resistant to weather conditions. Thewalls and roof should be made of dielectric materials, since they have less impact onhigher frequencies.


Receivingantenna Movable

Receiver

EUT

2F

Ellipseboundary

1.73F

F

Figure 11.10 Schematics of an open area test site.

Turntable

EUT

80 cm

3 or 10 m

Ferriteson cable

1–4m

Ground plane

Figure 11.11 Placement of EUT and the antenna.

The reflection coefficient from the vertical incident wave to the thin wall is:

( )r

l r

f

=−π ε

λ

1(11.2)

where l is the wall thickness (m), εr is the permittivity of the wall, and λf is the wave-length in free space.

If the reflection coefficient is to be less than 0.1 (typical value) at 1 GHz (λf =0.3m), then the acceptable width l for a given, nonferrite, nonconductible wall is:

( )l

r r

= ⋅−

≈−

03 011

0011

. . .π ε ε

(11.3)

The basic modular design of OATS is shown in Figure 11.12. The structure israised from the ground.

When making a study, the attenuation of OATS can be affected by the follow-ing parameters:

• Ground size;• Boundary conditions of the ground and surrounding terrain;• Influence of moisture in different types of soil;• Conductivity of soil as a function of temperature.

The size of the EUT can be up to 2F/λ (Rayleigh criteria). Surface irregularitiesmust be within ± 20 mm. The antenna should be placed at a 1–4-m height forobtaining the highest field level. It is also necessary to perform detailed ambientfield measurements before building the OATS.

11.3 Reverberation Chamber

The reverberation chamber (Figure 11.13) is a relatively new type of facility for test-ing emission and susceptibility. It consists of a plain shielded chamber with low losswalls. It should not radiate outside, does not contain absorbers, and can be of anysize. On resonant frequencies, the reverberation chambers are resonators with alarge Q factor. Inside the chambers mode tuners are built in, which change bound-ary conditions inside the chamber, thus ensuring that the EUT is exposed to a fullenergy amount and that all of the EUT emissions can be measured. Statistically, uni-form and isotropic wave propagation with uniformly distributed polarizationoccurs. Testing in a reverberation chamber can be performed on frequencies above

11.3 Reverberation Chamber 193

Ground Ground

Figure 11.12 Raised ground—side look.

the cutoff frequency (i.e., in the area where modes can exist inside the chamber).The Q factor can be calculated from

QVS

= ⋅32 δ

(11.4)

where V is the chamber volume, S is the surface of inner walls, and δ is the wallthickness (skin effect):

δπ σµ

= 1f

(11.5)

Reverberation chambers are usually from 75 to 100 m3, although they can bemuch smaller. They are used for testing at frequencies higher than 200 MHz (up to18 GHz). Working with frequencies below 200 MHz requires very large rooms. Forfrequencies above 1 GHz, smaller chambers can be used (volume ~ 0.25 m3).

The shape of the chamber is irrelevant—different designs prove quite good. Thevolume is the key factor. Around 50% of the volume is useful for testing, which ismore than with other types of chambers.

Mode tuners or propellers (Figure 11.14) are made of four equal boards turningaround a vertical axis. The turning of the tuners is regulated by step motors andmicrocontrollers, which must be placed outside of the room. Step motors andmicrocontrollers should not conductively be coupled with the interior of thechamber.

Reverberation chambers can only measure the isotropic radiation of EUT andnot the electrical field at a certain distance, which is often required in internationalstandards. The price of a reverberation chamber (6.55m × 5.85m × 3.50m) with thelowest usable frequency of 124 MHz is about $50,000.


Figure 11.13 Reverberation chamber.

11.4 TEM Cell

The appearance of the TEM cell (or Crawford cell) in 1974 intensified testing in thefields of electromagnetic compatibility, biomedical effects, and electromagnetic dis-turbances. The first TEM cell, still in use, underwent many improvements. TheTEM cell is a simple and economical surrogate for Open Area Test Systems. The testsystem should have an area with no outside interference, and at the same time a uni-form electromagnetic field inside. With increasing frequency, the cell dimensionsget smaller, thus becoming too small for testing larger devices, except for possiblyprinted circuit boards. The goal of technical improvements to various cell types isincreasing useful test area for higher frequencies using absorbers, obtaining a fieldas uniform as possible, and avoiding the appearance of higher-order modes of prop-agation. This is not a simple task and requires a lot of numerical modeling, use ofvarious numerical methods, and testing of prototypes. In the meantime, TEM-cellshave been acknowledged as standardized test systems for electromagnetic compati-bility and interference testing. Research is ongoing; new absorption materials arebeing developed and new geometrical structures tried.

Transversal electromagnetic (TEM) transmission cells are devices used forestablishing uniform electromagnetic fields in a shielded environment. They arestructures with three closed transmission lines for preventing radio frequency radia-tion and electrical isolation. The TEM cell is made of a quadrature transmission linewith pyramidal parts at the ends for adapting to standard coaxial connectors(Figure 11.15).

A uniform TEM field is established inside the cell on any desired frequencybelow the cutoff frequency, at which higher-order modes start to appear. TEM cellsare used for testing small equipment, calibrating of radio frequency probes, and bio-medical experiments. The wave propagating through the cell has a wave impedanceequal to free-space impedance (377Ω), thus enabling a good approximation of aplanar wave in a far field.

The cells are wide bandwidth and have linear phase and amplitude frequencycharacteristics from the DC to the cell cutoff frequency. This feature enables testingwith a continuous wave (CW) or over a selected frequency range, as well as with a

11.4 TEM Cell 195

Figure 11.14 Mode tuners.

pulse or modulated signal. The cell has its limitations—the main being that theupper cutoff frequency is determined solely by the physical dimensions of the cell.This results in limitations on the DUT size.

The expression for obtaining the electrical field in the cell, where V is the volt-age on the septum, b/2 is the distance from the septum to the cell wall, P is the powerlevel introduced into the cell, and Z0 is the characteristic impedance of the cell, isdefined as follows:

EV

b

PZ

b= =

2 20 (11.6)

11.4.1 Characteristic Impedance

Characteristic impedance of the symmetrical stripline, which has a metal shield atthe sides (Figure 11.16), is given with the values of cross section dimensions andunknown edge capacitance over the unit length Cf´:

( )[ ]Z

w b t Cf

0

3766

4=

− + ′.

ε(11.7)

where ε0 = 8.852 · 10−12, with an air dielectric. For a central conductor with smallthickness,

( )C b

b ta wb t

ta w

f

ε π

π=−

+ −−

⎛⎝⎜

⎞⎠⎟

⎡⎣⎢

⎤⎦⎥

+−

21

2ln coth (11.8)


Coaxialtermination

EUT

Septum

Outershield

Coaxialconnector

RF source

Figure 11.15 TEM cell.

a

b/2

b/2gg w

t

Figure 11.16 TEM cell cross section.

The upper expression is valid for (a w)/2b < 0.4.The Crawford cell (and other TEM cells) is designed to have a characteristic

impedance of 52Ω. This value is chosen because when inserting a DUT, the charac-teristic impedance will drop slightly.

11.4.2 Higher-Order Modes

The basic restriction of the TEM cell is the appearance of resonances, which destroythe uniform field distribution of the TEM mode of wave propagation. There arenumerical methods for establishing the cutoff frequency of the higher-order modesas a function of septum (inner conductor) width. The determination of the resonantcell length is not simple, because the cell pyramidal parts are acting differently ateach higher-order mode. The TEM cell is a resonant cavity with a high Q, where thehigher-order modes tend to appear at exactly determined frequencies. There is awindow between these resonances where the use of the TEM cell is still possible. Towhich degree these structures can be used with the presence of higher-order modesand whether it is possible to use them between these resonances will depend on thepractical implementation for which the cell is desired. Generally, the cutoff fre-quency in a perpendicular waveguide for TE10 mode, which is usually the firsthigher-order mode that starts to propagate, is given with the following expression:

( )f TEcac 10 2

= (11.9)

where c is the speed of light. The expression for the cutoff frequency for anyhigher-order mode TEmn is as follows:

( ) ( )f TE

c b m a n

bac m n, =+2 2 2 2 1 2

2(11.10)

where a and b are the dimensions of the waveguide cross section and m and n are thenumber of half-periods of the electric field in the x- and y-axes.

The TEM mode propagates through pyramidal parts of the cell without any sig-nificant change. Every higher-order mode is always reflected at the same place of thepyramidal part until it becomes too small to propagate. The energy of propagationof the higher-order mode suffers from repeated reflections inside the cell until itexhausts itself. The resonant conditions are fulfilled when the effective cell lengthfor a particular mode is equal to the number of half-lengths (p = 1, 2, 3, ...), p beingthe number of half wavelengths. On resonant frequencies, fR(mnp), there is a resonantfield of the TEmnp mode. If we use the expression:

( ) ( )l p pmn g mn= =λ 2 1 2 3; , , (11.11)

and

( )

1 1 12 2 2λ λ λ

= +g c mn

(11.12)

11.4 TEM Cell 197

where λ c cm( )2 represents the value of wavelength at the cutoff frequency, the follow-

ing expression, which predicts various resonant frequencies, can be obtained:

( ) ( )( )

f fpc

lR mnp c mnmn

2 2

2

2= +

⎛

⎝⎜⎜

⎞

⎠⎟⎟ (11.13)

with f cc mn c mn( ) ( )/= λ2 .Spreading of the useful frequency range of the cell can be achieved by filling the

cell with absorbers. It will lessen the quality factor (Q) inside the cell, which is fre-quency dependent. The absorbers improve the uniformity of the field between theseptum and upper and lower walls, thus increasing the vertical component of theelectric field on edges of the septum.

Higher-order modes have a special influence on the TEM cells. Their appear-ance disrupts the uniformity of the electromagnetic fields inside the cell. The firsthigher-order mode depends solely on cell dimensions. After the first higher-ordermode, the other higher-order modes starts to appear. The cell can be used even afterhigher-order modes start to appear. In the following text, a calculation of thehigher-order modes according to experimental equations is shown. Higher-ordermodes have two characteristic frequencies: cutoff and resonant. For every mode, theexpression for the cutoff frequency is as follows:

( ) [ ] [ ]fx c

bxbc m n, = ⎛

⎝⎜⎞⎠⎟

=2

150π π

Hz MHz (11.14)

where c = 3 × 108 m/s, and where x depends on the mode and is given later.The appropriate resonant frequency is determined by:

( ) ( )f fpc

LR m n p c m nmn

, , ,2 2

2

2= +

⎛⎝⎜

⎞⎠⎟ (11.15)

where Lmn is the effective cell length for every mode,

L L X Lmn c mn E= + (11.16)

and Lc is the length of the central section of the cell, and LE is the length of the twopyramidal ends. Xmn is an empirically defined multiplier. Table 11.3 gives the equa-tions for calculating cutoff frequencies of higher-order modes.

11.4.3 TEM Cell Construction

The TEM cell (like most other cells) should be designed to have a characteristicimpedance of 52Ω. When the TEM cell was designed at the Faculty of ElectricalEngineering and Computing, Zagreb (FER), its future purpose was taken into con-sideration (biomedical experiments, probe calibration, and electromagnetic com-patibility). The aim was to achieve as much space as possible for testing at thefrequency of 900 MHz, and sustain the characteristic impedance of the cell of 50Ω


at the same time, which was difficult to achieve. The usual cell dimensions are a > b(i.e., the cell is wider than it is tall). The value a < b was chosen, which provided formore space in the vertical dimension. This resulted in the change of the characteris-tic impedance from 50Ω to 75Ω. Even though most network analyzers and signalgenerators operate at 50Ω, the adaptation can be achieved through a 50/75Ω trans-former. Figure 11.17 shows the blueprints for the TEM cell designed at FER.

11.4 TEM Cell 199

Table 11.3 Expressions for Calculating the Cutoff Frequencies ofHigher-Order Modes

Mode Expressions

TE01x x b

aag

R

Rp

TE

TE

tan ln= ⎛⎝⎜

⎞⎠⎟

⎛⎝⎜

⎞⎠⎟ +⎡

⎣⎢

⎤

⎦⎥

=

−π

π22

1

01

01

1

coth cosp ba

p gap

π π−⎛⎝⎜

⎞⎠⎟

⎛⎝⎜

⎞⎠⎟

=

∞

∑ 1 2

1

TE10 x ba

= ⎛⎝⎜

⎞⎠⎟

π

2

TE11x y b

a

y yga

ba

= + ⎛⎝⎜

⎞⎠⎟

⎡

⎣⎢⎢

⎤

⎦⎥⎥

=⎛⎝⎜

⎞⎠⎟

22

2

2

2

8

π

ππ

tancos

ln ag

ga

R

Rp

TE

TE

π

π⎛⎝⎜

⎞⎠⎟ − ⎛

⎝⎜⎞⎠⎟

+⎡

⎣⎢

⎤

⎦⎥

=+

−

22

2 12

2

1

11

11

cos

( )1

2 1

21 2 1

22

1

coth cosp b

ap

ag

p

+−

⎛

⎝⎜⎜

⎞

⎠⎟⎟

+⎛⎝⎜

⎞⎠⎟

=

∞

∑π

π

TE02 x = π

TE12, TM12x b

a= + ⎛

⎝⎜⎞⎠⎟

⎡

⎣⎢⎢

⎤

⎦⎥⎥

π 12

2

TE20 x ba

= π

TE21

cotcos cot

ln cosxx

ga

y

yab

ag

ga

+

⎛⎝⎜

⎞⎠⎟ = ⎛

⎝⎜

⎞⎠⎟ −

22 2

2

2

π

π π

π⎛⎝⎜

⎞⎠⎟

+⎡

⎣⎢

⎤

⎦⎥

= − ⎛⎝⎜

⎞⎠⎟

⎡

⎣⎢⎢

⎤

⎦⎥⎥

=

R

y x ba

R

TE

TE

21

21

22

1

2

1

π

pp ba

p bap

coth cosπ π−⎛⎝⎜

⎞⎠⎟

⎛⎝⎜

⎞⎠⎟=

∞

∑ 1 2

2

TM11

x y ba

yy

ab

ag

= + ⎛⎝⎜

⎞⎠⎟

⎡

⎣⎢⎢

⎤

⎦⎥⎥

= ⎛⎝⎜

⎞⎠⎟ + −

22

1

2

2

2

2 2 1

π

π π

tan

( )

R

R ag

p b

a

TM

TMp

11

11

1

1

4 2 1

21

⎡

⎣⎢⎢

⎤

⎦⎥⎥

=+ +

−⎡

⎣⎢

⎤

⎦⎥

−

=

∞

∑πcoth

( )J

p g

a1

2 1

2

+⎡

⎣⎢

⎤

⎦⎥

π

The size of the device that can be tested in the cell is 5 cm high, which is 1/3 ofthe total TEM cell height. The cell is made of aluminum, whereas the septum ismade of 1.5-mm-thick copper; any metal available is acceptable. The connectors areBNC, and from the sides there are doors to insert the DUT. There are openings forthe wires below the doors. The septum is supported with dielectric material (Tef-lon). The cell can withhold up to 50W without cooling. Since it is made of alumi-num, it is very light for carrying and handling.

11.4.4 Parameter Measurements

The TEM cell was tested at the Department for Radiocommunications with Net-work Analyzer HP 8620B. VSWR (Figure 11.18) transmission characteristics (Fig-ure 11.19) were measured and the Smith chart (Figure 11.20) was obtained as aresult.

Transformers 50/75Ω were used for adaptation. At the frequency of 935 MHz,VSWR was measured to be only 1.06 and absorption 4.5 dB. Characteristic imped-ance was 80.8 − j0.92Ω, which should be higher than 75Ω because it will drop oncethe DUT is inserted in the cell. To ensure the field inside the cell, a probe is neces-sary. This will be discussed in the following chapters. Figure 11.21 shows the photo-graph of the TEM cell built at FER.


12.5 cm 25 cm

25 cm

25cm

12.5 cm

BNCconnector

Door

Guideholes

Dielectricsupporters

Septum

3.5 cmgap

18cm

30cm

Figure 11.17 Scheme of the TEM cell.

11.5 GTEM Cell

The GTEM cell is a transmission line based on the TEM cell approach. The letter Gstands for gigahertz, since the GTEM cell operates from DC to 18 GHz.

The slightly curved wave front (not a planar wave) travels from the source to the50Ω quadratic shielded transmission line (radial type) to the hybrid terminationwithout geometrical distortion of the TEM wave. This transmission line can be sym-metric or nonsymmetric (the latter being more frequent), in order to obtain a moreuseful testing area. Symmetrical transmission lines are also called coaxial. Since thewaveguide incident angle is small (20°), the wave can be considered planar. TheGTEM cell is an adaptable (pyramidal) part of a quadratic transmission line (TEMcell) with a characteristic impedance of 50Ω.

The GTEM cell (Figure 11.22) starts with a precise apex, where the transitionfrom the standard 50Ω N-type connector to a nonsymmetrical quadratic waveguideis done.

The distributed load consists of absorption material used for the termination ofthe electromagnetic wave, and of distributed resistance used for terminating lowfrequency currents. On low frequencies, the cell has an impedance of 50Ω. At higherfrequencies, the absorber attenuates the incident wave in much the same way as inthe anechoic chamber. Thus, the matching from DC to several gigahertz is achieved.

Wideband performance, which is enabled by termination load, lowers the influ-ence of higher-order modes. The absorbers significantly reduce the quality factor of

11.5 GTEM Cell 201

100 200 300 400 500 600 700 800 9001

1.5

2

2.5

3

3.5

Frequency (MHz)

VSW

R

Figure 11.18 VSWR of the TEM cell.

100 200 300 400 500 600 700 800 9002

3

4

5

6

7

8

Frequency (MHz)

Abs

orp

tion

(dB)

Figure 11.19 Absorption of the TEM cell.


Figure 11.20 Smith chart of the TEM cell.

Figure 11.21 TEM cell developed at FER.

the cell, thus lessening the influence of resonances. The TEM mode generated with acontinuous source or pulse generator simulates the planar wave for testing emissionand susceptibility.

11.5.1 GTEM Cell Characteristics

GTEM cell characteristics (Figure 11.23) are:

• Characteristic impedance of 50Ω;• Septum at 3/4 of the cell height (for larger EUT);• Width/height ratio of 2/3;• 15° angle between the septum and the lower shield;• 5° angle between the septum and the upper shield.

The N-type connector is placed at the end of the pyramidal part. The septum issupported with dielectric material. On the other side of the cell, there is a distributedtermination. The DUT size can be 1/3 of the size between the septum and shield.

11.5.2 GTEM Cell Construction

Figure 11.24 shows the schematic of the GTEM cell built at FER, Zagreb. The sep-tum, as well as the shielding, is made of copper. The N-type connector is placed atthe beginning of the pyramidal end.

11.5 GTEM Cell 203

Figure 11.22 GTEM cell.

2wgg

2b

2a

Figure 11.23 GTEM cell cross section.

Dielectric supporters of the septum are made of Teflon. On the other side of thecell there are pyramidal absorbers of 25 cm for matching electromagneticwaves and two parallel 100Ω resistors for current termination. The EUT size is20 cm × 20 cm.

The cell is designed in such a way as to enable replacement of pyramidal absorb-ers with ferrite (or some other more efficient absorbers) in the future. In this way,more testing area is obtained. The N-type connector can also be easily replaced incase of damage. The resistance array of 100Ω can also be replaced, or instead of two100Ω resistors some other combination of resistors can be introduced (6 × 300Ω,for example).

Figure 11.25 shows the GTEM cell cross section. The outer measures areslightly different from the inner due to the side connecting on edges. The sides areconnected with silver.

11.5.3 GTEM Cell Parameter Measurement

The GTEM cell must be tested for its voltage standing wave ratio (VSWR), trans-mission characteristics (reflection), and time-domain measurements.


SeptumDoor20 20cm×

115 cm

38,5cmN-type connector

60 cm

Figure 11.24 GTEM cell blueprint.

10 mm 10 mm600 mm inner

620 mm

400m

m

Figure 11.25 Cross section of GTEM cell.

Figures 11.26 to 11.29 show the VSWR and reflection from 1 GHz to 20 GHz.The resistors have more influence on lower frequencies, but their influence weakensgreatly after 100 MHz. After 500 MHz their influence is negligible and absorbersstart dominating above that frequency. Figure 11.30 shows the time-domainresponse of the GTEM cell. The higher magnitude levels are due to reflections at theconnector and dielectric supporters along the stripline.

11.5.3.1 Measuring Electric Field Strength Inside the Cell

In the FER project the measuring of the electric field was carried out with the radiofrequency signal generator HP 8656A (0.1–1,040 MHz), amplifier MiniCircuits 28dB (100–900 MHz), probe Holaday HI-4455, and readout device HI-4460.HI-4460 is a graphical device for reading the values of electromagnetic fields, and ithas a screen made of crystals for displaying numerical and graphical values. Thedevice can be connected to a computer through a RS232 interface. Probe HI-4455 isa battery-operated wide-bandwidth isotropic probe for measuring the electric fieldin the vicinity of the RF source. The application includes measuring microwavetransceivers and antennas, and monitoring electromagnetic interference (EMI). Theprobe uses optical isolation for keeping field changes low during measurements andhas a conical casing and sensor inside. The sensor is placed on one end of the sup-port, while the other part is connected to the electronics. With three orthogonallyplaced dipole antennas, the probe measures the field intensity in three directions,calculates the sum, and sends the results to the receiver over an optical cable. Thefrequency response is from 200 kHz to 40 GHz, whereas the dynamical range isfrom 1.5 to 300 V/m.

11.5 GTEM Cell 205

1.00

1.20

1.40

1.60

1.80

2.00

0.20 0.40 0.60 0.80 1.00

f (GHz)

VSW

R

Figure 11.26 VSWR up to 1 GHz of the GTEM cell.

0.001.00

1.50

2.00

2.50

3.00

3.50

VSW

R

5.00 10.00 15.00 20.00

f (GHz)

Figure 11.27 VSWR up to 20 GHz of the GTEM cell.

Figure 11.31 shows the field distribution inside the GTEM-cell at 1/2 of the sep-tum height—100 MHz. It is obvious that the measurements are in concordance withthe modeled values (FEM method).

The field was measured at different locations: the middle and at the edges of thecell. Figure 11.32 shows the frequency response of the GTEM cell in the middle ofthe area reserved for testing with an input power of 40 dBm. It is obvious that theresults are within 3 dB.


0.20 0.40 0 60. 0 80. 1 00.

−30

−25

−20

−15

−10

0

−5

f (GHz)

Refle

ctio

n(d

B)

Figure 11.28 Reflection up to 1 GHz of the GTEM cell.

0 00. 5 00. 10 00. 15 00. 20 00.

−30

−25

−20

−15−10

0

−5

f (GHz)

Refle

ctio

n(d

B)

−35

Figure 11.29 Reflection up to 20 GHz of the GTEM cell.

0.0 0.5 1.5 1.0 2.0 2.5 3.0 3.5

0.02

0.040.06

0.080.100.12

0.14

t (ns)

Mag

nitu

de

0.00

Figure 11.30 Time-domain response of the GTEM cell.

11.5.3.2 Higher-Order Modes

Higher-order modes do not play such an important role with GTEM cells as withTEM cells, because the GTEM cell is not a high Q factor cell. Higher-order modesare attenuated and therefore hard to measure. There are analytical calculation equa-tions of higher-order modes, which are valid only for the first several modes sincethe absorbers have a high influence at higher frequencies. Higher-order modesappear due to dimensions change, nonuniform mediums of absorbers, and finiteconductance of the walls. In the vicinity of the N connector higher-order modes can-not propagate, but when moving along the septum axis, the possibility of theirappearance increases. The modes stimulated by the TEM mode are called essential.Other modes, stimulated by disturbances or small discontinuities, do not achievelarge amplitudes and are called nonessential.

The first several modes that can propagate in the GTEM cell are H01(TE01), H10,H11, and H20, and after them E11 and E22. Even though mode H01 is the first to startpropagating, the first essential mode is H10. (Regarding resonances in the GTEMcell, it is the most important one.)

In nonuniform waveguides, transversal fields are the functions of the z coordi-nate (i.e., the direction of wave propagation). They can be expressed with the fol-lowing vector functions:

11.5 GTEM Cell 207

4.2

4.3

4.4

4.54.64.7

4.8

4.9

5.0

5.1

5.2

0.0 0.1

Num. method

Measured

0.2 0.3 0.4 0.5 0.6

Figure 11.31 Field distribution inside the GTEM cell at 1/2 of the septum height (100 MHz).

1

10

f (MHz)

E(d

b/(V

/m)

100

1000

100 1000

Figure 11.32 Measured electric field strength inside the GTEM cell with an input power of40 dBm.

( ) ( ) ( )

( ) ( ) ( )

H x y z I z h x y z

E x y z V z e x y z

tr i ii

tr i ii

, , , ,

, , , ,

=

=

=

∞

=

∞

∑

∑

1

1

(11.17)

The vectors

hi and

e i for TE and TM modes can be expressed as:

( ) ( ) ( ) ( )

e T h n TEtr

E Ez tr

E= −∇ = − × ∇ (11.18)

( ) ( ) ( ) ( )

e n T h THtr

H Htr

H= − × ∇ = −∇2 (11.19)

T(E) and T(H) fulfill differential equations and boundary conditions:

( ) ( ) ( )

( ) ( )∇ + =

=tr

Ec

E E

E

T k T

T L z

2 2 0

0 on(11.20)

( ) ( ) ( )

( )

( )

∇ + =

=

trH

cH HE

H

T k T

Tn

L z

2 2 0

0∂

∂on

(11.21)

where L is the boundary curve of the cross section.Putting (11.18) into the Maxwell equations, the following system of differential

equations is obtained:

( ) ( ) ( ) ( ) ( )

( )( ) ( )

dV

dzz Z z I z C z V z

dI

dz

z

Z zV z C

kk Lk k ki i

i

k k

Lkk k

= − +

= − −

=

∞

∑γ

γ1

( ) ( )i ii

z I z=

∞

∑1

(11.22)

The nonuniform waveguide can be regarded as the system of coupled transmis-sion lines, where the coupling factors Cik are the functions of z. Field strengths canbe expressed with voltages Vk and currents Ik. This equations system is known astelegraphic equations. The characteristic impedance values of ZL and spreadingconstant γ are given in Table 11.4.

Separating one equation from the other (11.22), the Schroedinger equation forcalculating higher-order modes is obtained:

( )( ) ( )( ) ( )d I

dzk z k Ik

E

cE

kE

2

2

2 2 0− − = (11.23)

( )( ) ( )( ) ( )d V

dzk z k Vk

H

cH

kH

2

2

2 2 0− − = (11.24)

for E (TM) modes (11.23) and H (TE) modes (11.24), where Vk and Ik are equivalentvoltages and currents of the electromagnetic field inside the waveguide.


The propagation constant in every cross section can be expressed as

( ) ( )γ k cz k z k= −2 2 (11.25)

( )γ E Hc

c

c

k k

k k

k k

, is

real

imaginary

2 2

2 2

2 2

0

<=

<

⎧

⎨⎪

⎩⎪

(11.26)

If γ is real, the mode will attenuate and (11.23) and (11.24) describe the wavethat is attenuated. If γ is imaginary, the mode is above the cutoff frequency andSchroedinger equations describe the propagating wave. Local cutoff frequencies canbe calculated according to the following expression:

fk k c

cc c= =

2 2π εµ π(11.27)

where kc is the wave number. For most TEM cells, the product of the wave numberand the width of the cell are constant:

k a konstc ⋅ = (11.28)

By solving (11.23) and (11.24), the expression for resonant frequencies of thehigher modes is obtained. For the GTEM cell, where kca = konst, the cross section ofthe cell changes linearly in the direction of wave propagation. The boundary curvecan be described with

( ) ( )a z mz a z= + = 0 (11.29)

Using (11.25) the following is obtained:

( )( )( )

d V

dz

konst

mz ak V

2

2

2

2

22

00−

+−

⎛

⎝

⎜⎜

⎞

⎠

⎟⎟ = (11.30)

where konst is different for every mode. If a replacement is introduced:

11.5 GTEM Cell 209

Table 11.4 Values of ZL and γ for TE and TM Modes

TE Modes TM Modes

( )( ) ( ) ( )γ Ec

Ez k z k= −2 2 ( )( ) ( ) ( )γ Hc

Hz k z k= −2 2

( ) ( ) ( )Z k k zL

Ec

E= −1 2 2

ωε( )

( ) ( )Z

k k zL

H

cH

=−

ωµ1

2 2

( )( ) ( )( ) ( ) ( )( )γωε

E

L

Ec

Ez Z zj

k z k= −1 2 2 ( )( ) ( )( )γ ωεH

L

Hz Z z j=

( )( )( )( )

γωε

E

L

E

z

Z zj=

( )( )( )( )

( ) ( )( )γ

ωε

H

L

H cHz

Z z jk z k= −1 2 2

( )a mz b

d

dzm

d

dam

b

l= + ⇒ = =

2

2

22

2

3 4(11.31)

Equation (11.31) takes a special form of the Bessel differential equation

d V

dz

CC V

2

212

2 22 0− −

⎛

⎝⎜

⎞

⎠⎟ =

α(11.32)

where

Ck a

mC

k

mc

12

22

2

2

2

2= ⎛

⎝⎜⎞⎠⎟

=i (11.33)

By solving the Schroedinger equation, the following is obtained:

( ) ( )V A mz b J k zbm

vk a

mH

vc= + +⎛

⎝⎜⎞⎠⎟

⎛⎝⎜

⎞⎠⎟ = + ⎛

⎝⎜⎞⎠⎟1

14 2

; (11.34)

where m is the ratio of opening b and cell length l. Jv are the zero points of Besselfunctions and constant A1 can be obtained by solving the telegraphic equations.

Figure 11.33 shows the amplitudes of the first, second, and third resonance inthe GTEM cell (made at FER) obtained by a specially developed software.

For a GTEM cell, the following is valid:

• Opening height in regard to width, b/a = 2/3;• Septum height of 3/4b;• Septum thickness in regard to opening height t/b = 0.004;• Septum width in regard to opening width w/a = 0.65;• Cell length l.

The values of kca are given in Table 11.5.By solving (11.34) for mode H10, three resonant frequencies are obtained, stim-

ulated by the cutoff frequency of that particular mode at 249.614 MHz: f101(H) =

406.01 MHz; f102(H) = 558.28 MHz; f103

(H) = 602.34 MHz. H01, which is the first


Table 11.5 Values ofkca Depending on thePropagation Mode

MOD kca

H01 2.640

H10 3.138

H11 5.418

H20 6.281

E11 6.002

E21 8.699

propagating mode (but actually the second mode), has the cutoff frequency of210.085 MHz, from which the following resonant frequencies appear: f011

(H) =362.03 MHz; f012

(H) = 509.16 MHz; f013(H) = 650.21 MHz. For other modes, the reso-

nant frequencies are even higher and the absorber influence is stronger above450 MHz.

11.5.4 Current Distribution at Septum

The resistance influence is more important at lower frequencies and can beneglected above 500 MHz. It is important to determine where to place the resistors;the VSWR must be kept in mind, and field must be uniform. This is why the resistorsare placed closer to the septum edges, so that the currents in this area flow close tothe edges and do not contribute to the nonuniformity of the field in the area desig-nated for testing. The testing area should be 25 to 45 cm away from the absorbersinside the cell. Placing the resistors close to the middle of the septum results in aworse VSWR at lower frequencies. The nonuniformity of the field increases as well.It is not important that the currents flow, but that they flow parallel with the septumedges (Figure 11.34). Figure 11.35 presents a photo of the GTEM cell built at FER.

11.5 GTEM Cell 211

−1.2 − .1 0

−0.10

−0.05

0.00

0.05

0.10

0.15

0.20

− .0 8 − .0 6 − .0 4 − .0 2 0 0.

f r1f r2

f r3

z

Figure 11.33 H01 amplitudes of the first, second, and third resonances in the GTEM cell.

Testing area

Good Not good

Figure 11.34 Current flow in the septum.


Crawford, M. L., “Generation of Standard EM Fields Using TEM Transmission Cells,” IEEETransactions on Electromagnetic Compatibility, Vol. EMC-16, No. 4, November 1974.Crawford, M. L., J. L. Workman, and C. L. Thomas, “Expanding the Bandwidth of TEM Cellsfor EMC,” IEEE Transactions on Electromagnetic Compatibility, Vol. EMC-206, No. 3, August1978.Hill, D. A., “Bandwidth Limitations of TEM Cells Due to Resonances,” Journal of MicrowavePower, Vol. 18, June 1983, pp 181–195.Koch, M, and H. Garbe, “Analytische und Numerische Parameterstudien in TEM/Zellenrechteckformigen Querschnitts,” Proc. Elektromagnetische Vertraglichkeit, Deutschland: VDE-Verlag Gmbh, 1998, pp. 255–262.Koenigstein, D., Hansen, D., “A New Family of TEM-Cells with Enlarged Bandwidth and Opti-mized Working Volume,” Proc. 7th Zurich Symp. and Techn. Exh. on EMC, March 1987,pp. 172–132.

Malaric, K., and J. Bartolic, “TEM Cell with 75Ω Impedance for EMC Measurements,” Proc.IEEE 1999 Int. Symposium on Electromagnetic Compatibility, Volume 1, August 2–6, 1999,Seattle, WA, pp. 234–238.Malaric, K., J. Bartolic, and B. Modlic, “Absorber and Resistor Contribution in the GTEM-Cell,”Proc. IEEE 2000 International Symposium on Electromagnetic Compatibility, August 21–25,2000, Washington, D.C., pp. 891–896.


Figure 11.35 GTEM cell built at FER.

Malaric, K., J. Bartolic, and B. Modlic, “TEM-Cell with Increased Usable Test Area,” Proc. Inter-national Conference on Telecommunications—ICT ‘99, Vol. 2, June 15–18, 1999, Cheju, Korea,pp. 370–374.Morgan, D., A Handbook for EMC Testing and Measurement, London, U.K.: Peter PeregrinusLtd., 1994.Nahman, N. S., et al., “Methodology for Standard Electromagnetic Field Measurements,” IEEETransactions on Instrumentation and Measurement, Vol. IM-34, No. 4, December 1986.Weil, C. M., “The Characteristic Impedance of Rectangular Transmission Lines with Thin CenterConductor and Air Dielectric,” IEEE Transactions on Microwave Theory and Techniques, Vol.MTT-26, No. 4, April 1978.Weil, C. M., and L. Gruner, “High-Order Mode Cutoff in Rectangular Striplines,” IEEE Trans-actions on Microwave Theory and Techniques, Vol. MTT-32, No. 6, June 1984.Wilson, P. F., et al., “Simple Approximate Expressions for Higher Order Mode Cutoff and Reso-nant Frequencies in TEM Cells,” IEEE Transactions on Electromagnetic Compatibility, Vol.EMC-28, No. 3, August 1986.Wilson, P., F. Gassmann, and H. Garbe, “Theoretical and Practical Investigation of the Field Dis-tribution Inside a Loaded/Unloaded GTEM Cell,” Proc. 10th International Zurich Symposium onEMC 1993, Zurich, Switzerland, 1993, pp. 595–598.

11.5 GTEM Cell 213

C H A P T E R 1 2

Typical Test EquipmentInterference into a piece of equipment or a system can be radiated or guided. Whiletest facilities for radiated emission were covered in Chapter 11, in this chapter moreattention will be paid to guided coupling. Coupling is performed either throughcapacitance or inductance. Typical test equipment includes: a line impedance stabi-lization network (LISN), coupling capacitors, coupling transformers, a parallelplate for a susceptibility test, coupling clamps and probes, injection clamps andprobes, an EMI receiver, spectrum analyzers, and oscilloscopes.

12.1 LISN—Line Impedance Stabilization Network

A line impedance stabilization network (LISN) is a device for direct coupling withequipment under test (EUT). It is also called an artificial mains network (AMN) andis used to measure distortion signals on the mains cord of electrical EUTs and comein many types. Distortion signals are usually generated or picked up inside the EUTand the mains cord acts as an antenna. European and international EMC regula-tions define maximum permissible signal levels and frequency bands for such distor-tion signals (CISPR 16-1-2, MIL-STD-461D and E).

For testing, LISN is placed between the power source and the equipment(device) to be tested during electromagnetic interference testing on power lines.Since the input impedance depends on frequency, LISN stabilizes the impedance at50Ω. Furthermore, LISN filters the radio frequency noise from the mains supply.Finally, LISN transfers the conducted interference voltage produced by EUT to aspectrum analyzer or EMI receiver. LISN (Figure 12.1) is used for measuring theguided RF signal from the mains to the EUT.

Every LISN has a low pass filter for rejecting noise on cables, as well as an effec-tive voltage transient limiter to protect sensitive analyzers or receivers from a strongenergy shock. Large capacitors and inductors absorb unwanted noise energy andpractically electrically isolate the EUT from the power mains. In this way, nounwanted guided noise from the power mains can enter the EUT, and vice versa.Thus, only the noise generated by the EUT will be measured by the spectrum ana-lyzer or EMI receiver.

LISN is usually placed on a metallic board near the EUT (Figure 12.2). The EMIreceiver is connected to the measuring port via a quality coaxial cable. Measuringunused LISN outputs are matched with 50Ω. LISN is widely used in commercialand military applications for electromagnetic compatibility measurements. How-ever, it is frequency limited. It can be used in the frequency range of 150 kHz to 30MHz, although some models can be used at frequencies as high as 400 MHz.

215

12.2 Coupling Capacitor

The coupling capacitor is used for testing spike voltages. Their typical value is 10µF, which represents the known RF impedance on mains. Furthermore, it preventsundesirable frequencies from contaminating the power source (Figure 12.3). 10 µFcapacitors should sustain voltages of up to 600-V DC and currents of up to 100Awithout significant losses.

216 Typical Test Equipment

To mains source To EUT

To 50 load or to50 input of meter

ΩΩ

50 Hµ

8 Fµ 0,25 Fµ

1 kΩ5 Ω

Figure 12.1 LISN schematics.

LISN forevery line

AC or DCport

Matching the unusedline with 50Ω

Coaxialcable 50Ω Input

impedance 50Ω

Groundedboard

EUT

Figure 12.2 Setup for measuring guided emission using LISN.

Transientgenerator

Osciloscope

EUT

10 Fµ 10 Fµ

Figure 12.3 Testing spike voltage.

Coupling capacitor dimensions are usually 0.85 × 0.85 × 0.7 cm. The attenua-tion of the RF signal on frequencies from 100 kHz to 1 GHz is approximately 60 dB.

12.3 Coupling Transformer

The coupling transformer is usually used to provide isolation of the power line fromthe main voltage. It should be connected parallel instead of with a serial connection,because the high current through the transformer can cause magnetic saturation ofthe transformer core. The values of R and C and the ratio of the transformer (Figure12.4) depend on the data rate, power, and signal frequency.

In industry, coupling transformers are used for ADSL, HDSL, VDSL, cables,and modems. Coupling transformers can prevent RF energy from entering theacoustic cables and are used in the frequency range from 30 Hz to 250 kHz.

12.4 Parallel Plate for Susceptibility Test

The parallel plate is used for radiated susceptibility tests from transient electromag-netic fields. The susceptibility of an EUT is the ability to withstand transient electro-magnetic fields. The parallel plate is shown in Figure 12.5. Its physicalcharacteristics are width, w, and height, d.

Characteristic impedance Z0 depends on the frequency, f, resistance, R, induc-tance, L, conductance, G, and capacitance, C, per unit length as shown:

ZR j fL

G j fC0

22

= ++

π

π(12.1)

where

12.3 Coupling Transformer 217

C R

L1 L2 OutputInput

N :N1 2

Figure 12.4 Coupling transformer.

Conductor

Conductor

Dielectricd

w

Figure 12.5 Parallel plate.

Rcond

= 2ωσ δ

(12.2)

Ld= µω

(12.3)

Gddiel= σω

(12.4)

Cd

= εω

(12.5)

σcond and diel are the conductivities of the conductor and dielectric; µ and ε arepermeability and permittivity of the dielectric. Skin depth, δ, is defined as:

δπ µσ

= 1

f cond

(12.6)

The parallel plate should be designed to have an impedance of 50Ω.The setup for testing susceptibility using the parallel plate is shown in Figure

12.6. Beside the plate, other necessary instruments include: a high voltage probe, atransient generator (ordinary monopulse), and a storage oscilloscope with a200-MHz minimum single shot bandwidth and variable sampling rate up to 1 Gsa/s.

12.5 Coupling Clamps and Probes

Coupling clamps and probes are used in EMC testing, especially in emission testing.Emission testing is used to establish how much a certain device is emitting (or radi-ating). Immunity test are used to establish how immune the device is to interference


Transientgenerator

Parallel plate line

Sensor

Load

High voltageprobe

Oscilloscope

Shieldedenclosure

Figure 12.6 Susceptibility setup with parallel plate line.

or outside radiation. Therefore there are instruments which measure the emissionthrough capacitive or inductive coupling. They are usually not used for injecting theinterference. The injection clamps or probes will be dealt with in the followingsection.

12.5.1 Capacitive Coupling Clamp

The capacitive coupling clamp is an instrument used for measuring conducted elec-tromagnetic interference where there is no galvanic connection between the cou-pling clamp and the measured cables. The work principle of a coupling clamp isbased on the capacity coupling that can exist between the measured cable and theclamp; it is used in testing cable resistance to fast transient sources like arcing oncable connectors when connecting the voltage network or ESD. If there is a sourceof continuous electromagnetic waves, the coupling clamp may be used for measur-ing the cables resistance. The capacitive coupling clamp conforms to the require-ments of ISO 61000-4-4. It guarantees that tests are carried out in strict compliancewith the standard. A coupling clamp with its physical characteristics is shown inFigure 12.7.

The clamp allows fast nanosecond pulse bursts (ISO 3a and 3b) to be injectedin cable runs. The characteristic impedance of the unit is 50Ω. The couplingclamp is fitted with appropriate BNC connectors at both sides and is connected tothe generator via a coaxial cable. The far side of the clamp has to be terminatedwith a 50Ω load resistor. It also provides a measurement output via a 40-dBattenuator.

The coupling clamp can test cables of up to a 40-mm diameter and is actually adistributed capacitance. The effective coupling capacitance depends on the crosssection and the material of the cable used, a typical value being around 100 pF.

Inside the capacitive coupling clamp, where the cable is placed, an electric fieldexists; this results in capacity coupling. The time varying electrical field of an exter-nal system produces time varying charges in the disturbed system.

For the capacitive coupling probe built at the Faculty of Electrical Engineeringand Computing, Zagreb (Figure 12.8), the maximum coupling frequency is around100 MHz, as it is with most models.

12.5 Coupling Clamps and Probes 219

1050 mmInsulating supports

High voltagecoaxial connector

High voltagecoaxial connector

140 mm

70m

m70m

m

1000 mm

Coupling plates

100m

m

Figure 12.7 Capacitive coupling clamp dimensions.

12.5.2 Current Probe

The current probe is a precision EMI measuring sensor, which clamps onto a wire,coaxial line, or cable carrying intentional or interference current. The diameterdimension is 5–10 cm. It is used to measure the current in a single wire, wire pair,coax, or bundle. Current probes are used on frequencies from 5 Hz to 1.2 GHz.

Current probes are an excellent diagnostic tool, especially for locating andquantifying ground loops. When measuring the currents in cables or wires, the wireshaving the highest current often point to a solution, which can be: breaking groundloops or increasing their impedance by isolating or floating, applying ferrites, usingbypass capacitors, applying a single-end grounded shield, or using filteredconnectors.

An important characteristic of current probes is transfer impedance. It is usedfor calibration. Figure 12.9 shows indirect measuring of the unknown current bymeasuring the voltage developed across a 50Ω load placed on the current probe’scoaxial cable (or input impedance of a spectrum analyzer).

The computed unknown current, IdB µA, in units of dB µA equals the measuredvoltage, VdB µV, in units of dB µV minus the probe’s transfer impedance in units of dB(dB above an Ω per meter length):

I V ZV TdB A dB dBµ µ= − Ω (12.7)

The spectrum analyzer/EMI receiver can measure up to several mA of current.The principle is shown in Figure 12.10.

There is a magnetic field around the wire through which the current flows. Theferrite core of the current probe concentrates this flow. At the current probe output,a voltage depending on permeability, ferrite core cross section, and the number ofcoils will appear:

V k ANfIout in= µ (12.8)


Figure 12.8 Capacitive coupling clamp built at FER, Zagreb.

Figure 12.9 Current probe.

where Vout is the output voltage, k is the constant, µ is the core permeability, A is thecore cross section, N is the coil number, f is the frequency, and Iin is the wire current.Toroidal ferrite concentrates the field around the wire of EUT using the test coil.When constructing a current probe, it is necessary to install electrostatic shielding toprevent capacitive coupling between the coil and the EUT wire.

12.6 Injection Clamps and Probes

Beside emission tests, there are also immunity tests in which EUT is tested with out-side interference. The instruments for introducing interference into the EUT arecalled injection clamps or probes. They use capacitive or inductive coupling. Themethod is usually quite simple but includes large losses. It is used only when LISNsare not available.

12.6.1 Current Injection Probe

The current injection probe is a method for injecting interference in immunity test-ing. It is simple, but relatively ineffective. It has high coupling losses (i.e., largepower is necessary, and the results are hardly repeatable). It is recommended only ifno other method is available or practical.

As a transformer, the probe introduces only inductive coupling with no capaci-tive coupling (Figure 12.11).

There is no isolation from other equipment, which is a serious drawback. Thevoltage on the EUT will depend on cable resonances at higher frequencies. Further-more, the parasitic capacitance between the probe and the cable will influence thelocal cable impedance. Thus, even though it is not necessary to ground the probe forthe coupling, it is useful to ground its casing to the ground reference plane to lowerthis effect. The probe has losses of about 5 dB. It is used in the frequency range of upto 400 MHz. Usually a high power amplifier (200W) is necessary to perform thetests.

12.6.2 EM Clamp

The EM clamp (Figure 12.12) is a tube made of slotted ferrite rings, which can beconnected to a cable under test. It is not invasive and can be used on any cable type.

12.6 Injection Clamps and Probes 221

Vout

H r= 1/2π

Output to EMIreceiverMagnetic field

around the wire

Current through wire

I in I in

Test coil

Figure 12.10 Cable current principle of work.

There are types with both inductive and capacitive coupling, which are used in thefrequency range of 150 kHz to 1 GHz. However, this method is not as good asLISN. The losses are not high and it is not necessary to use a high power amplifier asis the case with the current injection probe. It is desirable to ground the clamp forbetter repeatability of test results. The clamp (ferrites) can also be used as a couplingclamp for emission tests, in which the clamp absorbs interfering signals from thecable. The setback is that for every testing frequency it is necessary to move theclamp along the cable, which can take a lot of time. Tests are almost impossible toperform automatically and have to be done manually.

Table 12.1 shows the necessary power (in watts) for achieving 3 V/m and 10V/m for different methods of injecting interference into the EUT. The best are theLISN and EM clamps; the current injection probe is recommended only if there areno other methods available.


Figure 12.12 EM clamp.

GeneratorGround reference plane

Auxiliary equipment

Inductivecoupling EUT

Current injection probe

Figure 12.11 Injecting interference using a current injection probe.

12.6.3 Electrostatic Discharge (ESD) Generator

Electrostatics discharge from the human body to a device or discharge between twodevices can lead to interference or destruction of sensible electronic devices. Thegenerated voltages can be up to several kV. Electrostatic discharge is characterizedby a fast rise time (1 nanosecond), and intense discharge from humans, clothing,furniture, and other charged dielectric sources. The discharge resistance of humansmay vary from several hundred ohms to 10 kohms.

There are portable generators for testing ESD (IEC 1000-4-2, EN 61000-4-2),which can generate 50,000 pulses of 16.5 kV in the air and 10 kV in contact. Theysimulate ESD from a human or furniture. An ESD gun or generator is ahand-gun-shaped instrument containing a capacitor (typically about 150 pF—simu-lating a human), which can be charged up from 1 kV to 15 kV (sometimes more).One or more discharge resistors (approximately 1,500 ohms) and pulse-shapingnetworks acheive the undesired output waveform.

The waveform (Figure 12.13) is characterized by a subnanosecond rise time anda current value of a few amperes. Figure 12.14 shows an equivalent circuit of the

12.6 Injection Clamps and Probes 223

10%

90%

0.7–1 ns

30 ns 60 ns

I at 30 ns

I at 60 ns

Imax

Figure 12.13 Waveform of an ESD generator.

C

R

Figure 12.14 ESD generator circuit (R = 330Ω, C = 150 pF).

Table 12.1 Necessary Power (W)

LISN EM ClampCurrentInjection Probe

3V 10V 3V 10V 3V 10V

10 kHz — — — — 585 6,500

150 kHz 0.59 6.5 1.46 16.25 29.32 325.78

27 MHz 0.59 6.5 0.94 10.4 4.21 46.8

80 MHz 0.59 6.5 0.59 6.5 4.62 51.35

230 MHz 0.59 6.5 0.59 6.5 5.85 65

generator. The resistance-capacitance network would produce a zero rise time. Forgenerating the waveform, the actual ESD gun has a series inductance (due to thereturn current ground cable), which is responsible for the finite rise time in the cur-rent waveform.

For the ESD experiment (Figure 12.15), the discharge could not be performeddirectly on the PCB trace because the gun-radiated field would have coupled to thetraces, which could not be modeled.

The model considers an impulsive transverse electromagnetic mode (TEM)wave reaching the victim trace where the suppressor is mounted. Although this con-dition can be realized by connecting a coaxial cable to the PCB and performing thedischarge at the beginning of the cable, it must be done inside a shielded room toavoid propagation of the gun-radiated field. The PCB traces are then connected to a1-GHz oscilloscope to capture the structure’s response to the ESD. The correspond-ing response time of the oscilloscope should be 0.35 nanosecond—sufficiently lowerthan the rise times.

12.7 EMI Receiver

The EMI receiver measures conducted emissions using an LISN or current probe,and radiated emissions using antennas according to international standards (CISPR16-1). They measure RF signals with high accuracy. At the input, they have tunablepreselectors for overload protection, bandwidth selection, tuning capability, andseveral detector functions. Tunable preselectors are passive filters that split pulsedsignals into different frequency bands. At low frequencies these filters increase thesize and weight of the EMI receiver in comparison to the spectrum analyzer. Mea-surements are performed on one frequency at a certain time, but some models canbe programmed to sweep the frequency spectrum and record the resultsautomatically.

For covering the frequency range of 20 Hz to 40 GHz, at least three receivers arerequired. The most common digital interfaces are: RS232, Centronics, Ethernet,IEC-Bus (IEC625-2/IEEE 488-2), PS2-Keyboard, PS2-Mouse, USB, Userport, andVGA connectors. Impulse generators are used for broadband calibration of EMIreceivers. However, most EMI receivers have internal built-in calibrating impulsegenerators.

The main specifications of the EMI receiver include: interference, selectivity andshape factor, adjacent-channel interference, spurious response, intermodulation,cross-modulation, quasi-peak, peak and average value, frequency range at −6 dB,


PC OscilloscopePCB

Shielded room

ESDgenerator

Figure 12.15 ESD measurement setup.

time constants of charging and decharging, selectivity, intermodulation productlimitation, noise limitation, and accuracy.

The main advantages of an EMI receiver versus a spectrum analyzer are bettersensitivity, durable circuitry, higher accuracy at measuring frequency, and ampli-tude, as well as higher dynamics.

12.8 Spectrum Analyzer

Spectrum analyzers are less expensive than EMI receivers and usually do not havean RF preselector. They have a higher noise figure (less sensitivity), less dynamicrange for broadband emissions (such as an impulse generator), fewer detector func-tions, and less shielding in the case housing. Low-noise RF preamplifiers, preselec-tors, and CISPR quasi-peak adapters are optional. The spectrum analyzer usuallyhas a tracking generator, several half-decade bandwidth selections, scan rate/band-width interlocks, and data storage for statistical manipulation.

Similar to EMI receivers, spectrum analyzers measure the frequency and ampli-tude of electromagnetic signals in the frequency domain, but are used for differentpurposes. EMI receivers are used for distortion-free measurement of noncontinuousRF signals, while spectrum analyzers are used for high frequency, fast sweep, andcontinuous signals. EMI receivers respond properly to pulsed signals, which wouldoverload a spectrum analyzer’s input circuitry (especially those with a low pulserepetition rate). The advantage of spectrum analyzers is that they can be very small,which is useful for design and diagnostic purposes.

Cheap models have fewer functions and are used only for precompliance test-ing, while more sophisticated models can be used for the final compliance testing.With a current probe, the spectrum analyzer can be used to measure conductedemissions.

12.9 Oscilloscopes

An oscilloscope is a test instrument that displays waveforms of electronic and elec-trical circuits. In the past, oscilloscopes consisted of a cathode-ray tube and compo-nents that directed the electron beam based on the voltage of the input signal(vertical) and the scan produced by a time base (horizontal). Today most oscillo-scopes are digital. A digital oscilloscope converts the analog input signal into digitalform (a series of binary numbers), which is then displayed or stored in memory. Theability to store waveforms is especially important for viewing one-time events.

The oscilloscope is usually used with its probe. Probes usually have built-in 10:1attenuators. More sophisticated oscilloscopes have broadband amplifiers for mea-suring voltage in submillivolt levels.


C.I.S.P.R., “Specification for Radio Disturbance and Immunity Measuring Apparatus and Meth-ods,” International Electrotechnical Commission, Geneva, Switzerland, 1999.

12.8 Spectrum Analyzer 225

Cormier, B., and W. Boxleitner, “Electrical Fast Transient (EFT) Testing—An Overview,” Proc.IEEE International Symposium on Electromagnetic Compatibility, August 12–16, 1991, pp.291–296.De Leo, R., F. Moglie, and V. M. Primiani, “Analyzing ESD Transient Suppressors in Printed Cir-cuits,” Compliance Engineering, 2001.Dipak, L., D. L. Sengupta, and V. V. Liepa, Applied Electromagnetics and Electromagnetic Com-patibility, New York: Wiley-Interscience, 2006.Morgan, D., A Handbook for EMC Testing and Measurement, London, U.K.: Peter PeregrinusLtd., 1994.Kaires, R.G., “Stopping Electromagnetic Interference at the Printed Circuit Board,” Conformity,November 2003, pp. 12–21.Radman, S., I. Bacic, and K. Malaric, “Capacitive Coupling Clamp,” International Conference onSoftware, Telecommunications and Computer Networks, SOFTCOM, Split, CD-ROM, 2008.Reinhold, L., and P. Bretchko, RF Circuit Design: Theory and Applications, Upper Saddle River,NJ: Prentice-Hall, 2000.Sakulhirirak, D., V. Tarateeraseth, and W. Khanngern, “The Analysis and Design of Line Imped-ance Stabilization Network for an In-House Laboratory,” Proc. 2006 4th Asia-Pacific Confer-ence on Environmental Electromagnetics, August 2006, pp. 232–234.Sklar, B., Digital Communications: Fundamentals and Applications, Upper Saddle River, NJ:Prentice-Hall, 2001.Smith, D., “Current Probes, More Useful Than You Think,” Proc. 1998 IEEE International Sym-posium on EMC, 1998, pp. 284–289.


C H A P T E R 1 3

Control of Measurement Uncertainty

13.1 Evaluation of Standard Uncertainty

Measurement uncertainty is a parameter associated with the result of a measure-ment that characterizes the dispersion of the values that could reasonably be attrib-uted to the measurand. Uncertainty expresses doubt about the result of ameasurement. The real value of a measured quantity can never be knownexactly—it can only be estimated. The true value lies inside the uncertainty intervalwith a certain degree of probability (level of confidence). For most cases the suffi-cient level of confidence is 95%, obtained with coverage factor k = 2 (for k = 1 thelevel of confidence is 68%—one standard deviation). The evaluation of standarduncertainty is defined either by statistical analysis of a series of observations (i.e.,repeatable measurements) (type A), or by systematic components of uncertainty(type B). For combining uncertainty components of the measurement, a probabilitydensity function (pdf) must be chosen for each uncertainty component. If the uncer-tainty component has random errors (electrical noise, connector repeatability), thenthe pdf usually has a normal (Gaussian) distribution. For systematic errors (fromthe manufacturer’s data sheet) rectangular (uniform) distribution is used, whileuncertainties in measurements at microwave frequencies (phase influence) are bestdescribed with a U- shaped distribution.

13.1.1 Type A Evaluation of Standard Uncertainty

When evaluation of uncertainty is done by statistical analysis of a series of observa-tions, then it is called a type A evaluation. In this type of evaluation, standard uncer-tainty of a measurand is calculated from a series of repeated observations. Even iffor some measurements the random component of uncertainty is not relevant toother contributions of uncertainty, it is desirable to establish the scale of randomeffects on the measurement process. The average or mean value of the measure-ments should be calculated. If there are n independent repeated values for a quantityQ, then the mean value q is obtained by

qn

qq q q q

njn

j

n

= =+ +

=∑1 1 2 3

1

(13.1)

227

The measured results will be spread over a certain range, which depends on var-ious factors, such as: measurement method, measuring device used, and even theperson making measurements. The resulting dissipation is defined as standard devi-ation σ

( )σ = −=

∑1 2

1nq qj

j

n

(13.2)

The above expression gives the standard deviation σ for the particular set of nmeasurements. If the process is repeated at some later time, with a different numbern of measurements, different values of q and σ will be obtained. For a very large n,the mean value will approach the central limit of the distribution of all possible val-ues. The probability density function will have a normal distribution. From theresults of a single set of measurements and their standard deviation, σ, an estimatefor all possible values of the measurand s(qj) can be made with

( ) ( )s qn

q qj jj

n

=−

−=∑1

1

2

1

(13.3)

The mean value q is obtained from a finite number n of measurement results;the mean value is not the exact mean that would have been obtained if an infinitenumber of measurements could have been taken. Therefore, even the mean valuehas its uncertainty, which is called the standard deviation of the mean. Its value canbe obtained from the estimated standard deviation using

( ) ( )s q

s q

n

j= (13.4)

13.1.2 Type B Evaluation of Standard Uncertainty

The type B evaluation is defined as all uncertainty other than repeatable measure-ment uncertainty. It is associated with systematic errors. The evaluation of thesecontributions depends on previous experience, the measurement process, manufac-turer specifications, calibration data, and the environment.

When all possible systematic components of uncertainty are identified, proba-bility distributions should be assigned to them. Although probability distributionscan be of any form, the most common ones for the type B evaluation of standarduncertainty are the rectangular (uniform) and U-shaped distributions.

13.2 Distributions

The three most common distributions for the evaluation of uncertainties are normal(Gaussian), rectangular, and U-shaped. While normal distribution is used for thetype A evaluation, rectangular and U-shaped distributions are used for the type Bevaluation.

228 Control of Measurement Uncertainty

13.2.1 Normal (Gaussian) Distribution

Normal distribution is shown in Figure 13.1. The distribution size is described withthe standard deviation. The shaded area represents 1 standard deviations from thecenter of distribution. This is approximately 68% of the area under the curve. Thismeans that for coverage factor k = 1 there is a 68% probability that the measure-ment value is in this range. Table 13.1 shows the coverage probability versus cover-age factor. In some situations it is necessary to use a higher coverage factor for ahigher probability. Usually p = 95% (k = 1.96 or 2.00) is enough.

The values xi of the input quantities Xi all have their uncertainties, u(xi), whichare called standard uncertainties. Standard uncertainty for a normal distribution isequal to the standard deviation of the mean [i.e., s q( )]:

( ) ( )u x s qi = (13.5)

13.2.2 Rectangular Distribution

A rectangular distribution (Figure 13.2) is used for a measuring instrument with anaccuracy of ±x or ±dB without any statistical information. The result may lie any-where between −x to +x with equal probability. Outside of this range the probabil-ity of xi is zero. This distribution is used for type B evaluations. For someinstruments the resolution will be a = 0.5 of the least significant digit. When there isno previous knowledge about the measurement quantity, rectangular distributionmust be used. Standard uncertainty for a rectangular distribution is calculated from

13.2 Distributions 229

68%

Figure 13.1 Normal (Gaussian) probability distribution.

Table 13.1 CoverageProbability Depending onthe Coverage Factor

CoverageProbability p (%)

CoverageFactor k

68% 1.00

90% 1.64

95% 1.96

95.45% 2.00

99% 2.58

99.73% 3.00

( )u xa

ii=3

(13.6)

13.2.3 U-Shaped Distribution

Since measurements at microwave frequencies often involve vector quantities (bothmagnitude and phase), it is sometimes necessary to use a U-shaped distribution (Fig-ure 13.3). This distribution is most commonly used for the RF mismatch uncer-tainty where the phase information for a given vector is unknown. Mismatchuncertainty occurs when there is no perfect matching of impedance between thesource and load (termination). If the phase is unknown, the cosine function willdetermine the probability distribution. With this function, xi will probably be closerto one of the edges of the distribution rather than in the center. Standard uncertaintyfor a rectangular distribution is calculated from

( )u xa

ii=2

(13.7)

13.2.4 Combined Standard Uncertainty

When different input uncertainties are combined, the normal distribution will beused. Since normal distribution is described with standard deviation, all inputuncertainties have to be evaluated and combined with their sensitivity coefficients


a a

x a−i x a+ixi

Probabilityp

Figure 13.2 Rectangular distribution.

x a−i x a+ixi

Figure 13.3 U-shaped distribution.

to form the normal distribution. The standard uncertainties, xi, and their sensitivitycoefficients, ci, will give a single value to be associated with y of the measurand Y.The combined standard uncertainty will then be:

( ) ( ) ( )u y c u x u yc i ii

m

ii

m

= ≡= =∑ ∑2 2

1

2

1

(13.8)

When one contribution dominates, the resulting contribution will be very simi-lar to the dominating one. With the excepting of a few cases, the resulting contribu-tion will usually be normal, no matter what the contribution distribution is.

13.2.5 Expanded Uncertainty

When purchasing measurement equipment, a calibration certificate usually quotesan expanded uncertainty, U, with a high coverage probability. Using coverage fac-tor k, the standard uncertainty can be calculated as follows:

( )u xUki = (13.9)

13.3 Sources of Error

There are two types of measurement errors: random and systematic. Random errorsare evaluated in type A evaluations and are normally distributed as shown above.Systematic errors are evaluated in type B evaluations and can shift the mean value(probable value) and add an uncertainty.

13.3.1 Stability

All instrument performance changes with time; the value of resistors changes andmicrowave attenuators drift, which is evaluated by calibration. The drift is mostlikely not linear. The data over time should be displayed graphically and the mostprobable value selected.

13.3.2 Environment

Temperature and humidity can affect the performance of attenuators, power sen-sors, and other equipment. Therefore, measurements should be performed in labo-ratory conditions with defined temperature and humidity.

13.3.3 Calibration Data

Usually, calibration points are limited. Sometimes quantity values other than thecalibration point have to be measured, at which systematic errors can occur. If pos-sible, calibration should be performed with some other calibration instrument, orthe value prediction can be made from other similar models’ data.

13.3 Sources of Error 231

13.3.4 Resolution

Another systematic error of the measuring device is the digital rounding error. Aquantization error of ±0.5 is present, since the measured value is converted fromanalog to digital. Noise in the system can cause fluctuations of the last digit as well.

13.3.5 Device Positioning

The position between the measuring instrument and the device under test can alsolead to systematic error. There could be leakage currents to Earth as well as electro-magnetic leakage fields. Mutual heating can be avoided by placing the instrumentsfarther apart.

13.3.6 RF Mismatch Error

Characteristic impedance mismatch of the measurement transmission line is one ofthe most common systematic errors in power and attenuation measurementsbecause the phases of voltage reflection are usually unknown, making it hard tomake the corrections.

13.4 Definitions

The terms described in this chapter are given in the “ISO Guide to the ExpressionSystem of Uncertainty Measurement,” “IEEE Std 100-1988,” and in “InternationalVocabulary of Basic and General Terms in Metrology.” For definitions, please seethe Glossary.


Agilent Technologies, “Application Note 64-1B, Fundamentals of RF and Microwave PowerMeasurements Classic Application Note on Power Measurements,” 2000.Bronaugh, E., and J. Osburn, “A Process for the Analysis of the Physics of Measurement andDetermination of Measurement Uncertainty in EMC Test Procedures,” Proc. IEEE 1996 Interna-tional Symposium on Electromagnetic Compatibility, August 19–23, 1996, p. 245.“CISPR 16-4-2 (Ed.1.0). Specification for Radio Disturbance and Immunity Measuring Appara-tus and Methods—Part 4-2: Uncertainties, Statistics and Limit Modeling—Uncertainty in EMCMeasurements,” IEC Standard, November 2003, p. 43.Heise, E. R., and R. E. W. Heise, “A Method to Calculate Uncertainty of Radiated Measure-ments,” Proc. IEEE 1997 Symposium on Electromagnetic Compatibility, August 18–22, 1997,p. 359.“International Vocabulary of Metrology—Basic and General Concepts and Associated Terms(VIM),” ISO/IEC Guide 99, 2007.Kurosawa, T., et al., “Study on Measurement Uncertainty in Immunity Testing: IEC61000-4-6,”Proc. Electromagnetic Compatibility and 19th International Zurich Symposium on Electromag-netic Compatibility, May 19–23, 2008, pp. 598–601.Ridler, N., et al., “Measurement Uncertainty, Traceability, and the GUM,” IEEE MicrowaveMagazine, Vol. 8, No. 4, August 2007, pp. 44–53.


Taylor, B. N., and C. E. Kuyatt, Guidelines for Evaluating and Expressing the Uncertainty ofNIST Measurement Results, NIST Tech. Rep. TN1297, Gaithersburg, MD, 1994.

13.4 Definitions 233

Appendix ACommunication Frequency Allocations

A.1 Frequency Allocation in the United States

218–219-MHz Radio Service

30 MHz–300 MHz 300 MHz–3 GHz 3 GHz–30 GHz 30 GHz–300 GHz

218–2,190 — — —

700-MHz Guard Service (Digital TV)


— 746–747762–764776–777792–794

— —

3650–3700-MHz Radio Service


— — 3.650–3.700 —

Access Broadband over Power Line (BPL)


30–80 — — —

Advanced Wireless Services Including 3G


— 1,710–1,7551,915–1,9201,995–2,0002,020–2,0252,110–2,180

— —

235

Amateur Radio


50–54144–148219–220222–225

420–450902–9281,240–1,3002,300–2,3102,390–2,450

3.3–3.55.650–5.92510.0–10.524.0–24.25

47.0–47.276–81122.25–123.0134–141241–250275–300

Auditory Assistance Devices


72–7374.6–74.875.2–76216.75–217

— — —

Automatic Vehicle Identification Systems


— 2,900–3,000 3.0–3.263.267–3.3323.339–3.34583.358–3.6

—

Auxiliary Broadcasting


54–7276–108152.855–154157.45–161.575161.625–161.775162.0125–173.2174–216

450–454455–456470–608614–806944–9602,025–2,1102,450–2,483.5

6.425–6.5256.875–7.12512.7–13.2517.7–18.319.3–19.7

76–81

Aviation/Aeronautical

30 MHz–300 MHz 300 MHz–3,000 MHz3 GHz–30 GHz 30 GHz–300 GHz

72–7374.6–75.2108–137156.2475–157.0375

328.6–335.4849–851894–896960–1,2151,300–1,3501,435–1,5251,535–1,660.52,310–2,3202,345–2,3952,700–3,000

3.5–3.654.2–4.45.00–5.255.35–5.469.0–9.213.25–13.415.4–15.724.75–25.05

32.3–33.4

236 Appendix A

Basic Exchange Telephone Radio Service


152–159 450–460816–820861–865

— —

Biomedical Telemetry Devices


174–216 470–668 — —

Broadband Radio Service/Educational Broadband Service


— 2,495–2,690 — —

Cable TV Relay


— 2,025–2,110 6.425–6.5256.875–7.12512.7–13.2517.7–18.319.3–19.7

—

Cellular Services


— 824–849869–894

— —

Dedicated Short-Range Communication Services


— — 5.85–5.925 —

Digital Audio Broadcasting


— 1,452–1,4922,310–2,360

— —

Differential GPS


108–117.975 1,559–1,610 — —

Appendix A 237

Family Radio Service


— 462.5625–467.7125 — —

Field Disturbance Sensors


— 902–9282,435–2,465

5.785–5.81510.500–10.55024.075–24.175

57–64

Fixed Microwave


— 928–929932–935941–9601,850–2,0002,110–2,1802,450–2,483.5

3.7–4.25.925–6.87510.55–10.6810.45–10.6810.7–11.712.2–13.2517.7–18.319.3–19.721.2–23.624.25–24.4525.05–25.2527.5–29.5

31–31.337–39.542–42.571–7681–8692–9494.12–95

Fixed Satellite


— 1,390–1,3921,430–1,432

3.6–4.24.5–4.85.15–5.255.85–7.0757.25–7.757.9–8.410.7–12.212.7–13.2513.75–14.515.43–15.6317.3–21.224.75–25.2527.5–30

30–3137.5–4242.5–45.547.2–50.250.4–51.471–7681–86123–130158.5–164167–174.5209–226232–240265–275

238 Appendix A

FM Broadcasting


88–108 — — —

General Aviation Air-Ground Service


— 454.675–454.975459.675–459.975

—

General Mobile Radio Service


— 462–467 — —

Industrial/Business Radio Pool


25–5072–76150–174216–220

406–413421–430450–470470–512800900

3.4–3.6 —

Industrial, Scientific, and Medical Applications (ISM)


40–42 902–9282,400–2,500

5.65–5.92524.0–24.25

59.3–64116–123241–248

Location and Monitoring Service


— 902–928 — —

Low Power Radio Service


216.75–217 — — —

Lower 700 MHz Services (Digital TV)


— 698–746 — —

Appendix A 239

Maritime


154–161.625161.775–173.2216–220

454–455456–4601,525–1,5591,626.5–1,6602,900–3,000

3.0–3.15.47–5.659.2–9.3

—

Millimeter Wave 70-80-90 GHz Service


— — — 71–7681–8692–9494.1–95

Mobile Satellite


137–138148–150.05

399.9–400.05400.15–401406–406.11,525–1,5591,610–1,660.52,000–2,0202,180–2,2002,483.5–2,500

7.25–7.757.9–8.414–14.519.7–21.229.5–30

30–3139.5–4143.5–4750.4–51.466–7481–84123–130158.5–164191.8–200252–265

Multiple Address Systems


— 928–960 — —

Multiuse Radio Service


151.82151.88151.94154.57154.60

— — —

240 Appendix A

Offshore Radiotelephone Service


— 476–478479–481482–484485–487488–490491–493

— 40.5–43.5

Paging


35–3643–44152–159

454–460929931

— —

On–Site Paging


47.0–47.25 440–470 — —

PCS Broadband


— 1,850–1,9901,930–1,990

— —

PCS Narrowband


— 901–902930–931940–941

— —

PCS Unlicensed


1,920–1,930

Petroleum Radio Service


36.2541.71150.980154.585158.445159.480

454459

— —

Appendix A 241

Private and Public Land Mobile


30.56–3233–3435–3637–3839–46.647–49.672–7374.6–74.875.2–76150.8–156.2475157.0375–173.4216–222

406.1–455456–462.5375462.7375–467.5375467.7375–512698–901902–930931–940941–9601,427–1,4322,210–2,1802,450–2,483.52,900–3,000

3.0–3.74.94–4.995.25–5.655.85–5.92510.0–10.5513.4–1415.7–17.324.05–24.25

33.4–36

Public Safety Services


— 764–776794–806

— —

Radio Astronomy


37.50–38.2573–74.6

406.1–410608–6141,400–1,4271,610.6–1,613.81,660–1,6702,655–2,700

4.99–5.010.68–10.715.35–15.422.21–22.523.6–24

31.0–31.842.5–43.576–116123–158.5164–167182–185200–217226.0–231.5241–275

Radio Control Radio Service


72–7375.4–76

— — —

242 Appendix A

Space Operation/Space Research


137–143.65 400.15–402410–4201,215–1,3001,400–1,4291,660.5–1,668.41,755–1,8502,025–2,1202,200–2,3002,655–2,700

3.1–3.34.99–5.05.25–5.577.145–7.2358.4–8.58.55–8.659.5–9.810.6–10.713.25–14.214.5–15.416.6–17.117.2–17.318.6–18.821.2–21.422.21–22.523.6–2425.5–27

31.3–32.334.2–34.735.5–3840–40.550.2–50.452.6–59.365–6674–8486–9294–94.1100–102105–122.25148.5–151.5155.5–158.5164–167174.8–191.8200–209217–231.5235–238250–252

Specified Mobile Radio Service


— 809–824854–869896–901935–940

— —

Standard Frequency and Time Signal Satellite


— 400.05–400.15 13.4–14.020.2–21.225.25–27

30–31.3

Television (Digital)

30 MHz – 300 MHz 300 MHz – 3 GHz3 GHz – 30 GHz 30 GHz – 300 GHz

54–7276–88174–216

470–608614–6982,025–2,110

— —

Ultrawideband (UWB)


— Below 960 1.99–10.6 —

Appendix A 243

Unlicensed National Information Infrastructure (U–NII) Devices


— — 5.15–5.355.47–5.825

—

Vehicle Radar Systems


— — 16.2–17.723.12–29

46.7–46.976–77

Weather Instruments/Radar/Satellites


137–138 400.15–406460–4701,668.4–1,7102,700–3,000

5.6–5.657.75–7.858.175–8.2159.3–9.59.975–10.025

33.4–34.5

Wireless Communication Service


— 2,305–2,3202,345–2,360

— —

Wind Medical Telemetry


— 608–6141,395–1,4001,427–1,432

— —

Wireless Microphones


54–7276–88169.445169.505170.245170.305171.045171.105171.845171.905174–216

470–608614–806

— —

244 Appendix A

A.2 International Frequency Allocation

Active Sensors (Satellite)


30.0–37.5 402–406 — —

Aeronautical Radio Navigation

30 MHz–300 MHz 300 MHz–3,000 MHz 3 GHz–30 GHz 30 GHz–300 GHz

74.8–75.2108–117.975

328.6–335.4960–1,2151,300–1,3501,559–1,626.52,700–3,000

3.0–3.14.2–4.45.00–5.158.50–10.013.25–13.415.4–15.7

—

Amateur Applications


50–52144–146

430–4401,240–1,3002,300–2,450

3.4–3.55.650–5,83010.0–10.524.0–24.25

47.0–47.247.5–47.948.2–48.5475.5–81.5122.25–123.0134–141241–250

Amateur Satellite


144–146 434.79–4381,260–1,2702,400–2,450

5.650–5.7255.830–5.85010.45–10.524–24.05

47–47.275.5–81.5122.25–123134–141241–250

Analog/Digital Land Mobile Radio


30.01–40.6640.7–74.875.2–87.5146–156157.45–160.60160.975–161.475162.05–174.0

385–390395–399.9406.1–430440–470870–876915–921

— —

Appendix A 245

Automotive Short Range Radar (SRR)


— — — 76–81

Broadband Mobile Systems


— — — 40–40.542.5–43.562–6365–66

Defense Systems


29.7–74.875.2–87.5138–144230–242.95243.05–300

300–328.6335.4–399.9790–890915–9351,215–1,4001,427–1,4521,492–1,5251,660–1,6701,675–1,7102,025–2,1102,200–2,2902,520–2,6552,900–3,000

3.00–3.404.40–5.005.25–5.857.25–8.4013.4–1414.50–15.3515.7–17.724.05–24.2526.5–27.5

33.4–39.543.5–45.559.0–64.071–7481–84

Digital Enhanced Cordless Telephony Systems (DECT)


— 1,880–1,900 — —

Distress Signals


156.5125–156.5375156.7625–156.8375

1,544–1,545 — —

Equipment for Detecting Movement and Alert


— 2,400–2,483.5 9.2–1010.5–10.613.4–1424.05–24.25

33.4–35.2

246 Appendix A

Feeder Links


— — 5.15–5.256.925–7.07517.3–18.427.5–29.5

47.2–49.44

Fixed Links


— 2,025–2,1102,200–2,2902,483.5–2,5002,520–2,670

5.925–8.510.15–10.3010.45–10.6810.7–11.712.75–13.2514.5–15.3517.7–19.722–22.623–23.624.5–26.527.5–29.5

31–31.331.5–31.837–39.548.2–50.255.78–5964–6671–7681–86

Fixed Satellite


— — 3.4–4.24.5–4.85.15–5.255.725–7.0757.25–7.757.9–8.410.7–11.712.5–13.2513.75–14.515.43–15.6317.3–21.227.5–30

30–3137.5–40.542.5–45.547.2–50.250.4–51.471–7681–86123–130158.5–164167–174.5209–226232–240265–275

FM Radio Broadcasting


87.5–108 — — —

Appendix A 247

IMT (International Mobile Telecommunications)


— 880–915925–9601,710–1,7851,805–1,8801,900–2,0252,110–2,1702,500–2,690

3.4–3.6 —

Industrial, Scientific, and Medical Applications (ISM)


40.66–40.70 433.05–434.792,400–2,500

5.725–5.92524.0–24.25

59.3–62.0

Low Earth Orbiting Satellite


137–138148–150.05

400.15–401 — —

Maritime


156–156.5125156.5375–156.7625156.8375–157.45160.6–160.975161.475–162.05

456–459 — —

Meteorology (Including Satellites)


137–138 400.15–406460–4701,668.4–1,710

7.45–7.557.75–7.858.175–8.21518.1–18.3

35.2–36

Mobile Applications


137–144 1,785–1,8002,290–2,4002,483.5–2,500

3.40–3.604.40–5.08.025–8.215

—

248 Appendix A

Mobile Satellite


121.45–121.55137–138148–150.05242.95–243.05

399.9–400.05400.15–401406–406.11,518–1,5591,610–1,660.51,668–1,6751,980–2,0102,170–2,2002,483.5–2,500

7.25–7.3757.9–8.02510.7–11.714–14.5019.7–21.229.5–30

30–3139.5–40.543.5–4750.4–51.466–74123–130158.5–164191.8–200252–265

Multimedia Wireless Systems


— — — 40.5–43.5

Nonspecific Short Range Device (SRD)


40.66–40.70138.20–138.45

433.05–434.792400–2500

5.725–5.92524.0–24.25

59.3–62.0

On-Site Paging


47.0–47.25 440–470 — —

Passive Sensors (Satellite)


— 1,400–1,4272,690–2,700

4.2–4.44.8–4.996.425–7.2510.6–10.713.75–1415.35–15.418.6–18.821.2–21.422.21–22.523.6–24

31.3–31.836–3750.2–50.452.6–59.386–92100–102116–122.25148.5–158.5164–167174.8–191.8226–231.5235–238250–252

Appendix A 249

Position Fixing


— 432–4381,215–1,300

3.1–3.35.25–5.578.55–8.659.5–9.813.25–13.7524.05–24.25

35.2–36

Public Cellular Networks, GSM


— 455–470880–915925–9601,710–1,7851,805–1,880

— —

Radar and Navigation Systems


— 960–13502,700–3,000

3.0–3.15.0–5.03

—

Radio Astronomy


37.50–38.2570.45–74.880.25–84.6150.05–153.0

406.1–410608–6141,300–1,4001,400–1,4271,610.6–1,613.81,60–1,6702,200–2,2902,655–2,6902,690–2,700

4.80–5.038.215–8.4010.6–10.714.47–15.422–23.5523.6–24

31.0–31.836–3742.5–43.548.54–49.4451.4–52.658.2–5976–116123–158.5164–167182–185200–217226.0–231.5235–238241–275

Radio Frequency Identification (RFID)


— 865–8682,446–2,454

— —

250 Appendix A

Radio Microphones


29.7–47.0174–216

470–862863–8651,785–1,800

— —

Railway Applications


— 876–880921–9252,446–2,454

— —

Satellite Digital Audio Broadcasting (S–DAB)


— 1,479.5–1,492 — —

Satellite TV


— — 11.7–12.5 —

— — 21.4–22 —

Services Ancillary to Programming/Broadcasting (SAP/SAB)


— 470–8622,025–2,1102,200–2,2902,483.5–2,5002,520–2,670

3.4–3.610–10.6822–23.624–24.5

47.2–50.2

Shipborne and VTS Radar


— — 5.25–5.725 —

Appendix A 251

Space Operation/Space Research


48.5–50137–143.65

400–402410–420460–4701,215–1,3001,400–1,4291,525–1,5351,660.5–1,668.42,025–2,1202,290–2,3002,655–2,6702,690–2,700

3.1–3.35.0–5.035.25–5.577.145–7.258.4–8.58.55–8.658.75–1010.6–10.712.75–14.315.35–15.416.6–17.117.2–17.322–2323.6–2425.5–27

31.3–32.334.2–35.235.5–3840–40.550.2–50.452.6–55.7856.9–59.365–6674–7981–8486–9294–94.1100–102105–116122.02–122.25164–167174.8–191.8200–209217–231.5250–252

Terrestrial Digital Audio Broadcasting (TDAB)


174–240 1,452–1,479.5 — —

TV Broadcasting


174–230 470–862 — —

Ultrawideband (UWB)


— — 6–8.50 —

Very Small Aperture Terminal/Satellite News Gathering


— — 12.5–12.75 —

— — 14–14.50 —

Weather Radar


— — 5.25–5.85 —

— — 9.3–9.5 —

252 Appendix A

Wideband Data Transmission Systems


— 2,400–2,483.5 5.15–5.3 —

— — 5.47–5.725 —

— — 17.10–17.30 —

Wind Profiler Radar


46–68 440–450 — —

— 470–608 — —

— 1,270–1,300 — —

Wireless Applications in Healthcare


30.0–37.5 402–406 — —

Wireless Audio Applications


87.5–108 863–865 — —

— 1,795–1,800 — —

Appendix A 253

Appendix BList of EMC Standards RegardingEmission and Susceptibility

B.1 Cenelec

EN 50081-1:1992—Electromagnetic compatibility—Generic emissionstandard—Part 1: Residential, commercial, and light industry.

EN 50081-2:1994—Electromagnetic compatibility—Generic emissionstandard—Part 2: Industrial environment.

EN 50082-1:1998—Electromagnetic compatibility—Generic immunitystandard—Part 1: Residential, commercial, and light industry.

EN 50082-2:1995—Electromagnetic compatibility—Generic immunitystandard—Part 2: Industrial environment.

EN 55014-1:2001—Electromagnetic compatibility—Requirements forhousehold appliances, electric tools, and similar apparatus—Part 1:Emission—Product family standard.

EN 55014-2:1997—Electromagnetic compatibility—Requirements forhousehold appliances, electric tools, and similar apparatus—Part 2:Immunity—Product family standard.

EN 55015:2001—Limits and methods of measurement of radio disturbancecharacteristics of electrical lightning and similar equipment.

EN 55020:2002—Electromagnetic immunity of broadcast receivers andassociated equipment.

EN 55024:1998—Information technology equipment—Immunitycharacteristics—Limits and methods of measurement; Amendment A1:2001 toEN 55024:1998.

EN 55103-1:1997—Electromagnetic compatibility—Product family standardfor audio, video, audiovisual, and entertainment lighting control apparatus forprofessional use—Part 1: Emission.

EN 55103-2:1997—Electromagnetic compatibility—Product family standardfor audio, video, audiovisual, and entertainment lighting control apparatus forprofessional use—Part 2: Immunity.

EN 55104:1995—Electromagnetic compatibility—Immunity requirements forhousehold appliances, tools, and similar apparatus—Product family standard.

255

EN 61000-3-2:2001—Electromagnetic compatibility (EMC)—Part 3-2:Limits—Limits for harmonic current emissions (equipment input current up toand including 16A per phase).

EN 61000-6-1:2001—Electromagnetic compatibility (EMC)—Part 6-1:Generic standards—Immunity for residential, commercial, and light-industrialenvironments.

EN 61000-6-2:2001—Electromagnetic compatibility (EMC)—Part 6-2:Generic standards—Immunity for industrial environments.

EN 61000-6-3:2001—Electromagnetic compatibility (EMC)—Part 6-3:Generic standards—Emission standard for residential, commercial, andlight-industrial environments.

EN 61000-6-4:2001—Electromagnetic compatibility (EMC)—Part 6-4:Generic standards—Emission standard for industrial environments.

EN 61547:1996—Equipment for general lighting purposes—EMC immunityrequirements.

EN 12015:1998—Electromagnetic compatibility—Product family standard forlifts, escalators, and passenger conveyors—Emission.

EN 12016:1998—Electromagnetic compatibility—Product family standard forlifts, escalators, and passenger conveyors—Immunity.

B.2 Australian Standards

AS/NZS 4251.1:1999—Electromagnetic compatibility (EMC)—Genericemission standard—Residential, commercial, and light industry.

AS/NZS 4251.2: 1999—Electromagnetic compatibility (EMC)—Genericemission standard—Industrial environments.

B.3 Canadian Standards

CAN/CSA C108.8-M83 (R2000)—Limits and methods of measurement ofelectromagnetic emissions from data processing equipment and electronic officemachines.

CAN/CSA-C108.9-M91 (R1999)—Sound and television broadcasting receiversand associated equipment—Limits and methods of measurement of immunitycharacteristics.

CAN/CSA-CEI/IEC 61000-4-3-01—Electromagnetic compatibility(EMC)—Part 4-3: Testing and measurement techniques—Radiated,radio-frequency, electromagnetic field immunity test (Adopted CEI/IEC61000-4-3:1995 + A1:1998, Edition 1.1, 1998-11).

CAN/CSA-CEI/IEC 61000-4-4-01—Electromagnetic compatibility(EMC)—Part 4: Testing and measurement techniques—Section 4: Electrical fasttransient/burst immunity test—Basic EMC publication (Adopted CEI/IEC1000-4-4:1995, first edition, 1995-01, including Amendment 1:2000).

256 Appendix B

CAN/CSA-CEI/IEC 61000-4-5-01—Electromagnetic compatibility(EMC)—Part 4: Testing and measurement techniques—Section 5: Surgeimmunity test (Adopted CEI/IEC 1000-4-5:1995, first edition, 1995-02,including Corrigendum October 1995 and Amendment 1:2000).

CAN/CSA-CEI/IEC 61000-4-6-01—Electromagnetic compatibility(EMC)—Part 4: Testing and measurement techniques—Section 6: Immunity toconducted disturbances, induced by radio-frequency fields (Adopted CEI/IEC1000-4-6:1996, first edition, 1996-03, including Corrigendum September 1996and Amendment 1:2000).

CAN/CSA-CEI/IEC 61000-4-8-02—Electromagnetic compatibility(EMC)—Part 4-8: Testing and measurement techniques—Power frequencymagnetic field immunity test (Adopted CEI/IEC 61000-4-8:1993 + A1:2000,edition 1.1, 2001-03).

CAN/CSA-CEI/IEC 61000-4-9-02—Electromagnetic compatibility(EMC)—Part 4-9: Testing and measurement techniques—Pulse magnetic fieldimmunity test (Adopted CEI/IEC 61000-4-9:1993 + A1:2000, edition 1.1,2001-03).

CAN/CSA-CEI/IEC 61000-4-11-01—Electromagnetic compatibility(EMC)—Part 4: Testing and measuring techniques—Section 11: Voltage dips,short interruptions, and voltage variations immunity tests (Adopted CEI/IEC1000-4-11:1994, first edition, 1994-06 including Amendment 1:2000).

CAN/CSA-CEI/IEC 61000-4-12-01—Electromagnetic compatibility(EMC)—Part 4: Testing and measurement techniques—Section 12: Oscillatorywaves immunity test basic EMC publication (Adopted CEI/IEC1000-4-12:1995, first edition, 1995-05, including Amendment 1:2000).

CAN/CSA-CEI/IEC 61000-4-16-02—Electromagnetic compatibility(EMC)—Part 4-16: Testing and measurement techniques—Test for immunity toconducted, common mode disturbances in the frequency range 0 Hz to 150 kHz(Adopted CEI/IEC 61000-4-16:1998, first edition, 1998-01, includingAmendment 1:2001).

CAN/CSA-CEI/IEC 61000-4-17-02—Electromagnetic compatibility(EMC)—Part 4-17: Testing and measurement techniques—Ripple on DC inputpower port immunity test (Adopted CEI/IEC 61000-4-17:1999, first edition,1999-06, including Amendment 1:2001).

CAN/CSA-CEI/IEC 61000-4-27-01—Electromagnetic compatibility(EMC)—Part 4-27: Testing and measurement techniques—Unbalance,immunity test (Adopted CEI/IEC 61000-4-27:2000, first edition, 2000-08).

CAN/CSA-CEI/IEC 61000-4-28-01—Electromagnetic compatibility(EMC)—Part 4-28: Testing and measurement techniques—Variation of powerfrequency, immunity test (Adopted CEI/IEC 61000-4-28:1999, first edition,1999-11).

Appendix B 257

B.4 European Standards

CISPR 14-1, Amendment 2, Edition 4.0—EMC—Requirements for householdappliances, electric tools, and similar apparatus—Part 1: Emission.

CISPR 16-1, Consolidated, Edition 2.1—Specification for radio disturbanceand immunity measuring apparatus and methods—Part 1: Radio disturbanceand immunity measuring apparatus.

CISPR 16-2, Consolidated, Edition 1.2—Specification for radio disturbanceand immunity measuring apparatus and methods—Part 2: Methods ofmeasurement of disturbances and immunity; Amendment 1:1999 to CISPR16-2:1999.

CISPR 16-3, Consolidated, Edition 1.1—Specification for radio disturbanceand immunity measuring apparatus and methods—Part 3: Reports andrecommendations of CISPR.

CISPR 16-4, Edition 1.0—Specification for radio disturbance and immunitymeasuring apparatus and methods—Part 4: Uncertainty in EMC measurements.

CISPR 19:1983—Guidance on the use of substitution method formeasurements of radiation from microwave ovens for frequencies above 1 GHz.

CISPR 20:1999—Sound and television broadcast receivers and associatedequipment immunity characteristics—Limits and methods of measurement.

CISPR 24:1997—Information technology equipment—Immunitycharacteristics—Limits and methods of measurement.

CISPR 61000-6-3:1996—Electromagnetic compatibility (EMC)—Part 6:Generic standards—Section 3: Emission standard for residential, commercialand light-industrial environments.

CISPR/TR 16-3:2000—Specification for radio disturbance and immunitymeasuring apparatus and methods—Part 3: Reports and recommendations ofCISPR.

CISPR/TR 28:1997—Industrial, scientific and medical equipment(ISM)—Guidelines for emission levels within the bands designated by ITU.

EN 300 127:1999—Electromagnetic compatibility and radio spectrum matters(ERM); Radiated emission testing of physically large telecommunicationsystems.

EN 61000-2-9:1996—Electromagnetic compatibility (EMC)—Part 2:Environment—Section 9: Description of HEMP environment—Radiateddisturbance.

EN 61000-4-3:2002—Electromagnetic compatibility (EMC)—Part 4-3: Testingand measurement techniques—Radiated, radio-frequency, electromagnetic fieldimmunity test; Amendment A1:1998 to EN 61000-4-3:1996.

EN 61000-4-6:1996—Electromagnetic compatibility (EMC)—Part 4-6: Testingand measurement techniques—Immunity to conducted disturbances induced byradio-frequency fields.

EN 61000-4-8:1994—Electromagnetic compatibility (EMC)—Part 4-8: Testingand measurement techniques—Power frequency magnetic field immunity test.

258 Appendix B

EN 61000-4-9:1994—Electromagnetic compatibility (EMC)—Part 4-9: Testingand measurement techniques—Pulse magnetic field immunity test.

EN 61000-4-23:2001—Electromagnetic compatibility (EMC)—Part 4-23:Testing and measurement techniques—Test methods for protective devices forHEMP and other radiated disturbances.

EN 61000-4-28:2000—Electromagnetic compatibility (EMC)—Part 4-28:Testing and measurement techniques—Variation of power frequency, immunitytest.

ETS 300 127:1994—Equipment engineering (EE); Radiated emission testing ofphysically large telecommunication systems.

IEC 61000-3-2:2001—Electromagnetic compatibility (EMC)—Part 3: Limits.Section 2: Limits for harmonic current emissions for electronic equipment(equipment input current less than 16A per phase).

B.5 Other Standards

IEEE 139:1988 (R1999)—IEEE recommended practice for the measurement ofradio-frequency emission from industrial, scientific, and medical (ISM)equipment installed on user’s premises.

IEEE 213:1987 (R1998)—IEEE standard procedure for measuring conductedemissions in the range of 300 kHz to 25 MHz from television and FM broadcastreceivers to power lines.

IEEE C63.4:2000—Methods of measurement of radio-noise emissions fromlow-voltage electrical and electronic equipment in the range of 9 kHz to40 GHz.

IEEE C63.5:1998—Electromagnetic compatibility—Radiated emissionmeasurements in electromagnetic interference (EMI) control—Calibration ofantennas.

IEEE C63.7:1992—Construction of open-area test sites for performingradiated emission measurements.

IEEE C63.18:1997—Recommended practice for an on-site ad hoc test methodfor estimating radiated electromagnetic immunity of medical devices to specificradio-frequency transmitters.

Appendix B 259

Acronyms and AbbreviationsAWGN additive white Gaussian noiseAF antenna factorAGA air ground airALSE absorber-lined shielded environmentAM analog modulationANSI American National Standards InstituteASK amplitude shift keyingBCI bulk current injectionBPF band-pass filterBSS broadcasting satellite serviceCE conducted emissionsCENELEC European Committee for Electrotechnical

StandardizationCEPT European Conference of Postal and Telecommunications

AdministrationsCI conducted immunityCISPR International Special Committee on Radio InterferenceCS conducted susceptibilityCSA Canadian Standards AssociationCW continuous waveDCS 1800 Digital Communication SystemDECT Digital Enhanced Cordless Telecommunication SystemDUT device under testDVB-T terrestrial digital video broadcastingECA European Common AllocationECC Electronic Communications CommitteeEEE or E3 electromagnetic environmental effectsEESS Earth Exploration-Satellite ServiceEGSM Extended GSMEMC electromagnetic compatibilityEMD electromagnetic disturbanceEMI electromagnetic interferenceEMP electromagnetic pulse

261

ERC European Radiocommunications CommitteeERO European Radiocommunications OfficeERP effective radiated powerESD electrostatic dischargeEUT equipment under testFCC Federal Communications CommissionFDD frequency division duplexFIR finite response filterFM frequency modulationFSK frequency shift keyingFSS fixed satellite serviceFWA fixed wireless accessGNSS Global Navigation Satellite SystemGSM Global System for Mobile CommunicationsGTEM Gigahertz Transverse ElectromagneticHDTV High Definition TelevisionHEMP High-Altitude Electromagnetic PulseHIPERLAN High Performance Radio Local Area NetworkHPF highpass filterIEC International Electrotechnical CommissionIEEE Institute of Electrical and Electronics EngineersIEMI intentional electromagnetic interferenceIIR infinite response filterILS Instrument Landing SystemISM industrial, scientific, and medical equipmentISO International Organization for StandardizationITU International Telecommunication UnionLISN Line Impedance Stabilization NetworkLPF lowpass filterNATO North Atlantic Treaty OrganizationNEMP nuclear electromagnetic pulseNGSO nongeostationary satellite orbitOATS Open Area Test SitePAMR Public Access Mobile RadioPCM pulse code modulationPLT power line transientPM phase modulationPMR Professional Mobile Radio, Private Mobile RadioPSK phase shift keyingPWM pulse width modulationQAM quadrature amplitude modulation

262 Acronyms and Abbreviations

RA radio astronomyRE radiated emissionsRES radiated electromagnetic susceptibilityRF radio frequencyRFI radio frequency interferenceRFID radio frequency identification systemsRLAN radio local area networkRR radio regulationsRS radiated susceptibilitySC conducted susceptibilityT-DAB Terrestrial Digital Audio BroadcastingTEM transverse electromagneticTETRA Terrestrial Trunked RadioUMTS/IMT-2000 International Mobile TelecommunicationsUTP unshielded twisted pairVSAT very small aperture terminalVSWR voltage standing wave ratioXTALK Crosstalk

Acronyms and Abbreviations 263

GlossaryAccuracy of measurement The closeness of the agreement between theresult of a measurement and the value of the measurand.Corrected result The result of a measurement after correction for assumedsystematic error.Correction A value added algebraically to the uncorrected result of a mea-surement to compensate for systematic error (equal to the negative of estimatedsystematic error).Combined standard uncertainty The standard uncertainty of a measure-ment result when that result is obtained from the values of a number of otherquantities, equal to the positive square root of a sum of terms; the terms beingthe variances or covariances of these other weighted quantities.Coverage factor A numerical factor used as a multiplier of the combinedstandard uncertainty for obtaining an expanded uncertainty (e.g., a coveragefactor, k, is typically in the range of 2 to 3, but may range lower for special pur-poses; when k = 2, the probability level approximates 95%).Error of measurement The result of a measurement minus the value of themeasurand.Expanded uncertainty The quantity defining the interval about the result ofa measurement within which the values that reasonably could be attributed tothe measurand may be expected to be at a high level of confidence.Precision The quality of being exactly or sharply defined or stated; the mea-sure of accuracy of a representation is the number of distinguishable alterna-tives from which it was selected, which is sometimes indicated by the number ofsignificant digits it contains.Random error The result of a measurement minus the mean that wouldresult from an infinite number of measurements of the same measurand carriedout under repeatability conditions.Relative error of measurement The error of measurement divided by a truevalue of the measurand.Repeatability of measurement results The closeness of agreement betweenthe results of successive measurements of the same measurand carried out underthe same measurement conditions.Resolution The least value of the measured quantity that can be distin-guished.

265

Reproducibility of measurement results The closeness of agreementbetween the results of measurements of the same measurand carried out underchanged measurement conditions.Standard uncertainty The uncertainty of the result of a measurementexpressed as standard deviation.Systematic error The mean that would result from an infinite number ofmeasurements of the same measurand carried out under repeatability condi-tions minus the value of the measurand.Type A evaluation of standard uncertainty An evaluation method of stan-dard uncertainty by statistical analysis of a series of observations.Type B evaluation of standard uncertainty An evaluation method of stan-dard uncertainty by means other than statistical analysis of a series of observa-tions.Uncertainty of measurement A parameter, associated with the result of ameasurement, which characterizes the dispersion of the values that could rea-sonably be attributed to the measurand.Uncorrected result The result of a measurement before correction for theassumed systematic error.

266 Glossary

About the AuthorKresimirMalaric received a B.Sc., an M.Sc., and a Ph.D. from the Faculty of Electri-cal Engineering and Computing, University of Zagreb in 1991, 1994, and 2000,respectively. From 1992 to 1995 he worked at the Department of Fundamentals ofElectrical Engineering and Measurements. Since 1996 he has been with the Depart-ment of Wireless Communications and with the Faculty of Electrical Engineeringand Computing at the University of Zagreb, where he is an associate professor. Hiscurrent teaching and research areas include electromagnetic compatibility, satellitecommunications, and biomedical effects. He has published more than 100 confer-ence and journal scientific papers. Dr. Malaric is a member of the IEEE and the Cro-atian Academy of Technical Sciences.

267

IndexAAbsorbers, 185–89, 193, 201Absorption, 51, 53, 137, 141, 200–1Absorption loss, 132, 141–42, 144–45Active filters, 79, 85, 91ADC, 79, 91–93, 118Additive white Gaussian noise (AWGN), 57Advanced Encryption Standard (AES), 5Amplitude modulation, 10, 69, 105–6, 120

quadrature, 10, 105, 112Amplitude shift keying (ASK), 113–14Analog communication system, 1–2Analog filters, 79, 85–91Analog systems, 1, 17, 57Anechoic chambers, 185–88Antenna, 19–22, 192–93Antenna impedance, 21Antenna systems, 19, 21Antistatic wrist tape, 63Aperture dimensions, 138Attenuation, 36, 46

atmosphere, 65–69filter, 80–86

Attenuation constant, 36–37

BBandpass filter, 81–83Bandstop filters, 83Bessel filters, 87Bit errors, 13, 15Bolting, 163–64Bonding, 134, 153–54, 158–60, 162, 164–66Bonding classes, 160Bonding resistance, 160, 162–63BPSK, 117Brazing, 62–63Butterworth filters, 85–86

CCable screens, 133–34Cable shielding, 128, 132–33, 135Cables, tri-axial, 133Capacitance, stray, 128, 132, 158, 162Capacitive coupling, 128–29, 154, 221–22

Capacitive coupling clamp, 219–20, 226Capacitive reactance, 161Cavities, 19–20Characteristic impedance, 41, 98, 158,|

196–201Chebyshev, 86–87, 94, 98–99Clamp, 218–22Commercial radar systems, 74Conducted emission (CE), 170–72, 178–79Conductive adhesive, 164Control of system drift, 120–21Convolutional encoder, 7–8Coupling, 61, 99, 123–32, 134, 136–40, 142,

168–69, 215–22cable-to-cable, 125common mode, 130differential mode, 130inductive, 131, 219, 221–22

Coupling capacitors, 215–16Coupling clamp, 219, 222Coupling Transformer, 217CS (Conducted susceptibility), 170–72,

178–81Cutoff frequency, 79–81, 83, 85–87, 91, 97,

140, 194–95, 197–98, 209–11

DDecoder, 2, 9, 11, 13–15Decryption, 15Decryptor, 15Deinterleaver, 11–12Demodulation, 12, 17, 103–5, 107–13, 115,

117–19Demodulator, 1–2, 11–12, 105, 108, 115, 121Demultiplexer, 11, 16–17Derandomizer, 11, 15Detection, 5, 58, 103–5, 108

envelope, 12, 107–8, 112, 115, 118Dielectric resonators, 100–1Digital filters, 91–97, 101Digital modulation, 10, 105, 112Directivity, 20–22Dissimilar metal, 164Duplexer, 2, 19

269

EEffective radiated power (ERP), 72–73, 262Electric field, 40, 47, 100, 128, 130–32,

145–48, 172, 179, 181–82, 197–98,205

Electric field coupling to wires, 128–29Electrical grounding and bonding, 153–54,

156, 158, 160, 162, 164, 166Electromagnetic interference, 57–77, 126, 134,

151, 155, 167, 170, 215Electromagnetic spectrum, 27–29, 167Electrostatic discharge (ESD), 57, 59, 62–64,

154, 156, 169–70, 219, 223–24, 262EM clamp, 221EMI receivers, 170, 224Emission, conducted, 170–72, 178, 224–25,

261Emission tests, 182, 221–22Encoder, 2, 5, 7, 12–13

convolutional, 6–8differential, 6–7

Encoding, convolutional, 7–9Encryption, 2, 5, 15Equipment shielding, 147, 149, 151Equipotential plane, 156, 158–59Evaluation of standard uncertainty, 227–28Extra low frequency (ELF), 27–29

FFaraday cage, 133, 145–46, 148, 185Fault protection, 155–59Ferrite beads, 149Ferrite tiles, 185, 187, 189–91Field-to-aperture coupling, 137, 139Filter order, 86Filter types, 88, 97–98Filtering, 11–12, 103, 108, 126, 128, 136Filter order, 86–93Filters, 11–12, 79–81, 83–91, 93–94, 98–99,

102–4, 148–49, 155, 169–70active, 79, 91–92elliptic, 88ferrite, 126–7nonrecursive, 92, 95–96passive, 79, 85, 88, 91, 224recursive, 92, 95–6

Finite impulse response (FIR), 92, 94–95, 262FIR filters, 93Flicker noise, 58Floating ground, 156–57

Frequency modulation (FM), 10, 29, 69, 71,105, 109–12

signal, 109–10Frequency shift keying (FSK), 10, 105, 112,

114–15modulation, 114–15, 117signal, 115–16, 118

Fresnel knife-edge diffraction, 48Full anechoic and semianechoic chambers,

185, 187, 189

GGasketing, 147Golay decoder 15Graphical user interface (GUI), 18Ground

conductors, 155, 157fault, 153, 155–56plane, 148–49, 160, 192protection, 154–55reference plane, 159, 221–22wire, 155

Grounding, 128, 132–33, 153–56, 158, 166GTEM cell, 201, 203, 205, 207, 209, 211, 213

HHigh frequencies, 28–30, 123, 130, 133, 147,

160–62, 225Higher-order modes, 197–99, 207Highpass filters, 11, 80–81, 86

IIIR filters, 94–96Immunity, 76, 137, 177, 218Impedance

antenna, 21ferrite, 126

Industrial sources, 175Interference, 57, 69, 167

intentional, 76radiated, 167

Interleaving, 9Intermodulation products, 10, 12Injection clamps and probes, 221Ionosphere, 53–54, 69Isotropic radiator, 21

KKey

encryption, 5private, 5

270 Index

public, 5Kuroda transformation, 98

LLightning, 59–62, 153–62

protection, 154, 160strike, 156

Line impedance stabilization network (LISN),215–16

Line of sight, 50Load impedance, 39, 128Look-up tables (LUTs), 9Lowpass filter, 80, 99

RC filter, 89RL filter, 89

Lumped element filters, 97

MMagnetic field, 123, 146Maxwell equations, 44–45Measurement facilities, 185Measurement uncertainty, 227Microwave filters, 97–101Military requirements, 178Minimum shift keying (MSK), 114Mixer, 10, 12Modulated signal, 10, 105Modulating signal, 105Modulation index, 107Modulator, 10, 105Moisture, 63, 148, 154, 165, 193Multipath, 64–65Multiplexer, 2–4Multipoint ground, 158

NNoise

burst, 59flicker noise, 58shot, 58source, 131–33spectral density, 59thermal, 58voltages, 156white, 103

Noise factor (NF), 16–19

OOpen area test site (OATS), 191Oscilloscope, 225

PPassive filters, 88, 224Parallel plate, 217Path, 51

direct, 64–65loss, 51–53reflected, 64–65

Permeability, 35, 123, 150, 189Permittivity, 35, 100, 189Phase constant, 36Phase difference, 47, 65, 119Phase shift keying (PSK), 116Polarization, 42–43, 47–49, 67

circular, 47–48elliptical, 48linear, 47

Potential difference, 27, 60, 164Power

amplifiers, 23cables, 167density, 16, 38, 46, 51supply, 23

Private networks, 74Probability

distributions, 228functions, 104

Probes, 195, 218current 220

Propagation, 44, 54, 57Protection

fault, 155PCB, 148

Pulse code modulation (PCM), 118Pyramidal absorbers, 187–89

QQPSK, 117–18, 120Quadrature amplitude modulation (QAM),

119

RRadar, 74–75Radiated

EMI, 169–72emissions, 181power, effective, 72susceptibility, 182

Radiation pattern, 20Rain, 47, 54, 65–68

rate, 66–68

Index 271

Randomization, 4Reactance, 41, 97, 159Received power, 16Receiver, 11, 24

sensitivity, 17systems, 11

Rectangular distribution, 229Reed-Solomon

coding, 8decoding, 14

Reflected wave, 37–38, 40Reflection, 42, 141

coefficient, 37–38, 187, 193loss, 143–44total, 42

Refraction, 42Resistance, 123Resistivity, 123–24Resonances, 84–88, 147, 162, 197

parallel, 85serial, 84

Resonators, 98, 100–1Reverberation chambers, 193RF

mismatch error, 232susceptibility, 186

Richard transformation, 97Ripple, 85–88Room, shielded, 172

SS/UTP, 134Safety, 154–55Salisbury paper, 187Scattering, 53–54, 188Selectivity, 11–12, 83, 224Semianechoic chambers, 185Sensitivity, 17, 225, 230–31Shield, 132, 146, 149

apertures, 137magnetic, 149

Shielded twisted pair (STP), 134Shielding effectiveness (SE), 138Shock, 154, 215

hazard, 154–55, 160Shot noise, 58Single point ground, 157–58Skin depth, 37, 123, 132, 141, 218Skin effect, 123, 194Smith chart, 39–41Snell’s law, 42, 64Sources

industrial, 75manmade, 69natural, 59

Space wave, 20Spectrum analyzer, 225Spring fingers, 148Standard deviation, 227–28Standard uncertainty, 227–31Static electricity, 62–63, 153Stopbands, 88Strap, 160–62, 165Striplines, 149Subkey, 6Sunspot activity, 68–69Surface wave, 54, 64Susceptibility, 75, 167, 171

conducted, 172, 179radiated, 170, 172

SWR, 40Systematic errors, 227

TThermal noise, 17, 57Total reflection, 42Total transmission, 42–43Transfer function (filter), 79Transformers

λ/4, 41coupling, 217

Transients, 59, 170–71Transmission line, 38, 41, 44, 97, 149–50,

157, 162, 195, 201, 232Transmitter, 3Transverse electromagnetic mode (TEM)

cell, 195mode, 197

Tri-axial cable, 133Troposphere, 53–54Twisted pair

screened shielded, 134–35screened unshielded, 134–35shielded, 134unshielded, 125

UUninterruptible power supply (UPS), 23Unshielded twisted pair (UTP), 125User interface, 18

graphic, 18voice, 19

U-shaped distribution, 230

272 Index

VVictim, 167–69Viterbi decoder, 14Voice user interface, 19Voltage reference control, 156Voltage standing wave ratio (VSWR), 200,

204–5, 211

WWave

generation and propagation, 44impedance, 35–37, 195planar, 195, 201transmitted, 42, 190

Waveguide cavity filter, 98–99Welding, 162–63Wire coupling, 123, 130

Index 273

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