Wideband Microwave OTA with Tunable Transconductance using Feedforward Regulation and ...post. saavedra/research/papers/conference2016-06-02Wideband Microwave OTA with Tunable Transconductance

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  • Wideband Microwave OTA with TunableTransconductance using Feedforward Regulation

    and an Active Inductor LoadJiangtao Xu1,2, Carlos E. Saavedra1 and Guican Chen2

    1 Dept. of Electrical and Computer Engineering, Queens University, Kingston, Ontario, Canada2School of Electronic and Information, Xian Jiaotong University, Xian, China

    e-mail: saavedra@queensu.ca

    AbstractA microwave operational transconductance ampli-fier (OTA) using a feedforward-regulated cascode stage and anactive inductor load is proposed in this paper. The use of thefeedforward regulation mechanism provides the OTA with a verywide bandwidth of over 10 GHz while the active inductor loadcontributes to the OTAs high linearity performance. Further-more, the transconductance of the OTA can be varied from 5 mSto 20 mS. The OTA is designed in a 0.13-m CMOS technologyand it draws a maximum of 5.5 mA of current from a 1.2 Vsupply when the transconductance is set to 20 mS. The chipoccupies an area of 110 m 66 m.


    The operational transconductance amplifier (OTA) is animportant building block in anolog circuit design and it hasbeen extensively used in low frequency applications [1][6].However, with the rapid development of wireless communica-tions, microwave OTAs that can operate in the gigahertz rangeare of high interest.

    High-frequency OTAs have recently been used in RF fil-ters [7], oscillators [8] and phase shifters [9], for example.While CMOS technology continues to move toward smallerfeature sizes and power supply voltages are reduced, this cancompromise the linearity performance of circuits using thosedeep-submicron devices. Yet, circuits used in RF applicationsstill have to meet their specifications, some of which demandhigh linearity performance.

    One approach to addressing linearity requirements in RFand microwave circuits is to process signals in the currentdomain. Since the inner nodes in current-mode circuits usuallyhave low impedance, the voltage gain is small and the polesassociated with these nodes are located at high frequency,which result in high linearity and high operating frequency,respectively.

    In this paper, a high linearity, wideband OTA with tunabletransconductance is presented. The circuit uses the principleof feedforward regulation [10] which leads to its widebandperformance. An active inductor load is used to enhance thelinearity of the OTA. The operating principles and circuitimplementation of the OTA are discussed in Section II. Theperformance results are presented in Section III and SectionIV concludes the paper.

    Fig. 1. The proposed OTA


    The schematic of the proposed OTA is shown in Fig. 1and it is a variant of the amplifier designs presented in [10][11]. The OTA in Fig.(1) is a low-power, high-speed circuitthat uses an active inductor load. The feedforward-regulatedcascode topology makes it more appropriate for the low supplyvoltage while retaining the characteristics of high linearityand high speed performance [1][6]. A folded active inductortopology is employed here to improve the high frequencyperformance [12]. The input transconductor stage is formed bythe transistors M1 and M2 operating in the triode region. M3and M4 are utilized as the regulated cascode stage to enhancethe linearity and to boost the output impedance. The gatesof transistors M3 and M4 are biased with the same voltage,Vc. The transconductance, Gm, of the OTA can be changed byvarying the value of Vc. The folded active inductor, consistingof M5, M6, R5 and R6, is used as the active load to allow theOTA to work at high frequencies.

    A. Input Transconductor

    Using a short channel model to characterize the draincurrent to the gate-to-source voltage of a MOS transistor in

    978-1-4244-6805-8/10/$26.00 2010 IEEE


  • triode region, its transconductance can be expressed as follows:

    Gm = nCoxW



    1 + (VDS/EsatL)(1)

    where W and L is the width and length of the transistor,respectively; is the channel mobility; Cox is the oxide ca-pacitance; Esat is the saturated electrical field. As we can seein Eq:(1), the transconductance of the triode transistor is onlyrelated to VDS assuming all the physics-related parameters areconstant. So if we can keep the VDS constant as the amplitudeof the input small signal varies, then the high linearity willbe obtained. However, the channel mobility is actually notconstant. It is related to gate voltage due to mobility reductioneffect. The mobility compensation technique to solve thisproblem has been specified in detail in [4], [5].

    B. Feedforward-Regulated Cascode StageThe feedforward regulation is implemented by cross cou-

    pling between the differential cascode pairs M1/M3 andM2/M4. When the differential input signal increases, thedrain voltage of M1 decreases while the drain voltage of M2increases. The signal at the drain of M2 is then cross coupledto the gate of the cascode transistor M3. As VGS3 increases,M3 inversely injects more current into M1, which then elevatesVA and counteracts any change of VA. Therefore, by creatingan inter-locking regulation mechanism between VA and VB ,we get a stable DC voltage at the drains of M1 and M2.Besides, The cascoding transistor M3 also increases the totaloutput impedance, which is beneficial for the OTA.

    In [13], this function is performed by a feedback amplifierwith gain of -A directly from node A to the gate of M3.Higher gain means higher accuracy, but feedback and high-gain amplifier both result in low operating frequency. In thispaper, feedforward instead of feedback is employed by usingjust two capacitors to cross-couple the signal from nodes Band A. The variation at node B will be directly fed forwardto the gate of M1 and then compensate the variation at nodeA instantly, which is simpler and faster. At the same time,the RF signals at nodes A and B are further amplified by thecapacitative cross coupled cascode transistors M3 and M4.They perform small signal amplification with the same signas the RF input signals, and together with M1 and M2 theyare added to increase the total transconductance.

    C. Active Inductor LoadThe active load consisting of M5 and R5 is actually the

    PMOS version of the folded active inductor proposed in [12].To derive the parameters of the RLC equivalent circuit of thefolded active inductors, its small-signal equivalent circuit isshown in Fig. 2. To simplify analysis, we neglect Cgd, goalong with other parasitic capacitances of the transistor. It canbe shown that the input impedance is given by :

    Zin =sRCgs + 1

    sCgs + gm(2)

    It becomes evident that Zin has a zero at the frequency z =1

    RCgsand a pole at p = gmCgs . The active load is resistive at

    Fig. 2. The Active Inductor Load

    low frequencies < z with resistance R 1gm and inductivewhen z < < p. To derive the RLC equivalent circuit forthe network in Fig. 2b, we examine the input admittance ofthe circuit,

    Yin =1



    sRCgsgm 1R

    + 1gm 1R


    Eqn. (3) can be represented by a series RL network inparallel with a resistor Rp as shown in Fig. 2c, and thecomponent values are given by,

    Rp = R

    L =RCgsgm 1R

    Rs =1

    gm 1R


    It becomes evident that gm > 1R is required in order to haveL > 0 and Rs > 0.

    From a different perspective, the resistive self-biasing activeload also has the function of common mode feedback. Theoperating mechanism is as follows. Assuming a reduction inthe DC voltage appears at the output node, this will result inan equal increase in |VGS5|. The PMOS device M5 will injectmore DC current into the NMOS devices M3 and M1. Toaccommodate the variation of the current, M1 and M3 haveto increase their drain voltage to compensate the reductionof DC voltage at the output node. This is another benefit ofthe resistive self-biasing active load, which removes the high-gain amplifier used in common-mode feedback circuits, forinstance.

    III. SIMULATION RESULTSThe small-signal transconductance of the OTA as a function

    of frequency and control voltage, Vc, is shown in Fig. 3. Theproposed OTA works well up to and beyond 10 GHz. Tofurther investigate the behavior of the OTAs transconductanceversus Vc, the process corner simulation including three sigmavariations was carried out at a fixed frequency of 5.4 GHz andthe results are shown in Fig. 4. TT,FF,FS,SF and SS repre-sent TypicalNMOS/TypicalPMOS, FastN/FastP, FastN/SlowP,


  • 0 2 4 6 8 10

    x 109







    Frequency (Hz)








    Vc=0.5 V

    Vc=0.55 V

    Vc=0.6 V

    Vc=0.65 V

    Vc=0.7 V

    Vc=0.75 V

    Vc=0.8 V

    Fig. 3. Simulated OTA transconductance versus frequency and controlvoltage Vc

    0.45 0.5 0.55 0.6 0.65 0.7 0.75 0.80.01











    Vc (V)













    Fig. 4. Simulated OTA transconductance versus the control voltage Vc at5.4 GHz

    SlowN/FastP and SlowN/SlowP, respectively. We observe thatthe transconductance has a very linear response versus biascontrol voltage and it can be tuned from 5 mS to 20 mS asVc varies from 0.45 V to 0.8 V for TT corner situation.

    As a transconductance amplifier that translates the inputvoltage signal into output current , the OTA is usually loadedwith low impedance such as the switches of the passive mixeror transimpedance amplifier (TIA) stage. In the followinglinearity simulation, the OTA is loaded with 50 resistor,which will lower t


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