PFC boost converter design guide

Application Note 1 Revision1.1, 2016-02-22

About this document

Scope and purpose

This document introduces a design methodology for a Power Factor Correction (PFC) Continuous

Conduction Mode (CCM) boost converter, including:

Equations for design and power losses

Selection guide of semiconductor devices and passive components

Charts for CoolMOS™ optimum RDS(ON) selection

1200 W design example with calculated and experimental results.

Schematics, layout and bill of material

Intended audience

This document is intended for design engineers who want to design CCM PFC boost converter.

Table of contents

Introduction ................................................................................................................................... 2 1

Power stage design ........................................................................................................................ 4 2

ICE3PCS01G PFC boost controller ................................................................................................ 17 3

Efficiency and power losses modeling ......................................................................................... 19 4

Experimental results .................................................................................................................... 20 5

Board design ................................................................................................................................ 23 6

References ................................................................................................................................... 27 7

Symbols used in formulas ............................................................................................................ 28 8

PF C bo os t con verter desi gn gu i de 1200 W design example

Sam Abdel-Rahman

Franz Stückler

Ken Siu

Application Note

PFC boost converter design guide


Design Note DN 2013-01 V1.0 January 2013

Introduction 1Power Factor Correction (PFC) shapes the input current of the power supply to be in synchronization with

the mains voltage, in order to maximize the real power drawn from the mains. In a perfect PFC circuit, the

input current follows the input voltage as a pure resistor, without any input current harmonics. This document is to introduce a design methodology for the CCM PFC Boost converter, including equations for power losses estimation, selection guide of semiconductor devices and passive components, and a design example with experimental results.

1.1 Boost topology

Although active PFC can be achieved by several topologies, the boost converter (Figure 1) is the most

popular topology used in PFC applications, for the following reasons:

The line voltage varies from zero to some peak value typically 375 V; hence a step up converter is needed to output a DC bus voltage of 380 V or more. For that reason the buck converter is eliminated, and the

buck-boost converter has high switch voltage stress (Vin+Vo), therefore it is also not the popular one.

The boost converter has the filter inductor on the input side, which provides a smooth continuous input current waveform as opposed to the discontinuous input current of the buck or buck-boost topology.

The continuous input current is much easier to filter, which is a major advantage of this design because any additional filtering needed on the converter input will increase the cost and reduces the power

factor due to capacitive loading of the line.

Boost Key Waveforms

𝑉𝑜

𝑉in=

1

1 − D (CCM operation)

Figure 1 Structure and key waveforms of a boost converter

1.2 PFC modes of operation

The boost converter can operate in three modes: continuous conduction mode (CCM), discontinuous conduction mode (DCM), and critical conduction mode (CrCM). Figure 2 shows modeled waveforms to

illustrate the inductor and input currents in the three operating modes, for the same exact voltage and power conditions.

By comparing DCM among the others, DCM operation seems simpler than CrCM, since it may operate in

constant frequency operation; however DCM has the disadvantage that it has the highest peak current compared to CrCM and also to CCM, without any performance advantage compared to CrCM. For that

reason, CrCM is a more common practice design than DCM, therefore, this document will exclude the DCM

design.

AC

PFC

Converter

DC/DC

ConverterLoad

DC Bus

Vac

AC

L D

RoVac CoS

+

Vo

-

DC Bus

S

V_L

I_L

I_S

I_D

T=1/f

DT

Vin

Vin-VoI_Lmax

I_Lmin

I_Lmax

I_Lmax

I_Lmin




CrCM may be considered a special case of CCM, where the operation is controlled to stay at the boundary between CCM and DCM. CrCM usually uses constant on-time control; the line voltage is changing across the

60 Hz line cycle, the reset time for the boost inductor is varying, and the operating frequency will change as well in order to maintain the boundary mode operation. CrCM dictates the controller to sense the inductor

current zero crossing in order to trigger the start of the next switching cycle.

The inductor current ripple (or the peak current) in CrCM is twice of the average value, which greatly increases the MOSFET RMS currents and turn-off current. But since every switching cycle starts at zero

current, and usually with ZVS operation, turn-on loss of MOSFET is usually eliminated. Also, since the boost rectifier diode turns off at zero current as well, reverse recovery losses and noise in the boost diode are

eliminated too, another major advantage of CrCM mode. Still, on the balance, the high input ripple current and its impact on the input EMI filter tends to eliminate CrCM mode for high power designs unless

interleaved stages are used to reduce the input HF current ripple. A high efficiency design can be realized

that way, but at substantially higher cost. That discussion is beyond the scope of this application note.

The power stage equations and transfer functions for CrCM are the same as CCM. The main differences relate to the current ripple profile and switching frequency, which affects RMS current and switching power losses and filter design.

CCM operation requires a larger filter inductor compared to CrCM. While the main design concerns for a

CrCM inductor are low HF core loss, low HF winding loss, and the stable value over the operating range (the inductor is essentially part of the timing circuit), the CCM mode inductor takes a different approach. For the

CCM PFC, the full load inductor current ripple is typically designed to be 20-40% of the average input

current. This has several advantages:

Peak current is lower, and the RMS current factor with a trapezoidal waveform is reduced compared to a triangular waveform, reducing device conduction losses.

Turn-off losses are lower due to switch off at much lower maximum current.

The HF ripple current to be smoothed by the EMI filter is much lower in amplitude.

On the other side, CCM encounters the turn-on losses in the MOSFET, which can be exacerbated by the boost rectifier reverse recovery loss due to reverse recovery charge, Qrr. For this reason, ultra-fast recovery diodes

or silicon carbide Schottky Diodes with extreme low Qrr are needed for CCM mode.

In conclusion, we can say that for low power applications, the CrCM boost has the advantages in power

saving and improving power density. This advantage may extend to medium power ranges, however at some medium power level the low filtering ability and the high peak current starts to become severe

disadvantages. At this point the CCM boost starts being a better choice for high power applications.

Since this document is intended to support high power PFC applications, therefore a CCM PFC boost converter has been chosen in the application note with detailed design discussions and design examples for

demonstration.

0 1 104-

2 104-

3 104-

4 104-

0

5

10

15

Iin t( )

IL t( )

tContinuous Conduction Mode (CCM)

0 1 104-

2 104-

3 104-

4 104-

0

5

10

15

Iin t( )

IL t( )

tDiscontinuous Conduction Mode (DCM)

Figure 2 PFC Inductor and input line current waveforms in the three different operating modes

0 1 104-

2 104-

3 104-

4 104-

0

5

10

15

Iin t( )

IL t( )

tCritical Conduction Mode (CrCM)




Power stage design 2The following are the converter design and power losses equations for the CCM operated boost. The design example specifications listed in Table 1 will be used for all of the equations calculations. Also the boost converter encounters the maximum current stress and power losses at the minimum line voltage condition (𝑉𝑎𝑐.𝑚𝑖𝑛 ); hence, all design equations and power losses will be calculated using the low-line voltage

condition as an extreme case.

Specifications of the power stage Table 1

Input voltage 85-265 VAC 60 Hz

Output voltage 400 V

Maximum power steady state 1200 W

Switching frequency 100 kHz

Inductor current ripple 25% @ low line/full load

Output voltage 120Hz ripple 10 Vp-p

Hold-up time 16.6 ms @ VO,min=340 V

Figure 3 Block schematic for boost power stage with input rectifier

2.1 Main PFC inductor

Off-the-shelf inductors are available and usable for a first pass design, typically with single layer windings and a permeability drop of 30% or less.

In some circumstances it may be desirable to further optimize the inductor configuration in order to meet the requirements for high power factor over a wide input line current range. Many popular PFC controllers

use single cycle current loop control, which can provide good performance provided that the inductor remains in CCM operation. At low-line this is no problem, but for the operation in the high-line band (176 VAC to 265 VAC), the operating current will be much lower. If an inductor is used with a nominal “stable” value of

inductance, it works well at low-line but results in DCM mode operation for a significant part of the load range at high-line, with poorer power factor, THDi and higher EMI. A swinging choke (also called powder core), such as Arnold/Micrometals Sendust or Magnetics Inc Kool Mu, can address this if designed with the




right energy capability and with full load permeability drop by 75-80%, so that at lighter load the inductance swings up.

The filter inductor value and its maximum current are determined based on the specified maximum inductor current ripple as shown below:

𝐿 =1

%𝑅𝑖𝑝𝑝𝑙𝑒 ∙

𝑉𝑎𝑐.𝑚𝑖𝑛 2

𝑃𝑜

(1 −√2 ∙ 𝑉𝑎𝑐 .𝑚𝑖𝑛

𝑉𝑜

) ∙ 𝑇 =1

0.25 ∙

(85 𝑉)2

1200 𝑊(1 −

√2 ∙ 85 𝑉

400 𝑉) ∙

1

100 ∙ 103𝐻𝑧 = 168.5 𝜇𝐻

Eq. 1

𝐼𝐿.𝑚𝑎𝑥 =√2 ∙ 𝑃𝑜

𝑉𝑎𝑐.𝑚𝑖𝑛

∙ (1 +%𝑅𝑖𝑝𝑝𝑙𝑒

2) =

√2 ∙ 1200 𝑊

85 𝑉∙ (1 +

0.25

2) = 22.5 𝐴 Eq. 2

Note: Inductor saturation current must be rated at > 22.5 A.

In this evaluation board design, a 60 μ permeability Kool Mu core from Magnetics Inc. is used. It consists of two stacked of “Kool Mμ” 77083A7 toroids cores from Magnetics Inc., with 64 turns of 1.15 mm copper wire,

the DC resistance is about 70 mΩ, and the inductance ranges from 680 µH at no load dropping to about 165

µH at low-line full load, which is very close to the desired value calculated above.

Figure 4 Main PFC inductor

Since the inductance value is varying across the line and load range, and also across the line cycle, it would

be more accurate to model the inductance value as a function of Vin, Po and t. in order to obtain better estimation of the switching currents and losses. This specific inductor value was modeled as shown in Figure

5.

Figure 5 Swinging inductance accros load/line range

L (

Hen

ry

)

Po (W)

230Vac @

85Vac @

85Vac @

230Vac @




Inductor copper loss:

The inductor RMS current and the corresponding copper loss are:

𝐼𝐿.𝑟𝑚𝑠 ≅ 𝐼𝑖𝑛.𝑟𝑚𝑠 =𝑃𝑜

𝑉𝑎𝑐.𝑚𝑖𝑛 =

1200 𝑊

85 𝑉= 14.12 𝐴 Eq. 3

𝑃𝐿.𝑐𝑜𝑛𝑑 = 𝐼𝐿.𝑟𝑚𝑠2 ∙ 𝐷𝐶𝑅 = (14.12 𝐴)2 ∙ 0.07 Ω = 13.95 𝑊 Eq. 4

Inductor core losses:

At low-line, core loss across the line cycle is found to be close to a sinusoidal shape (Figure 6, left). Therefore a simple and accurate enough method to estimate the average core loss is to calculate the peak core loss at

the peak of the line cycle point, then multiply by 2/π. However, at high line, core losses are far from the sinusoidal shape (Figure 6, right), so the aforementioned method is not valid anymore, so it is necessary to

model the core loss across the line cycle as a function of time, and then integrate it to obtain the average loss.

Figure 6 Inductor core loss across the line cycle: low-line (left) , high-line (right)

Since in this document we are calculating losses at the minimum line voltage, as the worst case scenario,

thus we will use the first method discussed above to obtain the average core loss in the low-line, as detailed below:

In order to calculate the core loss, we must calculate the minimum and maximum inductor current and the associated minimum and maximum magnetic force (H), then we can use the fitted equation of that

magnetic material to calculate the minimum and maximum magnetic flux (B). Then the AC flux swing can be used to calculate the core loss by using another fitted equation.

For 2 stacked Kool Mμ 77083A7 toroids, we get:

𝑃𝑎𝑡ℎ 𝑙𝑒𝑛𝑔𝑡ℎ 𝑙𝑒 = 98.4 𝑚𝑚 𝐶𝑟𝑜𝑠𝑠 𝑠𝑒𝑐𝑡𝑖𝑜𝑛 𝑎𝑟𝑒𝑎 𝐴𝑒 = 2 ∙ 107 𝑚𝑚2 𝑉𝑜𝑙𝑢𝑚𝑒 𝑉𝑒 = 2 ∙ 10600 𝑚𝑚3

Using the maximum inductor current calculated in Eq. 2, the magnetic force at the peak of the line cycle can be found as:

𝐻𝑚𝑎𝑥 =0.4 ∙ 𝜋 ∙ 𝑁 ∙ 𝐼𝐿.𝑚𝑎𝑥

𝑙𝑒 (𝑖𝑛 𝑐𝑚)=

0.4 ∙ 𝜋 ∙ 64 𝑡𝑢𝑟𝑛𝑠 ∙ 22.51𝐴

98.4 𝑚𝑚/10= 184 𝑂𝑒𝑟𝑠𝑡𝑒𝑑𝑠

Eq. 5

The minimum inductor current and magnetic force at the peak of the line cycle are:

𝐼𝐿.𝑚𝑖𝑛 =𝑃𝑜 ∙ √2

𝑉𝑎𝑐.𝑚𝑖𝑛 (1 −

%𝑅𝑖𝑝𝑝𝑙𝑒

2) =

1200 𝑊 ∙ √2

85 𝑉(1 −

25%

2) = 17.42 𝐴

Eq. 6

𝐻𝑚𝑖𝑛 =0.4 ∙ 𝜋 ∙ 𝑁 ∙ 𝐼𝐿.𝑚𝑖𝑛

𝑙𝑒 (𝑖𝑛 𝑐𝑚)=

0.4 ∙ 𝜋 ∙ 64𝑡𝑢𝑟𝑛𝑠 ∙ 17.42 𝐴

98.4𝑚𝑚/10= 142.37 𝑂𝑒𝑟𝑠𝑡𝑒𝑑𝑠

Eq. 7




Flux density for 60 Koolu material is:

𝐵 = (𝑎 + 𝑏 ∙ 𝐻 + 𝑐 ∙ 𝐻2

𝑎 + 𝑑 ∙ 𝐻 + 𝑒 ∙ 𝐻2)

𝑥

Eq. 8

where a = 1.658e-2 b = 1.831e-3 c = 4.621e-3 d = 4.7e-3 e = 3.833e-5 x = 0.5

The minimum and maximum flux densities at the peak of the line cycle are:

𝐵𝑚𝑎𝑥 = (𝑎 + 𝑏 ∙ 𝐻𝑚𝑎𝑥 + 𝑐 ∙ 𝐻𝑚𝑎𝑥

2

𝑎 + 𝑑 ∙ 𝐻𝑚𝑎𝑥 + 𝑒 ∙ 𝐻𝑚𝑎𝑥2)

𝑥

= 8.483 𝑘𝐺𝑎𝑢𝑠𝑠 Eq. 9

𝐵𝑚𝑖𝑛 = (𝑎 + 𝑏 ∙ 𝐻𝑚𝑖𝑛 + 𝑐 ∙ 𝐻𝑚𝑖𝑛

2

𝑎 + 𝑑 ∙ 𝐻𝑚𝑖𝑛 + 𝑒 ∙ 𝐻𝑚𝑖𝑛2)

𝑥

= 8.014 𝑘𝐺𝑎𝑢𝑠𝑠 Eq. 10

The AC flux swing at the peak of the line cycle is:

∆𝐵 =𝐵𝑚𝑎𝑥 − 𝐵𝑚𝑖𝑛

2= 0.234 𝑘𝐺𝑎𝑢𝑠

Eq. 11

Peak core loss at the peak of the line cycle is:

𝑃𝑐𝑜𝑟𝑒.𝑝𝑘 = ∆𝐵2 ∙ (𝑓

103)

1.46

∙ 𝑉𝑒 ∙ 10−6 = 0.2342 ∙ (100 ∙ 103

103 )

1.46

∙ 2 ∙ 10600 ∙ 10−6 = 0.97 𝑊 Eq. 12

Average core loss across the line cycle is:

𝑃𝑐𝑜𝑟𝑒.𝑎𝑣 = 𝑃𝑐𝑜𝑟𝑒.𝑝𝑘 ∙2

𝜋= 0.968 𝑊 ∙

2

𝜋= 0.62 𝑊

Eq. 13

2.2 Rectifier bridge

Using a higher rated current bridge can reduce the forward voltage drop Vf, which reduces the total power

dissipation at a small incremental cost. Also using two parallel bridges is another approach to distribute the thermal dissipation. This is often a sound strategy, as with modern components, the bridge rectifier usually has the highest semiconductor loss for the PFC stage. In this evaluation board design, 2 parallel GSIB2580 are used.

The bridge total power loss is calculated using the average input current flowing through two of the bridge

rectifying diodes and is shown as:

𝐼𝑎𝑣𝑒𝑟𝑎𝑔𝑒 =2

𝜋∙

√2 ∙ 𝑃𝑜

𝑉𝑎𝑐.𝑚𝑖𝑛 =

2

𝜋∙

√2 ∙ 1200 𝑊

85 𝑉= 12.71 𝐴

Eq. 14

𝑃𝑏𝑟𝑖𝑑𝑔𝑒 = 2 ∙ 𝐼𝑎𝑣𝑒𝑟𝑎𝑔𝑒 ∙ 𝑉𝑓.𝑏𝑟𝑖𝑑𝑔𝑒 = 2 ∙ 12.71 𝐴 ∙ 1 𝑉 = 25.4 𝑊 Eq. 15

The selection on the heat sink is based on the process discussed in chapter 2.6 Heat sink design.

2.3 MOSFET

In order to select the optimum MOSFET, one must understand the MOSFET requirements in a CCM boost

converter. High voltage MOSFETS have several families based on different technologies. For a boost converter, the following are some major MOSFET selection considerations for high efficiency application

design:

Low figure-of merits - RDS(ON)*Qg and RDS(ON)*Eoss




Fast turn-on/off switching to reduce the device switching losses

Gate plateau near middle of gate drive range to balance turn-on/off losses

Low output capacitance Coss for low switching energy and to increase light load efficiency

Drain-source breakdown voltage VBR(DSS) to handle spikes/overshoots

Low thermal resistance RthJC. Package selection must consider the resulting total thermal resistance from junction to ambient, and the worst case surge dissipation, typically under low-line cycle skipping and recovery into highline while ramping the bulk voltage back up.

The body diode commutation speed and reverse recovery charge are not important, since body diode never conducts in the CCM boost converter.

Several CoolMOS™ series can be used for boost applications. C7 followed by CP provides the fastest switching (Figure 7) and best performance, but require careful design in terms of gate driving circuit and

PCB layout. The P6/C6/E6 series provides a cost advantage, with easier design. The P6 series approaches CP performance closely at a better price point, and is recommended for new designs that are cost sensitive. In this Evaluation board C7 is used to reach the best efficiency.

Figure 7 shows that CoolMOS™ C7 total gate charge Qg is less than one third of the C6 charge, and less than

two thirds of the CP charge. Moreover, it shows gate-drain charge Qgd reduction in the much shorter length

of the Miller Plateau compared to previous generations. These improvements in the gate charge profile are

an indication of the improvement of the gate driving related losses as well as of the MOSFET switching times and losses.

Figure 7 Gate charge and Eoss comparision for 41-45 mΩ CoolMOS™ C6, CP, C7

2.3.1 CoolMOS™ optimum RDS(ON) selection charts

Selection of the optimum on-state resistance of a specific CoolMOS™ series is based on the balancing

between switching losses and conduction losses of the device at a targeted load point. This can be done by

modeling all losses in software tool such as Mathcad® by evaluating different technologies and different values of the on-state resistance. This requires few iterations and entry of several parameters from the datasheet of each part.

An alternative way is using the CoolMOS™ selection charts shown in Figure 9 to Figure 12, specific charts

exist for each CoolMOS™ family, optimized at half load or full load. The following is a guide on how to use these CoolMOS™ RDS(ON) selection charts.

Let’s use the specifications in Table 1 as our design example:

Po= 1200 W, f=100 kHz; full load efficiency is critical; CoolMOS™ C7 is preferred for best performance.




Step 1: Find the correct chart, Figure 8 shows the chart for CoolMOS™ C7 optimized at full load.

Step 2: Find the 100 kHz curves (blue curves in this example)

Step3: Mark the 1200 W on the x-axis

Step 3: Find the 1200 W intersection with the solid blue line for the 100 kHz, read the left side y-axis, we

find that 0.055 Ohm is the optimum for CoolMOS™ C7, then we may choose 045C7.

Step 4: Find the 1200 W intersection with the dashed blue line for the 100 kHz, read the right side y-axis, we find that the 0.055 Ω will result in 13.9 W power loss at full load and low line 115 VAC.

Figure 8 Example on how to use the CoolMOS™ optimum RDS(on) selection charts.

45kHz65kHz

0.01

1

0.1

1

100

10

100 10,000

Output Power (W)

Op

tim

um

FE

T O

n-R

es

ista

nc

e (

oh

m)

Ma

x P

ow

er

Lo

ss

(W

) fo

r O

pti

mu

m F

ET

@ 1

15

Va

c045C7

065C7

095C7125C7

190C7225C7

019C7

100kHz150kHz200kHz

400kHz

Ron optimized at full load

Best thermal performance

1200W

13.9W

0.055ohm

45kHz65kHz100kHz150kHz200kHz

400kHz




Figure 9 CoolMOS™ C7 optimum RDS(on) selection charts

Figure 10 CoolMOS™ CP optimum RDS(on) selection charts

1000 10,000

45kHz65kHz

0.01

1

0.1

1

100

10

100

Output Power (W)

Op

tim

um

FE

T O

n-R

es

ista

nc

e (

oh

m)

Ma

x P

ow

er

Lo

ss

(W

) fo

r O

pti

mu

m F

ET

@ 1

15

Va

c045C7

065C7

095C7125C7

190C7225C7

019C7

100kHz150kHz200kHz

400kHz



45kHz65kHz

100kHz150kHz200kHz400kHz

0.01

1

0.1

1

100

10

100 10,000

Output Power (W)

Op

tim

um

FE

T O

n-R

es

ista

nc

e (

oh

m)

Ma

x P

ow

er

Lo

ss

(W

) fo

r O

pti

mu

m F

ET

@ 1

15

Va

c

045C7

065C7

095C7125C7

190C7225C7

019C7

Ron optimized at 50% load

Balanced light/full load efficiency


200kHz

400kHz

45kHz65kHz


1000

0.01

1

0.1

1

100

10

100 10,000

Output Power (W)

Op

tim

um

FE

T O

n-R

es

ista

nc

e (

oh

m)

Ma

x P

ow

er

Lo

ss

(W

) fo

r O

pti

mu

m F

ET

@ 1

15

Va

c

045CP

075CP

099CP125CP

165CP

199CP

250CP

299CP

385CP

600CP520CP



0.01

1

0.1

1

100

10

100 10,000

Output Power (W)

Op

tim

um

FE

T O

n-R

es

ista

nc

e (

oh

m)

Ma

x P

ow

er

Lo

ss

(W

) fo

r O

pti

mu

m F

ET

@ 1

15

Va

c

045CP

075CP

099CP125CP

165CP

199CP

250CP

299CP

385CP

600CP520CP



45kHz65kHz

100kHz150kHz200kHz

400kHz

45kHz65kHz

100kHz150kHz

200kHz

400kHz


400kHz

45kHz65kHz


1000 1000




Figure 11 CoolMOS™ P6 Optimum RDS(on) selection charts

Figure 12 CoolMOS™ C6 optimum RDS(on) selection charts

Since we found that 45 mΩ CoolMOS™ C7 “IPW65R045C7” is the optimum device for our design, we can base on it to calculate different power losses at the worst case of 85 Vac full load condition, as follows:

The MOSFET RMS current across the 60 Hz line cycle can be calculated by the following equation, and consequently the MOSFET conduction loss can be obtained as:

0.01 1

100

10

100 10,000

Output Power (W)

Op

tim

um

FE

T O

n-R

es

ista

nc

e (

oh

m)

Ma

x P

ow

er

Lo

ss

(W

) fo

r O

pti

mu

m F

ET

@ 1

15

Va

c

1

0.1

041P6

070P6

099P6125P6160P6

190P6230P6280P6330P6380P6

600P6



0.01

1

0.1

1

100

10

100 10,000

Output Power (W)

Op

tim

um

FE

T O

n-R

es

ista

nc

e (

oh

m)

Ma

x P

ow

er

Lo

ss

(W

) fo

r O

pti

mu

m F

ET

@ 1

15

Va

c

041P6

070P6

099P6125P6160P6

190P6230P6280P6330P6380P6

600P6



45kHz65kHz

100kHz150kHz200kHz

400kHz


200kHz

400kHz


400kHz


400kHz

1000 1000

100kHz

150kHz

200kHz

400kHz

45kHz

65kHz

0.01 1

100

10

100 10,000

Output Power (W)

Op

tim

um

FE

T O

n-R

es

ista

nc

e (

oh

m)

Ma

x P

ow

er

Lo

ss

(W

) fo

r O

pti

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m F

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@ 1

15

Va

c

1

0.1

041C6

070C6

099C6125C6160C6

190C6

280C6

380C6

600C6520C6

950C6



0.01 1

100

10

100 10,000

Output Power (W)

Op

tim

um

FE

T O

n-R

es

ista

nc

e (

oh

m)

Ma

x P

ow

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Lo

ss

(W

) fo

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pti

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m F

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@ 1

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Va

c

1

0.1

041C6

070C6

099C6125C6160C6

190C6

280C6

380C6

600C6520C6

950C6



45kHz65kHz

100kHz150kHz

200kHz

400kHz

45kHz65kHz

100kHz150kHz

200kHz

400kHz

45kHz

400kHz

45kHz

400kHz

1000 1000




𝐼𝑆.𝑟𝑚𝑠 =𝑃𝑜

𝑉𝑎𝑐.𝑚𝑖𝑛 ∙ √1 −

8 ∙ √2 ∙ 𝑉𝑎𝑐.𝑚𝑖𝑛

3 ∙ 𝜋 ∙ 𝑉𝑜=

1200 𝑊

85 𝑉∙ √1 −

8 ∙ √2 ∙ 85 𝑉

3 ∙ 𝜋 ∙ 400 𝑉= 12.2 𝐴

Eq. 16

𝑃𝑆.𝑐𝑜𝑛𝑑 = 𝐼𝑆.𝑟𝑚𝑠2 ∙ 𝑅𝑜𝑛(100) = 12.2 2 ∙ (0.045 ∙ 1.8) = 12 𝑊

(𝐴𝑠𝑠𝑢𝑚𝑖𝑛𝑔 𝑅𝑜𝑛(100) = 1.8 ∙ 𝑅𝑜𝑛(25) )

Eq. 17

For switching losses calculation, the average input current is used to estimate the losses over the line cycle. The calculation is based on the switching time consideration, where the triangular area between current and voltage changing references to the switching losses.

Figure 13 Simplified turn-on and turn-off waveforms

The average input current is given as:

𝐼𝐿.𝑎𝑣𝑔 =𝑃𝑜

𝑉𝑎𝑐.𝑚𝑖𝑛 ∙

2 ∙ √2

𝜋= =

1200 𝑊

85 𝑉∙

2 ∙ √2

𝜋= 12.71 𝐴

Eq. 18

Turn-on time and loss are:

𝑡𝑜𝑛 = 𝐶𝑖𝑠𝑠 ∗ 𝑅𝑔 ∙ 𝑙𝑛 (𝑉𝑔 − 𝑉𝑡ℎ

𝑉𝑔 − 𝑉𝑝𝑙) + 𝐶𝑟𝑠𝑠 ∙ 𝑅𝑔 ∙ (

𝑉𝑑𝑠 − 𝑉𝑝𝑙

𝑉𝑔 − 𝑉𝑝𝑙)

= 4340 ∙ 10−12𝐹 ∙ 1.8 Ω ∙ 𝑙𝑛 (12 𝑉 − 3.5 𝑉

12 𝑉 − 5.4 𝑉) + 75 ∙ 10−12𝐹 ∙ 1.8Ω ∙ (

400 𝑉 − 5.4 𝑉

12 𝑉 − 5.4 𝑉)

= 10 ∙ 10−9 𝑠

Eq. 19

(𝑉𝑑𝑠 = 𝑉𝑜 = 400 𝑉 , 𝐶𝑟𝑠𝑠 =𝑄𝑔𝑑

𝑉𝑑𝑠

=93 ∙ 10−9𝑛𝐶

400 𝑉= 75 ∙ 10−12𝐹)

𝑃𝑆.𝑜𝑛 = 0.5 ∙ 𝐼𝐿.𝑎𝑣𝑔 ∙ 𝑉𝑜 ∙ 𝑡𝑜𝑛 ∙ 𝑓 = 0.5 ∙ 12.71𝐴 ∙ 400 𝑉 ∙ 10 ∙ 10−9𝑠 ∙ 100 ∙ 103𝐻𝑧 = 2.5 𝑊 Eq. 20

Turn-off time and loss are:

𝑡𝑜𝑓𝑓 = 𝐶𝑟𝑠𝑠 ∗ 𝑅𝑔 ∙ (𝑉𝑑𝑠 − 𝑉𝑝𝑙

𝑉𝑝𝑙) + 𝐶𝑖𝑠𝑠 ∙ 𝑅𝑔 ∙ 𝑙𝑛 (

𝑉𝑝𝑙

𝑉𝑡ℎ)

= 75 ∙ 10−12𝐶 ∙ 1.8Ω ∙ (400 𝑉 − 5.4 𝑉

5.4 𝑉) + 4340 ∙ 10−12𝐶 ∙ 1.8Ω ∙ 𝑙𝑛 (

5.4 𝑉

3.5 𝑉) = 13.3 ∙ 10−9 𝑠

Eq. 21

𝑃𝑆.𝑜𝑓𝑓 = 0.5 ∙ 𝐼𝐿.𝑎𝑣𝑔 ∙ 𝑉𝑜 ∙ 𝑡𝑜𝑓𝑓 ∙ 𝑓 = 0.5 ∙ 12.71𝐴 ∙ 400𝑉 ∙ 13.25 ∙ 10−9𝑠 ∙ 100 ∙ 103𝐻𝑧 = 3.4 𝑊 Eq. 22

The above is the “classic” format for calculating turn-off time and loss; due to the high Qoss of Super Junction MOSFETs, the Coss acts like a nonlinear capacitive snubber, and actual turn-off losses with fast switching can




be up to 50% lower than calculated. The current flow through the drain during turn-off under these conditions is non-dissipative capacitive current, and with fast drive, the channel may be completely turned

off by the onset of drain voltage rise.

Output capacitance Coss switching loss are:

𝑃𝑆.𝑜𝑠𝑠 = 𝐸𝑜𝑠𝑠 ∙ 𝑓 = 11.7 ∙ 10−6 𝐽𝑜𝑢𝑙𝑒 ∙ 100 ∙ 103 𝐻𝑧 = 1.17 𝑊 Eq. 23

Gate drive loss is defined as:

𝑃𝑆.𝑔𝑎𝑡𝑒 = 𝑉𝑔 ∙ 𝑄𝑔 ∙ 𝑓 = 12 𝑉 ∙ 93 ∙ 10−9 𝑛𝐶 ∙ 100 ∙ 103𝐻𝑧 = 0.11 𝑊 Eq. 24

Total MOSFET loss is defined as:

𝑃𝑆.𝑡𝑜𝑡𝑎𝑙 = 𝑃𝑆.𝑐𝑜𝑛𝑑 + 𝑃𝑆.𝑜𝑛 + 𝑃𝑆.𝑜𝑓𝑓 + 𝑃𝑆.𝑜𝑠𝑠 = 19.2 𝑊 Eq. 25

2.3.2 TO-247 4-pin package with Kelvin source connection

In common gate drive arrangements, the fast current transient causes a voltage drop VLS across the parasitic

inductance of the source of the MOSFET that can counteracts the driving voltage. The induced source

voltage, VLS = L*di/dt, can reduce the gate current (Figure 14), therefore lead to slowing down the switching transient and increasing the associated energy loss. On the other hand, the kelvin-source package concept is to exclude the package source inductance and layout inductance from the driving loop, so that the L*di/dt

induced voltage is outside the gate drive loop and not affecting the gate current and switching losses. The

4pin MOSFET recommended for this design would be IPZ65R045C7.

Figure 14 a) Conventional package b) TO-247 4pin package

2.4 Boost diode

Selection of the boost diode is a major design decision in CCM boost converter. Since the diode is hard commutated at a high current, and the reverse recovery can cause significant power loss, noise and current

spikes. Reverse recovery can be a bottle neck for high switching frequency and high power density power supplies. Additionally, at low line, the available diode conduction duty cycle is quite low, and the forward current quite high in proportion to the average current. For that reason, the first criteria for selecting a diode

in CCM boost are fast recovery with low reverse recovery charge, followed by Vf operating at high forward current.

Since Silicon Carbide (SiC) Schottky Diodes have capacitive charge ,Qc, rather than reverse recovery charge, Qrr. Their switching loss and recovery time are much lower compared to silicon ultrafast diode, and will

show an enhanced performance. Moreover, SiC diodes allow higher switching frequency designs, hence, higher power density converters is achieved.




The capacitive charge for SiC diodes are not only low, but also independent on di/dt, current level, and temperature; which is different from Si diodes that have strong dependency on these conditions, as shown

in Figure 15.

Figure 15 Capacitive charge as a function of di/dt for Si pin double diode and SiC diode

The newer generations of SiC diodes are not just Schottky devices, but are merged structure diodes known

as MPS diodes - Merged PN/Schottky (Figure 16). They combine the relatively low Vf and capacitive charge

characteristics of Schottky Diodes with the high peak current capability of PN diodes, while avoiding the

high junction voltage penalty (typically 2.5-3 V at room temperature) of a pure PN wide bandgap diode.

Figure 16 Schottky and Merged PN/Schottky compared

The recommended diode for CCM boost applications is the 650 V CoolSiC™ Schottky Diode Generation 5, which include Infineon’s leading edge technologies, such as diffusion soldering process and wafer thinning technology. The result is a new family of products showing improved efficiency over all load conditions, coming from both the improved thermal characteristics and an improved figure-of-merit (Qc x Vf).

With the high surge current capability of the MPS diode, there is some latitude for the selection of the boost

diode. A simple rule of thumb that works well for a wide input range PFC is 1 A diode rating for each 150 W of output power for good cost/performance tradeoffs, or 1 A diode rating for each 75 W for a premium performance. For example, a 600 W application will only need a 4 A rated diode, but an 8 A diode would perform better at full load. Especially at low-line operation, where the input current is quite high with a

http://www.infineon.com/cms/en/product/power/sicarbide-sic/650v-thinq!-tm-sic-diode-generation-5/channel.html?channel=db3a3043399628450139b0536bed2187




short duty cycle, the higher rated diode will have a much lower Vf at the actual operating current, reducing the conduction losses.

Note that even when using the MPS type SiC diode, it is still preferred to use a bulk pre-charge diode as shown earlier in Figure 3. This is a low frequency standard diode with high I2t rating to support pre-charging

the bulk capacitor to the peak of the AC line voltage; this is a high initial surge current stress (which should be limited by a series NTC) that is best avoided for the HF boost rectifier diode.

In this design example, we are using the 1 A/75 W rule, so for a 1200 W we require a 16 A diode, therefore SiC

diode IDH16G65C5 is selected.

The boost diode average current and conduction loss are:

𝐼𝐷.𝑎𝑣𝑔 =𝑃𝑜

𝑉𝑜=

1200 𝑊

400 𝑉= 3 𝐴

Eq. 26

𝑃𝐷.𝑐𝑜𝑛𝑑 = 𝐼𝐷.𝑎𝑣𝑔 ∙ 𝑉𝑓.𝑑𝑖𝑜𝑑𝑒 = 3 𝐴 ∙ 1.5 𝑉 = 4.5 𝑊 Eq. 27

Diode switching loss, which is carried by the boost MOSFET is:

𝑃𝐷.𝑠𝑤𝑖𝑡 = 0.5 ∙ 𝑉𝑜 ∙ 𝑄𝐶 ∙ 𝑓 = 0.5 ∙ 400𝑉 ∙ 23 ∙ 10−9𝑛𝐶 ∙ 100 ∙ 103𝐻𝑧 = 0.46 𝑊 Eq. 28

Diode total loss is:

𝑃𝐷.𝑡𝑜𝑡𝑎𝑙 = 𝑃𝐷.𝑐𝑜𝑛𝑑 + 𝑃𝐷.𝑠𝑤𝑖𝑡 = 4.5 𝑊 + 0.46 𝑊 = 4.96 𝑊 Eq. 29

2.5 Output capacitor

The output capacitor is sized to meet both of the hold-up time (16.6 ms) and the low frequency voltage

ripple (10 V) requirements. The capacitor value is selected to have the larger value among the two equations

in below:

𝐶𝑜 ≥2 ∙ 𝑃𝑜 ∙ 𝑡ℎ𝑜𝑙𝑑

𝑉𝑜2 − 𝑉𝑜.𝑚𝑖𝑛

2 =2 ∙ 1200 𝑊 ∙ 16.6 ∙ 10−3𝑠𝑒𝑐

(400 𝑉)2 − (340 𝑉)2= 900.9 𝜇𝐹

Eq. 30

𝐶𝑜 ≥𝑃𝑜

2 ∙ 𝜋 ∙ 𝑓𝑙𝑖𝑛𝑒 ∙ ∆𝑉𝑜 ∙ 𝑉𝑜=

1200 𝑊

2 ∙ 𝜋 ∙ 60 𝐻𝑧 ∙ 10 𝑉 ∙ 400 𝑉= 795.8 𝜇𝐹

Eq. 31

→ 𝐶𝑜 = 𝑚𝑎𝑥 (900.9 𝜇𝐹 , 795.8 𝜇𝐹) = 900.9 𝜇𝐹

In this design we use two parallel 560 μF , 450 V, with dissipation factor DF=0.2, consequently the capacitor ESR loss is obtained as below:

𝐸𝑆𝑅 =𝐷𝐹

2 ∙ 𝜋 ∙ 𝑓 ∙ 𝐶𝑜=

0.2

2 ∙ 𝜋 ∙ 120𝐻𝑧 ∙ (2 ∙ 560 𝜇𝐹)= 0.237 Ω

Eq. 32

The capacitor RMS current across the 60Hz line cycle can be calculated by the following equation.

𝐼𝐶𝑜.𝑟𝑚𝑠 = √8 ∙ √2 ∙ 𝑃𝑜

2

3 ∙ 𝜋 ∙ 𝑉𝑎𝑐.𝑚𝑖𝑛 ∙ 𝑉𝑜−

𝑃𝑜2

𝑉𝑜2 = √

8 ∙ √2 ∙ (1200 𝑊)2

3 ∙ 𝜋 ∙ 85 𝑉 ∙ 400 𝑉−

(1200 𝑊)2

(400 𝑉)2= 6.47 𝐴

Eq. 33

𝑃𝐶𝑜 = 𝐼𝐶𝑜.𝑟𝑚𝑠2 ∙ 𝐸𝑆𝑅 = (6.47𝐴)2 ∙ 0.237 Ω = 9.91 𝑊 Eq. 34




2.6 Heat sink design

The MOSFET and boost diode share the same heat sink, thermal resistors are modeled as in Figure 17.

In this evaluation board the maximum heat sink temperature 𝑇𝑆 is regulated to 60°C, so we can calculate the average junction temperature for the MOSFET and diode as follows:

𝑇𝐽.𝑑𝑖𝑜𝑑𝑒 = 𝑇𝑆 + 𝑃𝑑𝑖𝑜𝑑𝑒 ∙ (𝑅𝑡ℎ𝐶𝑆.𝑑𝑖𝑜𝑑𝑒 + 𝑅𝑡ℎ𝐽𝐶.𝑑𝑖𝑜𝑑𝑒) Eq. 35

𝑇𝐽.𝐹𝐸𝑇 = 𝑇𝑆 + 𝑃𝐹𝐸𝑇 ∙ (𝑅𝑡ℎ𝐶𝑆.𝐹𝐸𝑇 + 𝑅𝑡ℎ𝐽𝐶.𝐹𝐸𝑇) Eq. 36

The 𝑇𝑆 can be regulated by choosing a heatsink that with certain airflow can reach the thermal resistance (𝑅𝑡ℎ𝑆𝐴) calculated below:

𝑅𝑡ℎ𝑆𝐴 =𝑇𝑆 − 𝑇𝐴

𝑃𝐹𝐸𝑇 + 𝑃𝑑𝑖𝑜𝑑𝑒

Eq. 37

Where:

𝑅𝑡ℎ𝐽𝐶 : Thermal resistance from junction to case, this is specified in the MOSFET and Diode datasheets.

𝑅𝑡ℎ𝐶𝑆 : Thermal resistace from case to heatsink, typically low compared to the overall thermal resistance, its

value depends on the the interface material, for example, thermal grease and thermal pad.

𝑅𝑡ℎ𝑆𝐴 : Thermal resistance from heatsink to ambient, this is specified in the heatsink datasheets, it depends

on the heatsink size and design, and is a function of the surroundings, for example, a heat sink could

have difference values for RthSA for different airflow conditions.

𝑇𝑆 : Heatsink temperature.

𝑇𝐶 : Case temperature.

𝑇𝐴 : Ambient temperature.

𝑃𝐹𝐸𝑇 : FET total power loss.

𝑃𝑑𝑖𝑜𝑑𝑒 : Diode total power loss.

PFET RthJC.FET RthCS.FET

RthJC.diode RthCS.diode

RthSA

TJ.FET

TJ.diode

TS TA

TC.FET

TC.diodePdiode

PFET+Pdiode

Figure 17 Schematic of thermal network




ICE3PCS01G PFC boost controller 3The ICE3PCS01G is a 14pins controller IC for power factor correction converters. It is suitable for wide range line input applications from 85 to 265 VAC and with overall efficiency above 97%. The IC supports the converters in boost topology and operates in continuous conduction mode (CCM) with average current control.

The IC operates with a cascaded control; the inner current loop and the outer voltage loop. The inner current loop of the IC controls the sinusoidal profile for the average input current. It uses the dependency of the PWM duty cycle on the line input voltage to determine the corresponding input current. This means the average input current follows the input voltage as long as the device operates in CCM. Under light load condition, depending on the choke inductance, the system may enter into discontinuous conduction mode

(DCM) resulting in a higher harmonics but still meeting the Class D requirement of IEC 1000-3-2.

The outer voltage loop of the IC controls the output bulk voltage and is integrated digitally within the IC.

Depending on the load condition, internal PI compensation output is converted to an appropriate DC voltage which controls the amplitude of the average input current.

The IC is equipped with various protection features to ensure safe operating for the system and the device.

3.1 Soft startup

During power up when the VOUT is less than 96% of the rated level, internal voltage loop output increases from initial voltage under the soft-start control. This results in a controlled linear increase of the input current from 0 A as can be seen in Figure 17. This helps to reduce the current stress in power components.

Once VOUT has reached 96% of the rated level, the soft-start control is released to achieve good regulation

and dynamic response and VB_OK pin outputs 5 V indicating PFC output voltage in normal range.

3.2 Gate switching frequency

The switching frequency of the PFC converter can be set with an external resistor RFREQ at pin FREQ with reference to pin SGND. The voltage at pin FREQ is typical 1V. The corresponding capacitor for the oscillator is

integrated in the device and the RFREQ/frequency is given in Figure 18. The recommended operating

frequency range is from 21 kHz to 250 kHz. As an example, a RFREQ of 43 kΩ at pin FREQ will set a switching

frequency fSW of 100 kHz typically.

Figure 18 Frequency setting

Frequency vs Resistance

0

20

40

60

80

100

120

140

160

180

200

220

240

260

10 20 30 40 50 60 70 80 90 100 110 120 130 140 150 160 170 180 190 200 210 220 230 240 250

Resistance/kohm

Fre

qu

en

cy/k

Hz

19.223243100

20.22214990

21.22105580

232006270

251917460

26.21698650

29.515010640

31.514014130

3413021120

3612024917

4011027815

Frequency

/kHz

Resistance

/kohm

Frequency

/kHz

Resistance

/kohm




3.3 Protection features

3.3.1 Open loop protection (OLP)

The open loop protection is available for this IC to safe-guard the output. Whenever voltage at pin VSENSE falls below 0.5 V, or equivalently VOUT falls below 20% of its rated value, it indicates an open loop condition (i.e. VSENSE pin not connected). In this case, most of the blocks within the IC will be shutdown. It is implemented using a comparator with a threshold of 0.5 V.

3.3.2 First over-voltage protection (OVP1)

Whenever VOUT exceeds the rated value by 8%, the first over-voltage protection OVP1 is active. This is

implemented by sensing the voltage at pin VSENSE with respect to a reference voltage of 2.7 V. A VSENSE voltage higher than 2.7 V will immediately block the gate signal. After bulk voltage falls below the rated value, gate drive resumes switching again.

3.3.3 Peak current limit

The IC provides a cycle by cycle peak current limitation (PCL). It is active when the voltage at pin ISENSE

reaches -0.2 V. This voltage is amplified by a factor of -5 and connected to comparator with a reference

voltage of 1.0 V. A deglitcher with 200 ns after the comparator improves noise immunity to the activation of this protection. In other words, the current sense resistor should be designed lower than -0.2 V PCL for normal operation.

3.3.4 IC supply under voltage lockout

When VCC voltage is below the under voltage lockout threshold VCC,UVLO, typical 11 V, IC is off and the gate

drive is internally pull low to maintain the off state. The current consumption is down to 1.4 mA only.

3.3.5 Bulk voltage monitor and enable function (VBTHL_EN)

The IC monitors the bulk voltage status through VSENSE pin and output a TTL signal to enable PWM IC or control inrush relay. During soft-start once the bulk voltage is higher than 95% rated value, pin VB_OK outputs a high level. The threshold to trigger the low level is decided by the pin VBTHL voltage adjustable

externally.

When pin VBTHL is pulled down externally lower than 0.5 V most function blocks are turned off and the IC enters into standby mode for low power consumption. When the disable signal is released the IC recovers by soft-start.




Efficiency and power losses modeling 4The design example discussed in this document was modeled in Mathcad®. All of the power losses equations

were written as a function of the output power in order to be able to plot an estimated efficiency curve

across the output power range as shown in Figure 19.

Figure 19 Calculated efficiency @ 85VAC vs experimental efficiency @ 90VAC

Figure 20 shows a breakdown of main power losses at the full load of both low and high line conditions.

Figure 20 Breakdown of main power losses

Figure 21 shows the simulated and experimental inductor current at the low line full load condition.

Figure 21 Simulated and experimental Inductor current waveforms at low-line full load condition

MOSFET losses26%

Boost diode losses

7%Inductor

losses20%

Rectifier bridge loss

34%

Output capacitor

loss13%

85Vac 100% load Total power loss = 74.1W

MOSFET losses16%

Boost diode losses20%

Inductor losses18%

Rectifier bridge

loss37%

Output capacitor

loss9%

230Vac 100% loadTotal power loss = 25.3W




Experimental results 5

Figure 22 Evaluation board

All test conditions are based on 60°C heat sink temperature.

5.1 Load and line measurement data

The test data of the evaluation board under different test conditions are listed in Table 2. The corresponding efficiency waveforms are shown in Figure 23 and Figure 24.

Efficiency test data with IPZ65R045C7 & IDH12G65C5 @ 100kHz , Rg= 1.8 Ω Table 2

Input VIN

[V]

IIN

[A]

PIN

[W]

UOUT

[V]

IOUT

[A]

POUT

[W] Efficiency

[%]

Power

factor

90 VAC 88.88 14.3962 1278.5 402.05 2.984 1200.02 93.829 0.9996

89.11 10.8743 968.8 402.09 2.282 917.68 94.719 0.9997

89.33 8.0877 722.4 402.12 1.711 688.38 95.290 0.9998

89.51 5.3667 480.2 402.15 1.141 459.05 95.597 0.9996

89.74 2.6953 241.5 402.18 0.571 229.82 95.161 0.9984

230 VAC 229.5 5.3365 1222.1 402.03 2.985 1200.01 98.186 0.9976

229.6 4.431 1014.9 402.06 2.479 996.66 98.198 0.9975

229.7 3.3171 758.7 402.10 1.852 744.59 98.138 0.9956

229.8 2.2298 508.8 402.12 1.239 498.18 97.910 0.9929

229.9 1.1304 253.4 402.16 0.612 246.15 97.120 0.9752




Figure 23 High-line efficiency with IPZ65R045C7 & IDH12G65C5 @ 100kHz, Rg= 1.8 Ω

Figure 24 Low-line efficiency with IPZ65R045C7 & IDH12G65C5 @ 100kHz, Rg= 1.8 Ω

5.2 Conductive EMI test

Compliance with EN55022 standard is a very important quality factor for a power supply. The EMI has to

consider the whole SMPS and is split into radiated and conductive EMI consideration. For the described evaluation PFC board it is most important to investigate on the conducted EMI-behavior since it is the input stage of any SMPS below a certain power range, as shown in Figure 25.




Figure 25 Conductive EMI measurement of the board with resistive load

5.3 Startup behavior

During power up when the VOUT is less than 96% of the rated level, internal voltage loop output increases

from initial voltage under the soft-start control. This results in a controlled linear increase of the input

current from 0 A thus reducing the current stress in the power components as can be seen on the yellow waveform in Figure 26.

Figure 26 Waveform capture during low-line startup




Board design 6

6.1 Schematics

Figure 27 Evaluation board schematic




6.2 PCB layout

Figure 28 PCB top layer view

Figure 29 PCB bottom layer view




6.3 Bill of material

Table 3 Bill of material

Designator Value Description

B1, B2 closed with 0 Ω Placeholder for Ferrite Bead, 0 Ω resistor

Bias1 12 V Bias Bias adapter

C1 10 µ 25 V

C2 4n7 25 V

C3 10 n 25 V

C4, C5 1 µ x-capacitor

C6 4.7 n 25 V

C7 10 n 25 V

C8, C30 100n500 V VJ1825Y104KXEAT

C10, C31 1 n 25 V

C11 10 µ 25 V

C12, C25, C32 100 n 25 V

C13 100 n 25 V

C14, C15 1 µ 25 V

C16 100 µ 25 V

C17, C18 1.1 n Y-capacitor

C19, C20 3.3 n Y-capacitor

C21, C24 560 µ EETHC2G561KA or EKMR421VSN561MR50S

C22 1u_400 V BFC237351105; Farnel 1215540

C23 1.5u_400 V

C26 220 n 25 V

C27 10 µ 25 V

C28 470p 25 V

C29 22n 25 V

D1 SS26

D2, D3 1N4148

D4 1N5408

D6 short 0 Ω

D10 ES1C 1 A 150 V fast diode

DUT1 IPZ65R045C7

D_Z3 ZMM15 1N4734A

EMI_1 not placed EMI Adapter

GL1, GL2 GSIB2580 GSIB2580

IC1 TDA2030 Mount with M2.5x6

IC2 LM4040 LM4040D20IDBZRG4

IC3 ICE3PCS01G PFC_CCM_Controller

IC4 1EDI60N12AF 6 A_isolated_MOSdriver

IC5 IFX91041 1.8A Step down switching regulator

J1, J11 Jumper_3Pin SPC20486

J2 Current measure bridge 1.25 mm isolated copper wire

J3 Current measure bridge U-shape-Cu-wire 1.25 mm 2 cm distance

J6 open Solder jumper; 4 pin as 3 pin

J7, J12 close with solder Solder jumper; driver ground to SS, Solder jumper; isolated driver power

J8, J10 open Solder jumper; 3pin ground, Solder jumper; driver power none isolated

J9 close with solder Solder jumper; isolated driver power

K1 SK426 100 mm long; mound with 2xM4x15

K2 KM75-1 KM75-1 +4clip 4597; Fischer

KL1 BNC Oscilloscope_ Funtion generator

KL01 HV in GMSTBVA 2,5 HC/ 3-G-7,62

KL01-S Complement GMSTB 2,5 HCV/ 3-ST-7,62

KL02 Vin_sense GMSTBVA 2,5 HC/ 2-G-7,62

KL02-S Complement GMSTB 2,5 HCV/ 2-ST-7,62




L1 L_PFC 2times 77083A7 64wind_1.15 mm

L2 10 A 100 µH Würth 744824101

L3 8120-RC BOURNS_8120-RC_2m4H_17 A

L4 33 µH 74454133

LED1 red Power on LED

LED2, LED3, LED4,

LED5 blue Power on LED

M1, M2 Fan 60mm PMD1206PTB1-A

M1, M2 finger guard for Fan 60mm LZ28CP

PWM-Signal SMA Oscilloscope_ Function generator

R1, R3, R13, R20, R56 1 k 5%

R2, R8, R15, R44 10 k 5%

R4 5 k 67WR20KLF

R5 680 5%

R6 220R 5%

R7, R11 10R 5%

R9, R16 20R 3314G-1-200E

R10, R25 47R 5%

R12, R42 330 k 5%

R14, R19 2M 5%

R17 27 k 5%

R18 36 k 5%

R21 LTO100 4R7 include two 20F2617 Bürklin connector

R22, R23 500 k 10%

R24 0R005 FCSL90R005FE

R27 np

R28 20 k 23AR20KLFTR

R29 22 k 5%

R30 1 k 10 V

R31 100 k 67WR100KLF

R35 2R 5%

R36 np

R37 np 500 k 5%

R45 200 k 5%

REL1 AZ762 12 V

REL2 G6D_1A_ASI 12 V

R_NTC1 5k B57560G502F mound in K1 under MOS

R_NTC2 3R3 R_SL22

S1, S2, S3, S4, S5, S6 SCREW_M4 3 cm Distance holder

S1, S2, S3, S4, S5, S6 M4 Screw nut M4 Screw nut

S1, S2, S3, S4, S5, S6 washer M4 washer M4

Vin HV_in GMSTBA_2.5HC_3G7.62

Vout Vout GMSTBA_2.5HC_2G7.62

Vout_sense Vout_sense GMSTBVA_2.5HC_2G7.62

X1 np (Heat sink) thermocouple plug

X2 np (MOS1) thermocouple plug

X3 np (Diode) thermocouple plug

X4 np (Choke) thermocouple plug

X5 np (MOS2) thermocouple plug

X6 Rg_4pin SPC20485

X7 Rg_3pin SPC20485

X8, X9 KL_STANDARD_2 SPC20485

X12 Np for adapter power supply




References 7 F. Stueckler, E. Vecino, Infineon Technologies Application Note: “Coo lMOSTM C7 650V Switch in a Kelvin [1]

Source Configuration”. May 2013.

http://www.infineon.com/dgdl/Infineon+-+Application+Note+-+TO-247-4pin+-+650V+CoolMOS™ +C7+Switch+in+a+Kelvin+Source+Configuration.pdf?folderId=db3a304333b8a7ca0133c6bec0956188&fi

leId=db3a30433e5a5024013e6a9908a26410

J. Hancock, F. Stueckler, E. Vecino, Infineon Technologies Application Note: “CoolMOS™ C7 : Mastering [2]the Art of Quickness”. April 2013.

http://www.infineon.com/dgdl/Infineon+-+Application+Note+-+650V+CoolMOS™ +C7+-

+Mastering+the+Art+of+Quickness.pdf?folderId=db3a304333b8a7ca0133c6bec0956188&fileId=db3a30

433e5a5024013e6a966779640b

ICE3PCS01G controller datasheet. [3]

http://www.infineon.com/dgdl/Infineon-ICE3PCS01-DS-v02_00-

en.pdf?folderId=5546d4694909da4801490a07012f053b&fileId=db3a304329a0f6ee0129a67ae8c02b46

Power Factor Correction (PFC) Parts Selection Guide [4]

http://www.infineon.com/dgdl/Infineon+-+Selection+Guide+-+PFC+-+Power+Factor+Correction+-+CoolMOS™ +-+SiC+Diodes+-

+Controllers.pdf?folderId=db3a30433e5a5024013e6a288c8f6352&fileId=db3a30433e5a5024013e6a35cb806364

http://www.infineon.com/dgdl/Infineon+-+Application+Note+-+TO-247-4pin+-+650V+CoolMOS+C7+Switch+in+a+Kelvin+Source+Configuration.pdf?folderId=db3a304333b8a7ca0133c6bec0956188&fileId=db3a30433e5a5024013e6a9908a26410



http://www.infineon.com/dgdl/Infineon+-+Application+Note+-+650V+CoolMOS+C7+-+Mastering+the+Art+of+Quickness.pdf?folderId=db3a304333b8a7ca0133c6bec0956188&fileId=db3a30433e5a5024013e6a966779640b



http://www.infineon.com/dgdl/Infineon-ICE3PCS01-DS-v02_00-en.pdf?folderId=5546d4694909da4801490a07012f053b&fileId=db3a304329a0f6ee0129a67ae8c02b46

http://www.infineon.com/dgdl/Infineon-ICE3PCS01-DS-v02_00-en.pdf?folderId=5546d4694909da4801490a07012f053b&fileId=db3a304329a0f6ee0129a67ae8c02b46

http://www.infineon.com/dgdl/Infineon+-+Selection+Guide+-+PFC+-+Power+Factor+Correction+-+CoolMOS+-+SiC+Diodes+-+Controllers.pdf?folderId=db3a30433e5a5024013e6a288c8f6352&fileId=db3a30433e5a5024013e6a35cb806364







Symbols used in formulas 8

Table 4 Symbols used in formulas

Vac Input voltage

Vac.min Minimum input voltage

Vo Output voltage

Po Output power

f Switching frequency

T Switching time period

fline line frequency

L Filter inductor

%Ripple Inductor current ripple percentage to input current

DCR Inductor DC resistance

Iin.rms Input rms current

IL.rms Inductor rms current

IL.avg Inductor average current across the line cycle

IL.pk Inductor peak current

PL.cond Inductor conduction loss

Vf.bridge Bridge diode forward voltage drop

Pbridge Bridge power loss

Ron(100C) MOSFET on-resistance at 100oC

Qgs MOSFET gate-source charge

Qgd MOSFET gate-drain charge

Qg MOSFET total gate charge

Rg MOSFET gate resistance

Vpl MOSFET gate plateau voltage

Vth MOSFET gate threshold voltage

ton MOSFET turn-on time

toff MOSFET turn-off time

Eoss MOSFET output capacitance switching energy

IS.rms MOSFET rms current over the line cycle

PS.cond MOSFET conduction loss

PS.on MOSFET turn-on power loss

PS.off MOSFET turn-off power loss

PS.oss MOSFET output capacitance switching loss

PS.gate MOSFET gate drive loss

ID.avg Boost diode average current

Vf.diode Boost diode forward voltage drop

Qrr Boost diode reverse recovery charge




Vac Input voltage

PDcond Boost diode conduction loss

PD.sw Boost diode switching loss

Co Output capacitor

ESR Output capacitor resistance

thold Hold-up time

Vo.mi Hold up minimum output voltage

∆Vo Output voltage ripple

ICo.rms Output capacitor rms current

PCo Output capacitor conduction loss

Revision history

Major changes since the last revision

Page or reference Description of change

-- First release (Revision 1.0)

Figure 8-12 Corrected output power on y-Axis: 10,000(Revision 1.1.)

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