Laporan Project 2 Nik Khatijah Nik Mustaffa-1

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    STUDY AND DESIGN OF A DUAL-MODE MICROSTRIP OPEN-LOOP

    RESONATORS AND FILTERS USING SONNET LITE SIMULATION SOFTWARE

    SUBMITTED BY:

    NIK KHATIJAH BINTI NIK MUSTAFFA

    (HE100082)

    SUPERVISOR:

    DR SAMSUL HAIMI BIN DAHLAN

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    DESIGNING A DUAL MODE MICROSTRIP OPEN LOOP RESONATORS AND

    FILTERS AT 5.7 GHZ FOR WLAN APPLICATIONS.

    ABSTRACT

    Bandpass filters play a significant role in wireless communication systems. Transmitted and

    received signals have to be filtered at a certain center frequency with a specific bandwidth. A

    dual mode microstrip open loop resonators and filter at 5.7 GHz for wireless local area network

    (WLAN) are designed using sonnet lite simulation software. Reflection coefficient and

    transmission coefficient of the filter are reported. The simulated and measured filter performance

    are presented.

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    CHAPTER 1

    INTRODUCTION

    1.1

    BACKGROUND

    With the rapid development of wireless and mobile communication systems, there is an

    increasingly demands for new technologies to meet the challenge for smaller size, high

    performance, multi-frequency and cheaper manufacturing cost. Filters are one of the most

    important components must be found in the transmitter or receiver. Filtering application

    is most crucial in a system as it will discriminate between wanted and unwanted signal

    frequencies. Bandpass filter is a passive component which is able to select signals inside

    a specific bandwidth at a certain center frequency and reject signals in another frequency

    regions, which have the potential to interfere the information signals. In designing the

    bandpass filter, we are faced the question, what is the maximal loss inside the pass

    region, and the minimal attenuation in the reject or stop regions, and how the filter

    characteristic must look like in transition region. Thus, filter size is one of the concern in

    the filter design. The filter is design to ensure that it can fit into the electronic devices.

    Besides, the losses of the filter are playing an important role because they will affect the

    performance of the filters. Microstrip filters is widely used and apply in many

    RF/microwave circuits and systems [1]. In general, microstrip bandpass filters can be

    designed using single or dual mode resonators. Band pass filter require precise

    transmission characteristics that allow a desired band of signals to pass through the two

    port network. Thus, between a transmitter and transmitting antenna, a band pass filter

    may be used to attenuate unwanted signals and harmonic components that may cause

    interference to other users of the electromagnetic spectrum. Conversely, between an

    antenna and a receiver, a band pass filter will reject out of band signals that may cause

    interference within the receiver, especially if they are at a high signal level in comparison

    with the desired signals. Dual mode resonators are attractive because each dual mode

    resonator can be used as a doubly tunable resonant circuit. Therefore the number of

    resonators in a filter can be reduced by half, thus resulting in a compact configuration.

    Various kinds of dual mode resonators have been investigated, including the circular ring

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    [2], square loop [3], and triangular patch [4]. Meanwhile, several new types of dual mode

    resonators [5] have been reported for filter applications recently. Practically, some special

    applications require high selectivity on only one side of the pass band, but less or none on

    the other side. In such cases, using filters with asymmetric frequency responses would be

    desirable [6].

    In this project, the type of miniature microstrip dual-mode open loop resonator for filter

    applications is introduced. The dual-mode open loop resonator is developed. The open-

    loop resonator is well known for its flexibility to design cross-coupled resonator filters, as

    well as its compact size. It will be shown that the proposed dual-mode open-loop

    resonator has a size that is the same as the single-mode open-loop resonator, which,

    however, is much smaller than the conventional dual-mode loop resonator [3]. The size

    of the dual-mode open-loop resonator is only approximately one-quarter of the dual-

    mode loop resonator, which is a significant size reduction. There have some different of

    distinct characteristics between the dual-mode open-loop resonator and the conventional

    dual-mode loop resonator. The study of filter parameters and the effect to filter response

    was also presented. This information is important especially for circuit optimization

    work.

    In this project bandpass filters based on dual-mode resonator are design at 5.7 GHz and

    verified using SONNET LITEPlus software to optimized and simulate the circuits with

    the microstrip calculator software to calculate related formulae for the design.

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    1.2 AIMS AND OBJECTIVES

    There are several objectives that have been outline to complete this analysis of dual

    mode microstrip open loop resonators and filters at 5.7 GHz. This project designs a

    dual-mode microstrip open loop resonators and filters at 5.7 GHz by using SONNET

    Lite simulation software. This objective helps to gain fundamental knowledge in

    microstrip filter. This project also to simulate and analyze the performance of

    bandpass filter operating at frequency range of 5 GHz to 6 GHz and to measure the

    frequency responses of the dual mode resonator and filter.

    The project aims to design a microstrip bandpass filter which is simple and easy to

    fabricate. The design filter must be a dual mode bandpass device that resonates in the

    frequency range of 5 GHz to 6 GHz.

    On the other hand, this project aims to design a dual-mode microstrip open loop

    resonators and filters at 5.7 GHz where the filter will be unique, simple, and high-

    performing. Through this project, we will have a better understanding of the

    microstrip bandpass filter properties.

    1.3 PROJECT MOTIVATION

    The project motivation is to design the microstrip bandpass filters that are to fulfill

    the masters project requirement. After completing the literature review throughout

    the journals, knowledge in filter design has widen. The goal of this project is todesign a simple, high performing, and small size microstrip bandpass filter that has

    results comparable to those published.

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    CHAPTER 2

    LITERATURE REVIEW

    2.1 Microstrip Filter

    In modern wireless and mobile communication systems, Radio Frequency (RF) and microwave

    filter are important and essential components. No doubt, filters are playing an important role in

    the RF and microwave applications. Filters with smaller size, lighter weight, higher performance,

    and lower insertion loss are of high demand. Microstrip filters can fulfil the requirements stated

    above. Besides wireless and mobile communication systems, low-temperature co fired ceramics

    (LTCC), high temperature superconductor (HTS), monolithic microwave integrated circuits

    (MMIC), micro electromechanic system (MEMS) and micromachining technology, have driven

    the rapid development of the new microstrip filter than other microwave and RF filters.

    Microstrip filters can be designed in various patterns depending on different requirements. Each

    microstrip filter consists of its individual properties and characteristics. The common microstrip

    designs that available in market are such as rectangular patch filter, circular patch filter,

    triangular patch filter and etc.

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    2.1.1 Microstrip Lines

    Microstrip is a planar transmission line that basically consists of a thin-film strip in intimate

    contact with one side of a flat dielectric substrate, with a similar thin-film ground-plane

    conductor on the other side of the substrate. It is similar to stripline and coplanar waveguide, and

    it is possible to integrate all three on the same substrate. According to Encyclopedia, microstrip

    was developed by ITT laboratories by Grieg and Engelmann in December 1952 as a competitor

    to stripline. According to Pozar, the early microstrip work used fat substrates, which allowed

    non-TEM7 waves to propagate which makes results unpredictable. In the 1960s the thin version

    of microstrip started became popular. Microstrip can be fabricated using printed circuit board

    (PCB) technology and is used to convey microwave frequency signals. It consist of a conducting

    strip separate from a ground plane by a dielectric layer which known as substrate.

    Microstrip can be used to design antennas, couplers, filters, power dividers and etc. The material

    costs are lower than the traditional waveguides since the entire microstrip is just a patterned

    metal and a ground plane only, and it also much lighter and more compact than the waveguides.

    Since microstrip is not enclosed as a waveguide, it is susceptible to crosstalk and unintentional

    radiation. Unlike stripline, microstrip can mount all the active components on top of the board.

    However, the disadvantages are that microstrip generally has lower power handling capacity and

    higher losses. External shield may be needed for circuits that require higher isolation such as

    switches. In this project, microstrip has been chosen for the filter design because it is a proven

    technology and the shielding problem can be solved easily by encapsulating the filter in a

    metallic box. Besides, crosstalk and unintentional radiation are not an issue in this project as

    well.

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    2.1.1.1 Dielectric Constant

    Microstrip can be fabricated on different substrates based on the requirement of the design. For a

    lower cost, microstrip devices can be built on the ordinary Duroid 6006 substrate. Dielectricconstant is a measure of the extent to which it concentrates electrostatic lines of flux. For

    example, FR-4 has a dielectric constant of 4.4 and R6006 duroid has 6.15 at frequency of 10

    GHz. And by definition, the dielectric constant of vacuum is equal to 1. Dielectric constant is

    frequency dependent.

    2.1.1.2 Effective Dielectric Constant

    In general, the dielectric constant of the substrate will be different (and greater) than air, so the

    wave is travelling in an inhomogeneous medium. In consequence, the propagation velocity is

    somewhere between the speed of radio waves in the substrate, and the speed of radio waves in

    air. This behaviour is commonly described by stating the effective dielectric constant of the

    microstrip. Effective dielectric constant is an equivalent dielectric constant of an equivalent

    homogeneous medium. As part of the fields from the microstrip conductor exists in air, the

    effective dielectric constant is less than the substrate's dielectric constant.

    2.1.1.3 Characteristic Impedance

    The characteristic impedance, Zo is important in microstrip lines design, it will affects the

    reflection loss, S11. The reflection loss is related to the characteristic impedance and also load

    impedance by formula, S11= . In order to minimize the reflection loss at the input port, the

    characteristic impedance of microstrip must be equal to the input of the impedance. Most of the

    input ports are designed at 50 ohm to prevent any reflections. The characteristic impedance is

    also a function of height (H) and width (W) of microstrip lines. Note that effective dielectric

    constant value is required for the microstrip characteristic impedance calculation. The

    characteristic impedance of microstrip lines can be calculated by using formulas below.

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    When

    (

    )

    When

    *

    +

    The very first step in this project is to design the microstrip lines that can achieve 50 ohm of

    characteristic impedance, . From the formulas above, there are three variables in thecalculation which are Width (W) and Height (H). Since the and height is constant forthe substrate, the only variable is the width of the microstrip line.

    Since the formula is quite complicated, we have to substitute different W value in order to get 50

    ohm characteristic impedance. At first, it has to verify the ratio of W/H and determine which

    formula to be used. It is time consuming for the calculation on different substrate. Fortunately,

    some of the web application and software are able to calculate the width of the microstrip lines

    by entering the desire characteristic impedance, height of the substrate and dielectric constant of

    the substrate. Then the software will perform the calculation within seconds and shows the value.

    It has simplified the microstrip lines design work and most importantly consume less time. For

    instance, one of the useful software was used for characteristic impedance calculation in this

    project is microstrip calculator. The software is easy to control and able to get the width value

    easily by key in height of the substrate, thickness of the strip, and dielectric constant of the

    substrate. The microstrip width calculation will be cover in Methodology part.

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    2.1.2 Microstrip Resonators

    Microstrip resonators can be designed in various patterns. Before designing the microstrip band

    pass filter, it has to do some literature studies through IEEE Xplore about the microstrip

    resonators. Besides, it is also to ensure that the effort spent in design has not been researched by

    others nor published in the IEEE Xplore before. The author has selected the common types of

    resonators to be discussed in the following chapters. The types of resonators that covered are

    dual mode microstrip open loop resonator.

    2.1.2.2 Microstrip Open Loop Resonators

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    CHAPTER 3

    METHODOLOGY

    There are different method carrying out research in the field of RF/Microwave system, like

    experiment and simulation. The building of a design and the testing of it in practical experiment

    is an expensive and time-consuming method. To minimize time and cost, this research focus on

    simulation experiment based on Sonnet Lite simulation software in order to simulate frequency

    response for resonators and filters.

    The Sonnet EM analysis is based on a method-of-moments technique. The circuit metal is

    first meshed into small subsections. The EM coupling between each possible pair of subsections

    is calculated and this fills a big matrix. The matrix is inverted, yielding all currents everywhere

    in the circuit metal. This, in turn, determines things like S-parameters, which can be used in other

    analysis programs.

    The Sonnet analysis has very high accuracy because it calculates all couplings between

    subsections using a 2-D Fast Fourier Transform (FFT). In signal processing, application of the

    FFT requires uniform time sampling of a signal. In EM analysis, the FFT requires uniform space

    sampling across the two dimensions of the substrate surface. Thus the analyzed circuit metal falls

    on a uniform underlying FFT mesh. The FFT approach also requires the circuit to exist inside a

    conducting, shielding box. Approximations are available to include radiation. A common

    alternative technique requires a four dimensional numerical integration to calculate each pair-

    wise coupling. This approach has the advantage of eliminating the underlying uniform mesh,

    however, it comes at the cost of sometimes extensive numerical integration time and the

    additional error involved in the numerical integration process. The FFT used by Sonnet quickly

    calculates all couplings to full numerical precision. In contrast to the FFT approach, numerical

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    integration requires an open environment allowing radiation while a shielded environment must

    be approximated.

    When and if a serious planar numerical integration based EM tool becomes freely

    available, the user will find the relative advantages and disadvantages tend to compliment

    Sonnet's FFT based tool.

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    CHAPTER 4

    DATA ANALYSIS AND RESULT

    4.1DESIGN OF THE DUAL MODE MICROSTRIP OPEN LOOP RESONATOR

    In this project, all the simulation are done by Sonnet Lite software. The design and result of

    dual mode microstrip open loop resonator is shown in Figure 4.1 (a) and Figure 4.1(b) below.

    Figure 4 (a) shows the layout of a dual mode microstrip open loop resonator on a substrate

    with a relative dielectric constant of 10.8 and a thickness of 1.27 mm. All dimensions are inmillimeters. A loading element with a variable parameter W is tapped from inside onto the

    open loop. When W varies, the modal resonant characteristic is changed. For the

    demonstration, we use a Sonnet Lite simulator to simulate the resonant frequency response of

    the proposed dual mode open loop resonator. To excite the resonator, two ports are weakly

    coupled to the resonator. The simulated result are plotted in Figure 4.1(b).

    Figure 4.1 (a): Layout of a dual-mode microstrip open loop resonator

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    For a loading element, W= 9.3 mm, two modes exhibits the same resonant frequency. The

    simulated result are in Figure 4 (b). Two modes will split if W is increased or decreased

    from value W = 9.3 mm.

    Specify unit for length and frequency where length is in mm and frequency in GHz.

    Dielectric layer. Set the thickness 1.27 mm and relative dielectric constant 10.8

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    Box size and cell size setting.

    Metal type

    Specify Frequency from 1.04 GHz to 1.12 GHz.

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    Figure 4.1(b) : Graph S11 and S21.

    Figure 4.1 (b) shows the modal resonant characteristic of the proposed dual mode microstrip

    open loop resonator for g=0.9 mm, d= 1.1 mm and w=9.3 mm. Figure 3.3 shows the simulated

    and measured return losses and insertionlosses. Rrom the figure, the measured center frequency is 1.655 GHz, which is very

    close to the simulated value of 1.69 GHz (error 2.07%). The measured and simulated

    3-dB bandwidths ( 11S10 dB) are given by 9.09% and 12.79%, respectively. Alsoshown in the figure are the simulated S11 and S21by using the network model. With30

    reference to the figure, the measured center frequency is 1.65 GHz, which is very

    close to the simulated value of 1.69 GHz (error 2.37%). The measured 3-dB

    bandwidth ( 11S10 dB) by using network model is given by 10.40%. Withreference to the figure, the models predict the S11 and S21 reasonably well.

    The error might caused by the gap distance. From the simulation design the

    capacitive coupling gap is 0.01 mm. However, such narrow gap is difficult tofabricate by the equipment in the laboratory. Few Duroid 6006 substrates have been

    fabricated and experiment. At the end of the day this is the best result the author able

    to collect.

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    Figure 4.2 (a): Layout of a dual-mode microstrip open loop resonator with W = 9.1

    mm.

    Figure 4: Graph S11 and S21 for w=9.10

    The smaller W results in a mode being shifted to a higher frequency.

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    Figure 5: Layout of a dual-mode microstrip open loop resonator with W = 9.5 mm.

    Figure 6: Graph for S11 and S21

    The larger W shifts a mode to a lower frequency.

    In all the cases, only one modal resonant frequency is affected, while the other one is

    hardly changed. It is shows that the even mode is affect to the resonant frequency but the odd

    mode in unaffected to the resonant frequency. The tapping point of the loading element, W is a

    virtual ground for the odd mode. Hence, the loading element does not affect the odd mode

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    characteristic and its resonant frequency. The loading element is affected to the even mode

    where the change of W makes the resonant frequency of the even mode shifted.

    The transmission zero is allocated on the high side of the two modes when the even

    mode frequency is higher than the odd mode frequency. The transmission zero is allocated on the

    low side when the even mode frequency is lower than the odd mode frequency. The transmission

    zero is closely with the even mode.

    The unique property of the even mode will proposed the dual mode resonator for

    designing the filters.

    Another characteristic of the dual mode open-loop resonator is the two modes are not

    coupled to each other even after the modes are split. From the theory, if the two split mode

    frequencies are equal to the two self resonant frequency, respectively there is no coupling

    between the two resonators.

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    FILTER APPLICATION

    n-pole filter can be constructed using n/2 dual mode open loop resonators. Basic coupling

    structures of this filter can be found in [6] where no coupling between the even and odd

    modes has been implicated.

    To design this filter, needs to allocate the modal resonant frequencies within the desired

    passband.

    For a given center frequency, the perimeter of the open loop is approximately a half

    wavelength, which sets the resonant frequency of the odd mode. Once the loop size is found,

    the odd mode resonant frequency can be tuned to the desired one by varying the gap gof the

    dual mode open loop resonators.

    When g is changed to the higher value, the odd mode resonant frequency is effectively

    shifted. The even mode resonant frequency is mainly determined by the size of the loading

    element and can easily be tuned by several parameters such as W and d.

    The W controls the capacitive loading so as to change the even mode resonant frequency. In

    addition, the even mode resonant frequency can be easily controlled by varying the

    dimension of d, when d is changed from 0.3 to 1.3 mm, the even mode resonant frequency is

    effectively shifted to the higher value, 0.983 to 1.152 GHz.

    The smaller d, results in a larger inductive loading and a lower resonant frequency

    for the even mode. The easy control of modal resonant frequencies allows one to control

    filterbandwidth because the separation of the modal resonant frequencies is proportional to

    the bandwidth.

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    Bandpass filter

    Figure 7 shows the filter where the characteristic of filter is a finite frequency zero

    on the low side of the passband. The filter use a single dual mode open loop resonator. The

    even mode frequency for filter B has to be lower than the odd mode frequency.

    Figure 10 shows the simulated performance of the filter. The measured center

    frequency of 0.95 GHz is slightly lower than the simulated 1.01 GHz and a normalized

    frequency axis is used for comparison. It can be seen that the measured frequency response

    is very similar to the simulated one. The filter exhibits a higher selectivity on the low side of

    the passband because of the finite frequency transmission zero. The first spurious does not

    occur at 2fo, but at a higher frequency.

    Figure 7: Layout of filter B. W=10.9 mm. because the even mode frequency in order to

    place a desired transmission zero on the low side of the passband.

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    Figure 8: Measured and simulated frequency response of filter B in Figure 7.

    Filterhigher order filter based on coupled dual mode open loop resonator.

    To prove that when two dual mode open loop resonator are coupled, there are only

    couplings between the same mode of resonator (odd mode resonator1=odd mode resonator

    2), (even mode resonator 1=even mode resonator 2)

    For the purpose, an inter resonator coupling structure is proposed in Figure below.

    Where two identical dual mode open loop resonators are approximately placed in opposite

    orientations with a spacing s to facilitate the coupling.

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    Figure 9: Coupled dual mode open loop resonator.

    Figure 10: resonant response of the coupled resonator structures

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    Four pole dual mode filter

    The coupling structure proposed in [6] for a multipole dual mode resonator filter is also

    applicable for the type of dual-mode resonator .In Figure 13, the filter has a very compact

    size and uses a resonator structure with a small offset of the two coupled resonator for a fine

    adjustment of the couplings. The simulated and measured performance of the filter are

    shown in Figure 14. The simulated performance is fo=?????.The filter shows a finite

    frequency zero on the high side of the passband, which is because the dual mode open loop

    resonator used have a higher even mode frequency.

    The above four pole filter exhibits a second order finite frequency transmission zero on

    one side of the passband for applications such as duplexers where asymmetric

    frequency selectivity is desired.

    Figure 11: Four pole filter (n=4) composed of two identical dual mode open loop resonators

    is developed. The filter has a very compact size and uses a coupled resonator structure in

    figure 11, with a small offset of the two coupled resonator for a fine adjustment of the

    couplings.

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    Figure 12: The filter shows a finite frequency zero on the high side of the passband, which is

    because the dual mode open loop resonators used have a higher even mode frequency.

    A four pole filter design to produce two transmission zeros, each on the one side of the

    passband, leading to a symmetric frequency response is required.

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    Figure 13: Layout of the for pole dual mode filter

    Figure 14: Symmetric passband with the two desired transmission zero.

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    As a matter of fact, a four pole filter can also be designed to produce two transmission zeros,

    each on the one side of the passband, leading to a symmetric frequency response if required. In

    this case, the two dual mode resonators without any direct couplings between them are cascadedthrough the two nonresonating nodes, as indicated. The two dual mode resonator are also not the

    same as shown, one will produce a zero on the high side of the passband, while the other will

    produce a zero on the low side of the passband. The result is plotted Figure 16, showing a

    symmetric passband with the two desired transmission zeros.

    CONCLUSION

    A miniature dual mode open loop resonator has been proposed for filter applications. The

    characteristics of this new type of microstrip dual mode resonator has been investigated. It has

    been shown that the loading element inside the open loop resonator does not affect the odd mode

    characteristic and the two modes of this type of dual mode resonator do not couple even with a

    mode split. It has also shown that a finite frequency transmission zero is inherently associated

    with the even mode. This makes it much easier to design bandpass filters with an asymmetric

    response, as no cross coupling is required to generate the transmission zero. The two pole filters

    of this type with opposite asymmetric responses have been demonstrated. To this end, a compact

    four pole filter using filter using two dual mode open loop resonators has been demonstrated

    with a promising performance. In addition, by implementing nonresonating nodes, it has been

    shown that the four pole filter can actually produce two separated transmission zeros, each on

    one side of the passband.

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    The expectation outcomes of this project are:

    1. Dual Mode Microstrip Open Loop Resonators can be used like a filters2. Varying of length gap between two resonators can be effect of the filter responses.

    REFFERENCE

    [1] Jia Sheng Hong, Hussein Shaman, Young Hoon Chun (2007), Dual Mode

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    [2] J.-S. Hong and M. J. Lancaster, Microstrip Filters for RF/Microwave

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    [3] I. Wolff, Microstrip bandpass filter using degenerate modes of a microstrip ring

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    [5] J. A. Curitis and S. J. Fiedziuszko, Miniature dual mode microstrip filters, in

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    [6] R. R. Mansour, Design of superconductive multiplexers using singlemode and

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    [8] C. Lugo and J. Papapolymerou, Bandpass filter design using a microstrip

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    [9] R. Wu and S. Amari, New triangular microstrip loop resonators for bandpass

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