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EE 447 VLSI Design. Lecture 8: Circuit Families. Outline. Pseudo-nMOS Logic Dynamic Logic Pass Transistor Logic. Introduction. What makes a circuit fast? I = C dV/dt -> t pd (C/I) D V low capacitance high current small swing Logical effort is proportional to C/I - PowerPoint PPT Presentation
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1EE 447 VLSI Design
EE 447 VLSI Design
Lecture 8: Circuit Families
2EE 447 VLSI Design
Outline
Pseudo-nMOS Logic Dynamic Logic Pass Transistor Logic
3EE 447 VLSI Design
Introduction
What makes a circuit fast? I = C dV/dt -> tpd (C/I) V low capacitance high current small swing
Logical effort is proportional to C/I pMOS are the enemy!
High capacitance for a given current Can we take the pMOS capacitance off the input? Various circuit families try to do this…
B
A
11
4
4
Y
4EE 447 VLSI Design
Pseudo-nMOS
In the old days, nMOS processes had no pMOS Instead, use pull-up transistor that is always ON
In CMOS, use a pMOS that is always ON Ratio issue Make pMOS about ¼ effective strength of pulldown
network
Vout
Vin
16/2
P/2
Ids
load
0 0.3 0.6 0.9 1.2 1.5 1.8
0
0.3
0.6
0.9
1.2
1.5
1.8
P = 24
P = 4
P = 14
Vin
Vout
5EE 447 VLSI Design
Pseudo-nMOS Gates
Design for unit current on output
to compare with unit inverter. pMOS fights nMOS
Inverter NAND2 NOR2
AY
B
AY
A B
gu =gd =gavg =pu =pd =pavg =
Y
gu =gd =gavg =pu =pd =pavg =
gu =gd =gavg =pu =pd =pavg =
finputs
Y
6EE 447 VLSI Design
Pseudo-nMOS Gates
Design for unit current on output
to compare with unit inverter. pMOS fights nMOS
Inverter NAND2 NOR2
4/3
2/3
AY
8/3
8/3
2/3
B
AY
A B 4/34/3
2/3
gu =gd =gavg =pu =pd =pavg =
Y
gu =gd =gavg =pu =pd =pavg =
gu =gd =gavg =pu =pd =pavg =
finputs
Y
7EE 447 VLSI Design
Pseudo-nMOS Gates
Design for unit current on output
to compare with unit inverter. pMOS fights nMOS
Inverter NAND2 NOR2
4/3
2/3
AY
8/3
8/3
2/3
B
AY
A B 4/34/3
2/3
gu = 4/3gd = 4/9gavg = 8/9pu =pd =pavg =
Y
gu = 8/3gd = 8/9gavg = 16/9pu =pd =pavg =
gu = 4/3gd = 4/9gavg = 8/9pu =pd =pavg =
finputs
Y
8EE 447 VLSI Design
Pseudo-nMOS Gates
Design for unit current on output
to compare with unit inverter. pMOS fights nMOS
Inverter NAND2 NOR2
4/3
2/3
AY
8/3
8/3
2/3
B
AY
A B 4/34/3
2/3
gu = 4/3gd = 4/9gavg = 8/9pu = 6/3pd = 6/9pavg = 12/9
Y
gu = 8/3gd = 8/9gavg = 16/9pu = 10/3pd = 10/9pavg = 20/9
gu = 4/3gd = 4/9gavg = 8/9pu = 10/3pd = 10/9pavg = 20/9
finputs
Y
9EE 447 VLSI Design
Pseudo-nMOS Design
Ex: Design a k-input AND gate using pseudo-nMOS. Estimate the delay driving a fanout of H
G = F = P = N = D =
In1
Ink
Y
Pseudo-nMOS1
1 H
10EE 447 VLSI Design
Pseudo-nMOS Design
Ex: Design a k-input AND gate using pseudo-nMOS. Estimate the delay driving a fanout of H
G = 1 * 8/9 = 8/9 F = GBH = 8H/9 P = 1 + (4+8k)/9 = (8k+13)/9 N = 2 D = NF1/N + P =
In1
Ink
Y
Pseudo-nMOS1
1 H
4 2 8 13
3 9
H k
11EE 447 VLSI Design
Pseudo-nMOS Power
Pseudo-nMOS draws power whenever Y = 0 Called static power P = I•VDD
A few mA / gate * 1M gates would be a problem This is why nMOS went extinct!
Use pseudo-nMOS sparingly for wide NORs Turn off pMOS when not in use
A BY
C
en
12EE 447 VLSI Design
Dynamic Logic
Dynamic gates uses a clocked pMOS pullup Two modes: precharge and evaluate
1
2A Y
4/3
2/3
AY
1
1
AY
Static Pseudo-nMOS Dynamic
Precharge Evaluate
Y
Precharge
13EE 447 VLSI Design
The Foot
What if pulldown network is ON during precharge? Use series evaluation transistor to prevent fight.
AY
foot
precharge transistor
Y
inputs
Y
inputs
footed unfooted
f f
14EE 447 VLSI Design
Logical Effort
Inverter NAND2 NOR2
1
1
AY
2
2
1
B
AY
A B 11
1
gd =pd =
gd =pd =
gd =pd =
Y
2
1
AY
3
3
1
B
AY
A B 22
1
gd =pd =
gd =pd =
gd =pd =
Y
footed
unfooted
32 2
15EE 447 VLSI Design
Logical Effort
Inverter NAND2 NOR2
1
1
AY
2
2
1
B
AY
A B 11
1
gd = 1/3pd = 2/3
gd = 2/3pd = 3/3
gd = 1/3pd = 3/3
Y
2
1
AY
3
3
1
B
AY
A B 22
1
gd = 2/3pd = 3/3
gd = 3/3pd = 4/3
gd = 2/3pd = 5/3
Y
footed
unfooted
32 2
16EE 447 VLSI Design
Monotonicity
Dynamic gates require monotonically rising inputs during evaluation 0 -> 0 0 -> 1 1 -> 1 But not 1 -> 0
Precharge Evaluate
Y
Precharge
A
Output should rise but does not
violates monotonicity during evaluation
A
17EE 447 VLSI Design
Monotonicity Woes
But dynamic gates produce monotonically falling outputs during evaluation
Illegal for one dynamic gate to drive another!
AX
Y
Precharge Evaluate
X
Precharge
A = 1
Y
18EE 447 VLSI Design
Monotonicity Woes
But dynamic gates produce monotonically falling outputs during evaluation
Illegal for one dynamic gate to drive another!
AX
Y
Precharge Evaluate
X
Precharge
A = 1
Y should rise but cannot
Y
X monotonically falls during evaluation
19EE 447 VLSI Design
Domino Gates
Follow dynamic stage with inverting static gate Dynamic / static pair is called domino gate Produces monotonic outputs
Precharge Evaluate
W
Precharge
X
Y
Z
A
BC
C
AB
W XY Z =
XZH H
A
W
B C
X Y Z
domino AND
dynamicNAND
staticinverter
20EE 447 VLSI Design
Domino Optimizations
Each domino gate triggers next one, like a string of dominos toppling over
Gates evaluate sequentially but precharge in parallel Thus evaluation is more critical than precharge HI-skewed static stages can perform logic
S0
D0
S1
D1
S2
D2
S3
D3
S4
D4
S5
D5
S6
D6
S7
D7
YH
21EE 447 VLSI Design
Dual-Rail Domino
Domino only performs noninverting functions: AND, OR but not NAND, NOR, or XOR
Dual-rail domino solves this problem Takes true and complementary inputs Produces true and complementary outputs
sig_h sig_l Meaning
0 0 Precharged
0 1 ‘0’
1 0 ‘1’
1 1 invalid
Y_h
f
inputs
Y_l
f
22EE 447 VLSI Design
Example: AND/NAND
Given A_h, A_l, B_h, B_l Compute Y_h = A * B, Y_l = ~(A * B)
23EE 447 VLSI Design
Example: AND/NAND
Given A_h, A_l, B_h, B_l Compute Y_h = A * B, Y_l = ~(A * B) Pulldown networks are conduction complements
Y_h
Y_l
A_h
B_hB_lA_l
= A*B= A*B
24EE 447 VLSI Design
Example: XOR/XNOR
Sometimes possible to share transistors
Y_h
Y_l
A_l
B_h
= A xor B
B_l
A_hA_lA_h= A xnor B
25EE 447 VLSI Design
Leakage
Dynamic node floats high during evaluation Transistors are leaky (IOFF 0) Dynamic value will leak away over time Formerly miliseconds, now nanoseconds!
Use keeper to hold dynamic node Must be weak enough not to fight evaluation
A
H
2
2
1 kX
Y
weak keeper
26EE 447 VLSI Design
Charge Sharing
Dynamic gates suffer from charge sharing
B = 0
AY
x
Cx
CY
A
x
Y
27EE 447 VLSI Design
Charge Sharing
Dynamic gates suffer from charge sharing
B = 0
AY
x
Cx
CY
A
x
Y
Charge sharing noise
x YV V
28EE 447 VLSI Design
Charge Sharing
Dynamic gates suffer from charge sharing
B = 0
AY
x
Cx
CY
A
x
Y
Charge sharing noise
Yx Y DD
x Y
CV V V
C C
29EE 447 VLSI Design
Secondary Precharge
Solution: add secondary precharge transistors Typically need to precharge every other node
Big load capacitance CY helps as well
B
AY
x
secondaryprechargetransistor
30EE 447 VLSI Design
Noise Sensitivity
Dynamic gates are very sensitive to noise Inputs: VIH Vtn
Outputs: floating output susceptible noise Noise sources
Capacitive crosstalk Charge sharing Power supply noise Feedthrough noise And more!
31EE 447 VLSI Design
Domino Summary
Domino logic is attractive for high-speed circuits 1.5 – 2x faster than static CMOS But many challenges:
Monotonicity Leakage Charge sharing Noise
Widely used in high-performance microprocessors
32EE 447 VLSI Design
Pass Transistor Circuits
Use pass transistors like switches to do logic Inputs drive diffusion terminals as well as gates
CMOS + Transmission Gates: 2-input multiplexer Gates should be restoring
A
B
S
S
S
Y
A
B
S
S
S
Y
33EE 447 VLSI Design
LEAP
LEAn integration with Pass transistors Get rid of pMOS transistors
Use weak pMOS feedback to pull fully high Ratio constraint
B
S
S
AYL
34EE 447 VLSI Design
CPL
Complementary Pass-transistor Logic Dual-rail form of pass transistor logic Avoids need for ratioed feedback Optional cross-coupling for rail-to-rail swing
B
S
S
S
S
A
B
AY
YL
L