128

Chapter 7

DESIGN MODIFICATION OF

INDUCTION MOTOR FOR PROPULSION

PURPOSES

7.1. Introduction

Electric drives are well known and wide spread in the industry and

propulsion system. However, the constraints and the requirements on these

kinds of systems embedded in propulsion applications are different, when

compared to a classical industrial application. Generally, transport

propulsion systems are operating during relatively short duration, with a lot

of transient phases and on a wide range of operating conditions. Even

though permanent magnet machines are getting more widespread in drive

applications due to their superior power density, compactness and current

availability of power electronics needed for effective control, recent increase

in price of permanent magnet materials restricts its wide spread usage in

propulsion system. Switched reluctance machines have also been

considered as a candidate for propulsion application, but they are still less

widespread. Synchronous wound rotor machine is also not selected due to

rotor copper losses which is difficult to evacuate. Induction motors are

considered a better choice for propulsion applications due to their

robustness, reliability, low price etc.

In this chapter, adaptability of induction motor for propulsion system

is discussed with emphasis given to motor-load interaction. Design options

are generated with the spread sheets developed for design of induction

motor. Simulations are carried out in various design options for motor

characteristics and compared the results.

129

7.2. Motor Characteristics for Propulsion Systems

The major requirements of electric propulsion are listed as below:

i. High instant power and high power density for better

controllability.

ii. High torque at low speeds for starting and climbing, as well as

constant power at high speed for cruising.

iii. Four quadrant operation with parabolic torque speed curve

iv. Very wide speed range including constant torque and constant

power regions.

v. Fast torque response.

vi. High efficiency over the entire speed range including regeneration.

vii. High reliability and robustness with sensorless speed control.

viii. Techno economically viable cost along with controlled drives.

Moreover, in the event of faulty operation, the electric propulsion

should be fault-tolerant.

7.3. Induction Motor for Propulsion Systems

The selection of the electrical motor is an important task that requires

special characteristics like high torque response and efficiency, or low

maintenance and cost [88]. Nowadays, three–phase squirrel cage induction

motors and permanent magnet machines are more appropriate solutions

due to their lower cost and higher reliability [89].The regenerative braking

system, which allows delivering power back to the batteries while braking, or

even when vehicles go downhill, is also possible when using induction

motors along with controlled drives.

Development of precise digital algorithms using microcontrollers to

control power inverters for driving induction motors made them an ideal

candidate in propulsion applications. Cage induction motors are widely

accepted as the most potential candidate for the propulsion according to

their reliability, ruggedness, low maintenance, low cost, and ability to

operate in hostile environments [90], [91]. Field oriented control of induction

motors can decouple its torque control from field control and extended speed

range operation with constant power beyond base speed is accomplished by

130

flux weakening. The torque-speed characteristics of induction motor with

extended field weakening operation can meet nearly series excited DC motor

characteristics is shown in Fig. 7.1 for comparison.

Fig. 7.1 Torque-speed curves

7.4. Design Modification of Induction Motor to meet propulsion characteristics along with suitable drive

Existing constraints for the operation at field weakening are the

maximum output voltage and the permitted maximum current of the

inverter. To produce the maximum torque that the machine could possibly

develop, the excitation level at field weakening must be appropriately

adjusted. Most commonly used methods of excitation control do not fully

utilize the installed inverter power which can lead to a reduction of torque

and power down to 65% in field weakening [92]. Hence it is required to

explore novel methods to produce maximum torque of the induction motor

in FW region. The innovative method proposed in this work is to carry out

appropriate design modification of the induction motor by changing the

rated value of power factor so as to produce maximum torque in field

weakening region.

131

7.4.1. Design Procedures for Induction Motors

The main specifications for the design of a three phase squirrel cage

induction motor are: rated output power in HP or kW, frequency in Hz,

voltage in volts, speed in rpm, efficiency, power factor and full load current

in ampere.

The standard specification for the design of an induction motor are

materials (lamination thickness, conductor diameter), performance indexes

(efficiency, power factor, starting torque, starting current, breakdown

torque), temperature by insulation class, frame sizes, shaft height, cooling

types, service classes, protection classes, etc. are specified in national (or

international) standards (NEMA, IEEE, IEC, EU, etc.) to facilitate the

induction motor for various applications.

The main purpose of designing an induction motor is to obtain the

complete physical dimensions of all the parts of the machine as mentioned

below to satisfy the given specifications and stator and rotor laminations of

an induction motor are shown in Fig. 7.2.

i. The main dimensions of the stator

ii. Details of stator windings

iii. Design details of rotor and its windings

iv. Performance characteristics

Fig. 7.2 Stator and rotor laminations of an induction motor

132

The stator and rotor dimensions are determined by independent

variables which are: stator slot height, stator tooth width, rotor slot height,

rotor tooth width, air-gap length, air gap flux density, stack length, outer

stator diameter, stator wire size and electrical steel type. Besides the above

independent variables, the design involves some non-linear constraints

which concern mainly the motor performances. They are stator winding

temperature, rotor bar temperature, flux density in the stator.

The type of winding adopted in this design is the squirrel cage type as

it is simpler and economical, which consists of a number of bars embedded

in the rotor slots and connected at both ends by means of end rings.

The major steps in designing an IM may be divided into 5 areas:

electrical, dielectric, magnetic, thermal and mechanical.

Electrical design: To supply the IM, the supply voltage, frequency, and

number of phases are specified. From this data and the minimum power

factor and a target efficiency, the phase connection (star or delta), winding

type, number of poles, slot numbers and winding factors are calculated.

Current densities are imposed.

Magnetic design: Based on output coefficients, power, speed, number of

poles, type of cooling, and the rotor diameter is calculated. Then, based on a

specific current loading (in A/m) and air gap flux density, the stack length is

determined. Fixing the flux densities in various parts of the magnetic circuit

with given current densities and slot mmfs, the slot sizing, core height, and

external stator diameter Do are all calculated. Choosing Do, which is

standardized, the stack length is modified until the initial current density in

the slot is secured. It is evident that sizing the stator and rotor core may be

done many ways based on various criteria.

Insulation design: Insulation material and its thickness, be it slot/core

insulation, conductor insulation, end connection insulation, or terminal

leads insulation depends on machine voltage insulation class and the

environment in which the motor operates. There are low voltage 400V/50Hz,

230V/60Hz, 460V/60Hz 690V/60Hz or less or high voltage machines

(2.3kV/60Hz, 4kV/50Hz, 6kV/50Hz). When PWM converter fed IMs are

used, care must be exercised in reducing the voltage stress on the first 20%

of phase coils or to enforce their insulation or to use random wound coils.

133

Thermal design: Extracting the heat caused by losses from the IM is

imperative to keep the windings, core, and frame temperatures within safe

limits. Depending on application or power level, various types of cooling are

used. Air cooling is predominant but stator water cooling in the stator of

high speed induction motors (above 10,000 rpm) is frequently used.

Calculating the loss and temperature distribution and the cooling system

represents the thermal design.

Mechanical design: Mechanical design refers to critical rotating speed, noise,

and vibration modes, mechanical stress in the shaft, and its deformation

displacement, bearings design, inertia calculation, and forces on the winding

end coils during most severe current transients.

Output Equation:

Output equation is the mathematical expression which relates the

output of the machine with its main dimensions. Rating of a three phase

induction motor, Q in kVA is given by:

33 10ph phQ E I−=

(7.1)

33 4.44 10w ph phQ x K f T Iφ −= (7.2)

where

2

spn

f =

Since the rating is given in kW, converted to kVA by the relation:

c o s

k Wk V A

η φ=

(7.3)

Specific magnetic loading, Bav is obtained by dividing total flux around the

air gap by area of flux path in the air gap as:

av

pB

DL

φ

π=

(7.4)

Specific electric loading, ac is obtained by dividing total armature ampere

conductors by armature periphery at the air gap.

6 ph phzT II Z

acD Dπ π

= =

(7.5)

Rearranging (7.4) and (7.5) gives:

134

a vB D L

p

πφ =

(7.6)

6p h p h

ac DT I

π=

(7.7)

Substituting (7.6) and (7.7) in (7.1) gives output equation of an AC machine,

2

0 sQ C D L n=

(7.8)

where output coefficient, 2 3

01 .1 1 1 0

a v wC B a c Kπ −=

31 1 1 0a v w

B a c K −=

Step 1: Choice of Specific loadings

Specific Magnetic loading or Air gap flux density, Bav is selected

normally from standard design tables and the value is limited by losses in

the teeth and magnetizing current. Higher value of Bav means the winding

requires less number of turns per phase, but higher value of overload

capacity. For 50 Hz machine, the value of Bav is between 0.35 and

0.60Wb/m2.

Specific electric loading or ampere conductors per meter of air gap

circumference, ac is selected from traditional design tables. High value of ac

means less electric material, but higher electric losses and lower overload

capacity. Normal range of ac is between10000 ac/m and 450000 ac/m.

Bav and ac values are taken from table A2.1 in Appendix II.

Step 2: Calculation of the main dimensions

The armature diameter and stator core length are known as the main

dimensions.

Aspect ratio: The output equation, (7.8) gives the relation between D2L

product and output of the machine,

2

0 s

QD L

C n=

(7.9)

The operating characteristics of an induction motor are mainly

influenced by the ratio L/τ. The ratio, D/L determines the shape of a pole,

135

square or rectangular. λ is selected from Fig. A2.1 and L and D are

calculated using (7.9) and (7.10).

L L

D

p

λτ π

= =

(7.10)

Following are the various design considerations based on which a

suitable value for λ can also be assumed from table 7.1.

Table 7.1 Value of λ

Sl No.

Design consideration Value of λ

1 Minimum overall cost 1.5 to 2.0

2 Good efficiency 1.4 to 1.6

3 Good overall design 1.0 to 1.1

4 Good power factor 1.0 to 1.3

For obtaining the best power factor, the following relation is usually

assumed for calculating D and L.

0.18Lτ = (7.11)

Check for peripheral velocity: The peripheral velocity is calculated using

(7.12). For normal construction of induction motors, the calculated diameter

of the motor should be such that the peripheral velocity must be below 30

m/sec.

v Dnπ= (7.12)

Step 3: Assume efficiency and power factor

Power factor and efficiency under full load conditions will increase

with increase in rating of the machine. The power factor and efficiency will

be higher for a high speed machine than the same rated low speed machine

because of better cooling conditions. Taking into considerations all these

factors the above parameters will vary in a range based on the output of the

machine. Efficiency and power factor values are selected from Figs. A2. 2

and A2. 3 in Appendix-II.

136

Step 4: Calculation of air gap length from empirical formula

The air gap length depends on both electromagnetic factors such as

magnetizing current, pulsation losses; and mechanical factors such as

mechanical tolerances, bearing, shaft deflection, unbalanced magnetic pull.

The length of airgap primarily determines the magnetizing current drawn by

the machine. The air gap length is calculated from [93], [94] as,

1/2

35 10

2g

Dl x τ−

=

(7.13)

( )1/233 10

gl x pτ−=

(7.14)

0.2 2gl DL= +

(7.15)

where

D

p

πτ =

Step 5: Calculation of rotor diameter

Rotor diameter can be calculated from the length of air gap as:

Rotor diameter, Dr = Stator bore – 2 x length of airgap 2g

D l= − (7.16)

Step 6: Calculation of number of stator turns per phase

Generally the induced emf can be assumed to be equal to the applied

voltage per phase. Thus stator turns per phase is obtained from emf

equation as:

4.44

ph

ph

w

ET

f Kφ=

(7.17)

where Flux/pole, a veB D L

p

πφ =

and winding factor, Kw is assumed as 0.955 for full pitch distributed

winding.

Step 7: Calculation of number of stator conductors

Number of conductors per phase and the total number of stator

conductors are obtained using the expressions (7.18) and (7.19)

2ph ph

Z T= (7.18)

137

6s ph

S T= (7.19)

Step 8: Calculation of cross sectional area of stator conductors

Stator current per phase is calculated using (7.20) as,

3 coss

ph

QI

V φ=

(7.20)

Area of cross section of stator conductors can be estimated from the

stator current per phase and suitably assumed value of current density for

the stator windings. Cross sectional area of the stator conductor,

ss

s

Ia

δ=

(7.21)

where δs is the current density in stator windings.

Usual value of current density for stator windings is 3 to 5 A/mm2.

Based on the cross sectional area, shape and size of the conductor can be

decided. If the sectional area of the conductors is below 5 mm2 then usually

circular conductors are employed. If it is above 5 mm2 then rectangular

conductors will be selected. Standard bare size of round and rectangular

conductors is obtained from the standard tables. In case of rectangular

conductors width to thickness ratio must be between 2.5 to 3.5.

Step 9: Calculation of stator slot pitch

Number of stator slots shown in Fig. 7.3 is selected based on the

criteria that less number of slots provides less cost and less space lost due

to insulation and slot opening while more number of slots causes smaller

leakage inductance, larger breakdown torque, small MMF harmonics and

better cooling. The total conductor per stator slot is calculated using (7.22)

as:

6p h

s

s

TZ

S=

(7.22)

where, Zs is an integer for single layer winding and even number for double

layer winding. Stator slot pitch at the air gap surface is obtained by dividing

gap surface by total number of stator slots as,

138

ss

s

DY

S

π=

(7.23)

The slot pitch at the air gap surface for open type of slots should be between

15mm to 25mm.

Fig. 7.3 Stator slots

Step 10: Calculation of area of stator slot

After calculating the number of conductors per slot, approximate area

of the slot is estimated as copper area in the slot divided by slot space

factor.

Area of each slot =

s sz a

sp a ce fa c to r (7.24)

The space factor varies between 0.25 and 0.4. The ratio of slot depth to slot

width is assumed as 3 to 6.

Step 11: Check for specific electric loading

The Specific electric loading is recalculated using (7.25).

6 p h p hT I

a cDπ

=

(7.25)

Then initial value chosen for ac is checked with the recalculated value.

Steps (1) to (11) is repeated with different value for ac until both the values

are approximately equal.

Step 12: Calculation of stator teeth dimensions

The dimensions of the slot determine the value of flux density in the

teeth. Stator tooth area is selected depending on mechanical strength

139

without teeth flux density being too high (normally between 1.6 to 1.8

Wb/m2).

Minimum tooth area per pole = 1 .7

mφ

(7.26)

Minimum width of stator tooth = ( )

/s i

to o th a rea p er p o le

S p L (7.27)

Step 13: Select suitable values of flux density in stator

Select suitable values of flux density in stator back iron from table

A2.2 and compute stator outer diameter.

Step 14: Calculation of outer diameter stator laminations

Area of stator core can be obtained by dividing flux through the core,

which is half of the flux per pole, by flux density in stator core. Suitable

values of flux density (between 1.2 to 1.4Wb/m2) in stator back iron are

selected and stator outer diameter is computed using (7.28).

Area of stator core, 2

mi cs

cs

L dB

φ= (7.28)

Depth of stator core, 2

mcs

cs i

dB L

φ= (7.29)

Outside diameter of stator laminations = 2 ( )cs ssD d d+ + (7.30)

Step 15: Calculation of stator winding resistance

Resistance of the stator winding per phase is calculated using the

expression,

c mt ph

s

s

L TR

a

ρ=

(7.31)

where Resistivity of copper, ρc = 0.021Ωm

Length of mean turn, 2 2.3 0.24mtL L τ= + +

Using the calculated resistance of stator winding, copper losses in stator

winding can be calculated as:

Total copper losses in stator winding 23s s

I R=

(7.32)

140

Step 16: Calculation of number of rotor slots

Proper number of rotor slots is selected in relation to number of stator

slots otherwise undesirable effects will be found at the starting of the motor.

The number of rotor slots is 15 to 30 percent larger or smaller than the

number of stator slots. The number of rotor slots is selected using the

following guide lines.

(i) To avoid cogging and crawling: (a) Ss ≠ Sr, (b) Ss- Sr ≠ ±3p

(ii) To avoid synchronous hooks and cusps in torque speed

characteristics Ss - Sr ≠ ±p, ±2p, ±5p

(iii) To avoid noisy operation Ss - Sr ≠ ±1, ±2, (±p ±1), (±p±2)

The value of stator slot number and rotor slot number for different pole is

shown in table A2.3.

Step 17: Calculation of area of rotor bars

The cross section of the bars and the end rings must be so selected

that a proper value of rotor resistance is obtained, which meets both the

requirements of starting torque as well as efficiency. Bar current in the rotor

of a squirrel cage induction motor may be determined by comparing the mmf

developed in rotor and stator. Hence the current per rotor bar is given by,

2coss ws s

b ph

r

m K TI I

Sφ=

(7.33)

Area of each bar is obtained by,

bb

b

Ia

δ=

(7.34)

where δb is the current density in rotor bars which normally varies between

4 to 7 A/mm2. For aluminum bar it is between 2.2 and 4.5 A/mm2 [95]. For

deep bar rotor it is between 5.5 and 7.5 A/mm2. For load with large inertia

and high rated speed, it should not exceed 6.5 to 7 A/mm2. Once the cross

sectional area is known the size of the conductor is selected form standard

design table.

Step 18: Select rotor slot shape and size

Semi-closed slots or closed slots with very small or narrow openings

are selected for the rotor slots shown in Fig. 7.4, as these are giving better

performance to the motor in the following ways:

141

i. As the rotor is closed the rotor surface is smooth at the air gap and

hence the motor draws lower magnetizing current,

ii. Reduced noise as the air gap characteristics are better,

iii. Increased leakage reactance,

iv. Reduced starting current,

v. Over load capacity is reduced and

vi. Undesirable and complex air gap characteristics.

Fig. 7.4 Types of rotor slots

Step 19: Calculation of copper losses in rotor bar

Knowing the length of the rotor bars and resistance of the rotor bars

copper losses in the rotor bars is calculated as,

Copper loss in rotor bars = 2

r b bS I r

(7.35)

where rotor bar resistance, 0 .0 2 1

bb

b

Lr

A=

Length of rotor bar, Lb = L + allowance for skewing

Step 20: Calculation of end ring current

The maximum end ring current is taken as the sum of the average

current in half of the number of bars under one pole. Current is not the

maximum in all the bars under one pole at the same time but varies

sinusoidally.

1

2 1.11

brbav

ISI

p=

(7.36)

rms value of end ring current,

142

1

1.11

brrms

ISI

pπ=

(7.37)

Step 21: Calculation of area of end rings

A suitable value for the current density in the end rings is assumed

and cross sectional area for the end ring is calculated as:

ee

e

IA

δ=

(7.38)

where δe is the current density in rotor bars which normally varies between

4.5 and 7.5 A/mm2. Area of end ring is obtained from another expression as

the product of depth of end ring and thickness of end ring.

Area of ring = depth of end ring x thickness of end ring =e ed t

(7.39)

Step 22: Calculation of copper losses in end rings

Knowing the length of current path and resistance of end ring, copper

losses in the rotor bars is calculated as,

Copper loss in end ring = 22e e

I r

(7.40)

where end ring resistance, 0 .0 2 1

m ee

e

Lr

A=

Mean length of the current path in end ring, m e m eL Dπ=

Mean diameter of end ring should be between 4 to 6 cm less than rotor

diameter.

Step 23: Calculation of Magnetizing current

Magnetizing MMF is obtained as.

0

2( )g

m c g mts mtr mcs mcr

BF K l F F F F

µ= + + + +

(7.41)

where, Kc - the Carter coefficient to account for the effective air gap

length increase due to slot opening which is usually range

from1.0 to 1.5 [93]-[96].

Fmts - MMF drop along stator teeth

Fmtr - MMF drop along rotor teeth

143

Fmcs - MMF drop along stator core

Fmcr - MMF drop along rotor core

MMF drop along stator teeth, stator core, rotor teeth and rotor core

are estimated from assigned flux density and B- H curve.

Based on the total ampere turns of the magnetic circuit, the magnetizing

current is calculated as [96],

0.427 mm

ph w

pFI

T K=

(7.42)

Step 24: Calculation of no load losses

Iron loss has two components, hysteresis and eddy current losses,

occurring in the iron parts depend upon the frequency of the applied

voltage. The frequency of the induced voltage in rotor is equal to the slip

frequency which is very low and hence the iron losses occurring in the rotor

is negligibly small. Hence the iron losses occurring in the induction motor is

mainly due to the losses in the stator alone. The total iron loss in induction

motor is taken as the sum of iron loss in stator core and iron losses in stator

teeth. The iron loss in stator teeth and core is obtained by calculating their

respective weights.

In addition to iron losses, friction and windage loss is to be taken into

account by assuming it as 1.0 to 2.0 % of the output of the motor.

total no load losses = Total iron losses + Friction and windage loss (7.43)

Step 25: Calculation of no load current

The no load current of an induction motor has two components,

magnetizing component, Im and iron loss component, Iw and is calculated as,

2 2

0 mI wI I= +

(7.44)

where, Iw = Total no load loss/(3 x phase voltage)

Step 26: Calculation of no load Power Factor

No load power factor is calculated knowing the components of no load

current.

144

0

0

cos wI

Iφ =

(7.45)

Step 27: Calculation of stator leakage reactance

The leakage reactance is calculated by considering several

components by using some equations and some empirical formulas.

2

02 ( )

ph

sl sls ds ecs

TX fL

pqπµ λ λ λ= + +

(7.46)

where q – Stator slots/pole/phase

slsλ -Stator slot leakage coefficients

dsλ -Stator differential leakage coefficients

ecsλ -Stator end leakage coefficients

2

1

0 12 ( ) ( )

ph

sl sls ds ecs s sls ds ecs sls ds ecs

TX f L C X X X

pqπµ λ λ λ λ λ λ= + + = + + = + +

(7.47)

where slsX -Stator slot leakage reactance

dsX -Stator differential leakage reactance

ecsX -Stator end leakage reactance

Slot leakage coefficients: Deeper slot indicates larger slot leakage reactance

and wider slot means smaller leakage reactance. Figs. 7.5 and 7.6 show the

slot leakage flux in slots and slot dimensions.

Fig. 7.5 Slot leakage flux in slots

145

Fig. 7.6 Slot dimensions

1 2 1

22 1 3[ ]( )3 ( ) ( ) 4

s w ossls

s s os s os

h h h

b b b b b

βλ

+= + +

+ +

(7.48)

Differential leakage coefficients: Fig 7.7 shows the zig-zag stator and rotor

leakage flux lines.

Fig. 7.7 Zig-zag leakage flux lines

5 / 3 1

5 4 / 4

g c o

ds

g c o

l K b

l K b

βλ

+=

+

(7.49)

End leakage coefficients: Fig 7.8 shows the coil dimensions.

146

Fig. 7.8 Coil dimensions

0.34 ( 0.64 )ecs end p

ql

Lλ βτ= −

(7.50)

7.4.2. Development of Design Sheet

Based on the design procedure, design spread sheets are developed for

squirrel cage induction motor. The rated values are entered in input cells

and designed values are obtained from output cells of the design sheet as

shown in Fig. A2. 4 in Appendix II.

7.4.3. Design Modifications and Design Options

The induction motor is singly excited, means the electrical power is

applied only to the stator winding and the current flows through the rotor

winding by induction. As a consequence both the magnetizing current,

which sets up the magnetizing field, and the power current which allows

energy to be delivered to the shaft load, flow through the stator winding.

Thus for obtaining a larger power component for a given rating, either

keeping the magnetizing current as small as possible or keeping the air gap

as small as possible. For reduced value of magnetizing current, the power

factor can be reduced as per design equations.

Design options are generated by reducing the specified value of power

factor and the options are named as Design-A, Design-B, Design-C and

Design-D for power factors equal to 0.9, 0.8, 0.7 and 0.6 respectively. The

design values obtained from the design sheet for the four design options are

tabulated and presented in table. 7.2.

147

Table 7.2 Values of parameters for design options

Parameters

Design options for 10kW Design options for 30kW

A. Rating

A B C D A B C D unit

Full load output P 10.0 10.0 10.0 10.0 30.0 30.0 30.0 30.0 kW

Line voltage V 220 220 220 220 220 220 220 220 Volts

Frequency f 50 50 50 50 50 50 50 50 Hz

Efficiency η 90 90 90 90 90 90 90 90

Synchronous speed Ns 1500 1500 1500 1500 1500 1500 1500 1500 rpm

Number of poles p 4 4 4 4 4 4 4 4

kVA input

18.5 15.9 13.9 12.3 55.56 47.62 41.67 37.04 kVA

Full load line current IL 48.6 41.6 36.4 32.4 145.7 124.9 109.3 97.1 A

Power factor pf 0.60 0.70 0.80 0.90 0.60 0.70 0.80 0.90

B. Specific Loading

Magnetic loading Bav 0.44 0.44 0.44 0.44 0.45 0.45 0.45 0.45 Wb/m2

Electic loading ac 22000 22000 22000 22000 23000 23000 23000 23000 A/m

Output coefficient C0 102.5 102.5 102.5 102.5 108.9 108.9 108.9 108.9

C. Main dimensions

Stator bore D 210 200 190 185 295 280 270 260 mm

Core length L 165 155 150 145 230 220 210 205 mm

Pole pitch t 165 157 149 145 232 220 212 204 mm

D. Stator

Flux/pole ϕ 0.012 0.011 0.01 0.009 0.024 0.022 0.02 0.019 Wb

Turns/phase Ts 86 96 106 112 44 48 52 56 Nos.

Slot pitch Yss 18.3 17.4 16.6 16.1 25.73 24.42 23.55 22.680 mm

Required bare dia. of conductor

2.11 1.96 1.83 1.73 3.27 3.03 2.84 2.67 mm

Provided bare dia. of conductor

2.12 2.06 1.90 1.80 3.25 3.25 3.25 3.25 mm

Current density δs 3.98 3.61 3.71 3.68 5.08 4.35 3.81 3.38 A/mm2

Length of mean turn Lmts 0.95 0.91 0.88 0.86 1.233 1.186 1.147 1.119 mm

Phase resistance at 75⁰C Rs 0.24 0.28 0.35 0.40 0.069 0.072 0.076 0.079 W

Copper loss at full load

574 478 461 419 1460 1125 903 750 watts

Depth of stator core dcs 37 35 33 33 51 49 47 45 mm

Depth of stator slot dss 23 22 25 24 17.00 20.54 20.54 20.54 mm

Outer diameter of stator D0 329 319 306 299 431 419 405 391 mm

148

Table 7.2 (contd.)

Parameters

Design options for 10kW Design options for 30kW

E. Rotor:

A B C D A B C D unit

Length of air gap

0.57 0.55 0.54 0.53 0.72 0.70 0.68 0.66 mm

Diameter of Rotor

209 199 189 184 294 279 269 259 mm

Slot pitch

17.3 16.4 15.6 15.2 24.3 24.4 23.6 22.7 mm

Rotor bar current

218 244 269 284 335 366 396 426 A

Required c/s area of rotor bar

44 49 54 57 67 73 79 85 mm2

Provided c/s area of rotor bar

45 54 54 57 70 77 84 88 mm2

Resistance of each bar

100.3 79.7 77.8 68.3 84.0 73.6 65.0 60.9 W

Copper loss in bars

182 180 214 210 358 374 387 421 watts

End ring current

661 737 814 860 1014 1106 1198 1290 A

Required c/s area of rotor ring

132 147 163 172 203 221 240 258 mm2

Provided c/s area of rotor ring

135 150 170 176 204 228 247 260 mm2

Copper loss in end rings

72.4 76.4 76.1 78.7 166.5 163.9 168.3 177.9 watts

Total copper loss

254.1 256.3 290.1 288.3 524.9 537.8 555.7 598.5 watts

Full load slip

0.02 0.02 0.03 0.03 0.02 0.02 0.02 0.02

Resistance of rotor refers to stator

0.3 0.3 0.34 0.34 0.07 0.07 0.07 0.08 W

Depth of rotor core

36 34 33 32 51 48 46 45 mm

F. No Load Current:

Magnetizing MMF/pole

1050 865 662 591 967 1058 1051 805 A

Magnetizing current Im 41.5 33.7 28.9 24.3 66.9 64.5 63.2 56.8 A

Magnetizing reactance Xm 10.10 13.70 19.70 23.3 5.60 5.58 6.09 8.56 W

Core loss (CL)

573 509 454 415 1288 1220 1086 988 watts

Friction&Windage loss(FWL)

100 100 100 100 300 300 300 300 watts

No load loss

673 609 554 515 1588 1520 1386 1288 watts

Loss component

1.02 0.92 0.84 0.78 2.41 2.30 2.10 1.95 A

No load current (NLC)

41.5 33.7 28.9 24.3 66.9 64.5 63.2 56.8 A

G. Short circuit current

Slot leakage reactance

0.40 0.46 0.56 0.62 0.11 0.13 0.15 0.16 W

Overhang leakage reactance

0.46 0.55 0.63 0.69 0.17 0.19 0.22 0.24 W

Zigzag leakage reactance

0.20 0.27 0.38 0.45 0.11 0.11 0.12 0.17 W

Total leakage reactance

1.05 1.27 1.58 1.76 0.39 0.43 0.49 0.57 W

H. Performance at full load

Total losses

1501 1344 1305 1223 3573 3183 2846 2636 watts

Input

11501 11344 11305 11223 33573 33183 32846 32636 watts

Efficiency

87.0 88.2 88.5 89.1 89.4 90.4 91.3 91.9 %

7.5. Simulation Results and Discussion

The simulation model for sensorless FOC induction motor using SVM

inverter for FW operation is developed based on Fig. 6.11 in chapter 6 which

is used for the simulation of various design options for 10 kW and 30 kW.

For comparing the performance of various design options Design-A, Design-

149

B, Design-C and Design-D for 10kW and 30kW are named as Case-1, Case-

2 respectively.-

In the simulation, the motor starts from standstill state in no load

condition for all the cases and the speed responses are shown in Figs. 7.9

and 7.10 respectively. From the results, it is clear that in all the cases the

rotor speed increases from Design-A to Design-D, which means the rotor

speed increases with the reduction in power factor. The maximum attainable

rotor speed is increased from 2.94 p.u. to 3.36 p.u. and 4.80 p.u. to 5.30

p.u. for Cases 1 and 2 respectively as given in table 7.3.Variation of torque

with respect to rotor speed is presented in Figs. 7.11 and 7.12 for Cases 1

and 2 respectively for various design options. Torque capability is improved

in the FW region when power factor decreases as demonstrated in table 7.4.

Table 7.3 Rotor speed attained with time

Time

(sec) Unit

Case-1 (10 kW) Case-2 (30 kW)

A B C D A B C D

2 rpm 2399 2473 2545 2672 4267 4341 4658 5089

p.u. 1.65 1.70 1.75 1.84 2.94 2.99 3.21 3.50

4 rpm 3308 3417 3520 3731 5817 5934 6284 6571

p.u. 2.28 2.35 2.42 2.57 4.01 4.09 4.33 4.53

6 rpm 3864 4005 4106 4374 6607 6798 6880 7221

p.u. 2.66 2.76 2.83 3.01 4.55 4.68 4.74 4.98

8 rpm 4270 4430 4525 4885 6973 7225 7335 7687

p.u. 2.94 3.05 3.12 3.36 4.80 4.98 5.05 5.30

Table 7.4 Maximum torque (p.u.)

Rotor

Speed

(p.u.)

Case-1 (10 kW) Case-2 (30 kW)

A B C D A B C D

1.0 0.611 0.728 0.836 0.865 0.705 0.870 0.886 0.976

2.0 0.309 0.331 0.350 0.395 0.292 0.353 0.405 0.464

3.0 0.201 0.217 0.234 0.268 0.196 0.229 0.271 0.307

5.0 - - - - 0.093 0.101 0.104 0.166

150

Fig. 7.9 Speed responses for Case-1 (10 kW)

Fig. 7.10 Speed responses for Case-2 (30 kW)

0

1000

2000

3000

4000

5000

6000

0 1 2 3 4 5 6 7 8

Design - A

Design - B

Design - C

Design - D

Ro

tor

spee

d(r

pm

)

Time(sec)

0

1000

2000

3000

4000

5000

6000

7000

8000

0 1 2 3 4 5 6 7 8

Design - A

Design - B

Design - C

Design - D

Ro

tor

spee

d(r

pm

)

Time(sec)

151

Fig. 7.11 Torque vs. Rotor speed for Case-1 (10 kW)

Fig. 7.12 Torque vs. Rotor speed for Case-2 (30 kW)

0.0

0.2

0.4

0.6

0.8

1.0

1.2

0.0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0

Design - A

Design - B

Design - C

Design - D

To

rqu

e (p

.u.)

Rotor speed (p.u.)

0.0

0.2

0.4

0.6

0.8

1.0

1.2

0.0 1.0 2.0 3.0 4.0 5.0 6.0

Design - A

Design - B

Design - C

Design - D

To

rqu

e (p

.u.)

Rotor speed(p.u.)

152

Variation of power with respect to rotor speed is presented in Figs.

7.13 and 7.14 for Cases 1 and 2 respectively for various design options.

Power is increased in the constant power region when power factor

decreases as demonstrated in table 7.5. Increase of power for Design-D

compared to Design-A is about 33% and 60% for Cases 1 and 2 respectively.

The variation of rotor magnetizing current (p.u.) with respect to rotor

speed (p.u.) is shown in Figs. 7.15 and 7.16 for Cases 1 and 2 respectively.

The magnetizing current curves for various design options are varying for

the entire speed range for various design options in both cases.

Table 7.5 Maximum power (p.u.)

Rotor Speed (p.u.)

Case-1 (10 kW) Case-2 (30 kW)

A B C D A B C D

1.0 0.611 0.728 0.836 0.865 0.705 0.870 0.886 0.976

2.0 0.618 0.662 0.700 0.790 0.580 0.706 0.810 0.928

3.0 0.603 0.651 0.702 0.804 0.588 0.687 0.813 0.921

5.0 - - - - 0.465 0.505 0.520 0.830

Fig. 7.13 Power vs. Rotor speed in Case-1 (10 kW)

0.00

0.20

0.40

0.60

0.80

1.00

1.20

0.0 1.0 2.0 3.0 4.0

Design - A

Design - B

Design - C

Design - D

Ideal curve

Po

wer

(p

.u.)

Rotor speed (p.u.)

153

Fig. 7.14 Power vs. Rotor speed in Case-2 (30 kW)

Fig. 7.15 Magnetizing current vs. Rotor speed in Case-1 (10 kW)

0.00

0.20

0.40

0.60

0.80

1.00

1.20

0.0 1.0 2.0 3.0 4.0 5.0

Design - A

Design - B

Design - C

Design - D

Ideal curve

Po

wer

(p

.u.)

Rotor speed (p.u.)

0.0

0.2

0.4

0.6

0.8

1.0

0.0 1.0 2.0 3.0 4.0

Design - A

Design - B

Design - C

Design - D

Ma

gn

etiz

ing c

urr

ent

(p.u

.)

Rotor speed (p.u.)

154

Fig. 7.16 Magnetizing current vs. Rotor speed in Case-2 (30 kW)

By observing the simulation results, it ensures that improvement in

torque capability and power of sensorless induction motor drive in FW

region is possible by decreasing the power factor.

7.6. Summary

In this chapter, adaptability of induction motor for propulsion

application is discussed with emphasis to motor-load interaction. Design

options are generated with the spread sheets developed for design of

induction motor. Simulations are carried out in various design options for

motor characteristics. By analyzing the result, the major finding of this

research work is evolved, which is, improvement in torque capability and

power of sensorless induction motor drive in FW region is possible by

decreasing the power factor.

0.0

0.2

0.4

0.6

0.8

1.0

0.0 1.0 2.0 3.0 4.0 5.0

Design - A

Design - B

Design - C

Design - D

Ma

gn

etiz

ing

curr

ent

(p.u

.)

Rotor speed(p.u.)

155

7.7. Publications related to this Chapter

International Conference:

1. G. K. Nisha, Z. V. Lakaparampil and S. Ushakumari, “Effect of Leakage inductance

on Torque Capability of Field Oriented Controlled Induction Machine in Field

Weakening Region,” International Conference on Advances in Engineering and

Technology (ICAET’2014), Singapore, 29-30 March2014.

International Journal:

1. G. K. Nisha, Z. V. Lakaparampil and S. Ushakumari, “Torque Capability

Improvement of Sensorless FOC Induction Machine in Field Weakening Region for

Propulsion Purposes,” communicated for publication in Journal of Computers and

Electrical Engineering: Elsievier. (under Review)

2. G. K. Nisha, Z. V. Lakaparampil and S. Ushakumari, “Effect of Power factor on

Torque Capability of FOC Induction Machine in Field Weakening Region for

Propulsion Systems,” communicated for publication in Journal of Institution of

Engineers (India) Series–B: Springer. (under Review)