Hindawi Publishing CorporationActive and Passive Electronic ComponentsVolume 2011, Article ID 919240, 6 pagesdoi:10.1155/2011/919240

Research Article

Comb-Line Filter with Coupling Capacitor in Ground Plane

Toshiaki Kitamura

Faculty of Engineering Science, Kansai University, Suita, Osaka 564-8680, Japan

Correspondence should be addressed to Toshiaki Kitamura, kita@kansai-u.ac.jp

Received 18 January 2011; Accepted 1 March 2011

Academic Editor: Tzyy-Sheng Horng

Copyright © 2011 Toshiaki Kitamura. This is an open access article distributed under the Creative Commons Attribution License,which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.

A comb-line filter with a coupling capacitor in the ground plane is proposed. The filter consists of two quarter-wavelengthmicrostrip resonators. A coupling capacitor is inserted into the ground plane in order to build strong coupling locally alongthe resonators. The filtering characteristics are investigated through numerical simulations as well as experiments. Filteringcharacteristics that have attenuation poles at both sides of the passband are obtained. The input susceptances of even and oddmodes and coupling coefficients are discussed. The filters using stepped impedance resonators (SIRs) are also discussed, and theeffects of the coupling capacitor for an SIR structure are shown.

1. Introduction

Miniaturization of microwave filters is highly demanded. Formobile telephones especially, ceramic laminated filters [1–4]have been widely used, and in particular, comb-line filtershave extensively made practical use.

In this study, comb-line filters in which both sides ofthe substrate are utilized are considered. Comb-line filtersconsist of two quarter-wavelength resonators, and attenua-tion poles can be created in the frequency characteristics ofthe transmission parameter by changing the coupling locallyalong the resonators [5]. The stopband characteristics canbe improved by arranging the attenuation poles around thepassband. In [1, 2], strong coupling between two resonatorsis obtained by installing a patch conductor on the dielectricsubstrate above the resonators. The patch conductor isreferred to as a coupling capacitor (Cc). The method ofinstalling a coupling capacitor by inserting slots into theground plane is discussed. By this method, a couplingcapacitor can be achieved without using a multilayeredstructure. As a method of inserting slots into a ground plane,a defected ground structure (DGS) has been attracting muchattention [6–8]. In a broad sense, the proposed structure isa kind of DGS. The filtering characteristics are investigatedthrough numerical simulations as well as experiments. Thefilters using stepped impedance resonators (SIRs) are alsodiscussed, and the effects of the coupling capacitor for an SIRstructure are shown.

2. Filter Structure

Figure 1 shows an overview of the proposed comb-line filter.Two microstrip resonators are arranged on the substrate, andeach resonator is terminated through the ground plane atone end using a through hole. As an I/O port, a microstripline with a characteristic impedance of 50 ohm is directlyconnected to each resonator. A square-shaped couplingcapacitor is fabricated by inserting slots into the groundplane. The coupling capacitor is a patch conductor thatis not terminated through the ground plane and producesstrong coupling locally along the resonators. The thicknessand relative permittivity of the substrate are assumed tobe 1.27 mm and 10.2, respectively, and the diameter of thethrough hole is 0.3 mm.

The metallization patterns and dimensions of the filterare shown in Figure 2. The dimensions and position of thecoupling capacitor are also shown. The coupling capacitoris a × 5.2 mm2 and is located d mm from the open endsof the microstrip resonators. The filtering characteristics areinvestigated with a and d as parameters. The other structuralparameters are also shown in Figure 2.

3. Results and Discussion

The filtering characteristics are investigated through numer-ical simulations by the full-wave EM simulator Ansoft HFSS

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I/O port I/O port

1.27 mm

Symmetric planeShading: conductor

Figure 1: Whole structure of filter.

0.40.4

a

5.2

2.4

Symmetric plane

1.2

1

0.4

0.4

0.4

0.4

10.2d

Unit: mmShading: conductor

Figure 2: Metallization patterns.

Ver. 11. The frequency characteristics of the scatteringparameters when a = 5.2 mm and d = 1.0 mm are shownin Figure 3. For comparison, the results when there is nocoupling capacitor (Cc) are also shown. It can be seen thatan attenuation pole was created at each side of the passbandby installing the coupling capacitor. The center frequencyof the passband was also increased slightly by insertingslots into the ground plane. The slots also cause radiationloss, and it is understood from Figure 3 that 1 − |S11|2 −|S21|2 = 0.090 at 2.1 GHz when a coupling capacitor is used.Figure 4 illustrates the equivalent circuit of the comb-linefilter. Attenuation poles appear at the frequencies where theinput susceptances of the even and odd modes are equal toeach other.

The frequency characteristics of the input susceptancesof the even and odd modes Be

in and Boin are shown in Figures

5(a) and 5(b), respectively. The structural parameters were

1 2 3 4−70

−60

−50

−40

−30

−20

−10

0−30

−20

−10

0

Frequency (GHz)

With Cc

Without Cc

|S 21|(

dB)

|S 11|(

dB)

Figure 3: Frequency characteristics of scattering parameters.

jBo

in − Bein

2

jBo

in + Bein

2jBe

in − Boin

2jBe

in − Boin

2jBo

in + Bein

2

Figure 4: Equivalent circuit.

the same as those in Figure 3. The input susceptances ofthe even and odd modes were calculated by setting themagnetic and electric wall, respectively, on the symmetricplane shown in Figure 1. As shown in these figures, theodd-mode susceptances hardly changed, whether or notthere was a coupling capacitor. The coupling capacitor doesnot have much effect on the electromagnetic fields of theodd mode. On the other hand, the even-mode susceptanceswere decreased by inserting the coupling capacitor, and theyintersected with the odd-mode ones at 1.62 and 2.81 GHz, asshown in Figures 5(a) and 5(b), respectively. The frequenciesof the intersections correspond with the attenuation-polefrequencies in Figure 3.

The frequency characteristics of the scattering parame-ters with d as a parameter when a = 5.2 mm are shown inFigures 6(a) and 6(b). Here, the parameter d correspondsto the position of the coupling capacitor. As shown inFigure 6(a), when d is small (from 0 to 1.0 mm), twoattenuation poles appear at both sides of the passband,and the parameter d mainly affects the attenuation-polefrequency above the passband. On the other hand, as shownin Figure 6(b), no attenuation pole appears when d is large(from 4.0 to 6.0 mm). The resonant frequency of the odd

Active and Passive Electronic Components 3

1.55 1.6 1.65

−0.006

−0.004

−0.002

Even mode (with Cc)Odd mode (with Cc)

Even mode (without Cc)Odd mode (without Cc)

Frequency (GHz)

Bin

(S)

(a)

Frequency (GHz)

Bin

(S)

2.75 2.8 2.85−0.001

0

0.001

0.002

Even mode (with Cc)Odd mode (with Cc)

Even mode (without Cc)Odd mode (without Cc)

(b)

Figure 5: Frequency characteristics of input susceptances around attenuation-pole frequencies (a) below and (b) above passband.

−70

−60

−50

−40

−30

−20

−10

0

1 2 3 4

−30

−20

−10

0

Frequency (GHz)

d = 0 mmd = 0.5 mmd = 1 mm

|S 21|(

dB)

|S 11|(

dB)

(a)

d = 4 mmd = 5 mmd = 6 mm

1 2 3 4

Frequency (GHz)

−70

−60

−50

−40

−30

−20

−10

0−30

−20

−10

0

|S 21|(

dB)

|S 11|(

dB)

(b)

Figure 6: Frequency characteristics of scattering parameters when (a) d = 0 ∼ 1.0 mm and (b) d = 4.0 ∼ 6.0 mm (a = 5.2 mm).

mode is about 2.0 GHz and changes very little when changingd. However, the resonant frequency of the even modedecreases as d increases. It was confirmed that the even-moderesonant frequency becomes lower than the odd-mode onewhen d exceeds about 2.5 mm.

The frequency characteristics of the input susceptancesof an even mode are shown in Figure 7. Here, the frequencyrange is chosen so as to be close to the attenuation-polefrequency above the passband. The structural parameterswere the same as in Figure 6(a). As shown in this figure,

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2.6 2.8 3 3.2 3.4

−0.005

0

0.005

0.01

d = 0.5 mmd = 1 mm

d = 0 mm

Frequency (GHz)

Bin

(S)

Figure 7: Frequency characteristics of input susceptances of evenmode (a = 5.2 mm).

the input susceptances of an even mode change almost inparallel to each other with changing d. According to this, theattenuation-pole frequencies above the passband change asshown in Figure 6(a).

However, from Figure 6, the parameter d also has alarge effect on the bandwidth of the passband. The couplingcoefficients as a function of d are shown in Figure 8. Thecoupling coefficient k is calculated using the followingequation

k = f 2e − f 2

o

f 2e + f 2

o. (1)

Here, fe and fo are the resonant frequencies of the even andodd modes, respectively, and they are determined from thezero-crossing points of the input susceptance curves of theeven and odd modes. As can be seen, the coupling coefficientsdecrease steadily as d increases.

Figure 9 shows the frequency characteristics of the scat-tering parameters with a as a parameter when d = 0 mm.Here, the parameter a is the length of the patch conductorthat makes up the coupling capacitor. As shown in this figure,the attenuation-pole frequencies decrease, and, in contrast,the passband frequency shifts slightly higher as a increases.The coupling coefficients as a function of a are shownin Figure 10. It can be seen that the coupling coefficientsincrease almost linearly as a increases.

Next, an SIR filter shown in Figure 11 is studied.Filters can be miniaturized by using an SIR structure. Thefrequency characteristics of scattering parameters are shownin Figure 12. The solid line shows the results of a normal SIR

0 0.5 1 1.5 2

0.02

0.04

0.06

0.08

0.1

d (mm)

Cou

plin

gco

effici

ents

Figure 8: Coupling coefficients as a function of d (a = 5.2 mm).

1 2 3 4−70

−60

−50

−40

−30

−20

−10

0

Frequency (GHz)

a = 5.2 mma = 6.2 mma = 7.2 mm

−30

−20

−10

0

|S 21|(

dB)

|S 11|(

dB)

Figure 9: Frequency characteristics of scattering parameters (d =0 mm).

filter (without Cc) when s1 = 0.1 mm and s2 = 0.2 mm.It is shown that the passband frequency becomes lowercompared with that in Figure 3, meaning that it is possible tominiaturize the filter. In addition, two attenuation poles canbe created at both of the passbands by choosing appropriatevalues for parameters s1 and s2. However, the values of s1 ands2 are quite small. The dotted line shows the results of anSIR filter with Cc when s1 = 0.4 mm and s2 = 0.9 mm. Itis understood that attenuation poles can be created near the

Active and Passive Electronic Components 5

0.04

0.08

0.12

0.16

a (mm)

Cou

plin

gco

effici

ents

4 5 6 7 8

Figure 10: Coupling coefficients as a function of a (d = 0 mm).

0.4 0.4

0.4

5.2

2.4

1.2

1

0.4

0.4

6.2

4

5.2

1

s2

s13.6

Symmetric plane

Unit: mmShading: conductor

Figure 11: Metallization patterns of SIR filter.

passband by choosing s1 and s2 that are easy to fabricate, andtherefore, the structural limitation can be relaxed by usinga coupling capacitor [2]. However, a drawback is that thepassband frequency becomes higher compared with a normalSIR filter. This is due to the decrease of the effective relativepermittivity by the installation of slots in the ground plane.For comparison, the results of the filter shown in Figure 2(a = 5.2 mm, and d = 1.0 mm) are also shown (dashed line).

1 2 3 4−70

−60

−50

−40

−30

−20

−10

0

Frequency (GHz)

−30

−20

−10

0

SIR (without Cc)SIR (with Cc)Straight (with Cc)

|S 21|(

dB)

|S 11|(

dB)

Figure 12: Frequency characteristics of scattering parameters (solidline: s1 = 0.1 mm and s2 = 0.2 mm (without Cc), dotted line: s1 =0.4 mm and s2 = 0.9 mm (with Cc), and dashed line: a = 5.2 mmand d = 1.0 mm (Figure 2)).

1 2 3 4−70

−60

−50

−40

−30

−20

−10

0

Frequency (GHz)

−30

−20

−10

0

ExperimentSimulation

|S 21|(

dB)

|S 11|(

dB)

Figure 13: Frequency characteristics of scattering parameters (a =5.2 mm and d = 1.0 mm (Figure 2)).

Finally, the filtering characteristics of our developed filterare investigated through experiments. The proposed filtershown in Figure 2 is manufactured on an RT/duroid 6010LMsubstrate of 1.27-mm thickness and 10.2 relative permittivity.The frequency characteristics of the scattering parameterswhen a = 5.2 mm and d = 1.0 mm are shown in Figure 13.Here, the solid and dashed lines indicate the experimentaland numerical results, respectively. As can be seen, bandpass

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characteristics with an attenuation pole both below andabove the passband were achieved. The experimental resultswere also in good agreement with the numerical ones. Fromthis figure, it is estimated that the influence of processvariation may cause the degradation of impedance matching.

4. Conclusion

A comb-line filter with a coupling capacitor in the groundplane was proposed. The insertion of the coupling capacitorbuilds strong coupling locally along the resonators in thefilter. The filtering characteristics were investigated throughnumerical simulations as well as experiments, and thefiltering characteristics having attenuation poles at bothsides of the passband were obtained. The input susceptancesof even and odd modes and coupling coefficients werediscussed. The filters using SIRs were also discussed, and theeffects of the coupling capacitor for an SIR structure wereshown.

References

[1] T. Ishizaki, M. Fujita, H. Kagata, T. Uwano, and H. Miyake,“Very small dielectric planar filter for portable telephones,”IEEE Transactions on Microwave Theory and Techniques, vol. 42,no. 11, pp. 2017–2022, 1994.

[2] T. Ishizakl, T. Uwano, and H. Miyake, “An extended con-figuration of a stepped impedance comb-line filter,” IEICETransactions on Electronics, vol. 79, no. 5, pp. 671–677, 1996.

[3] T. Ishizaki, T. Kitamura, M. Geshiro, and S. Sawa, “Study ofthe Influence of Grounding for Microstrip Resonators,” IEEETransactions on Microwave Theory and Techniques, vol. 45, no.12, pp. 2089–2093, 1997.

[4] T. Kitamura, M. Geshiro, T. Ishizaki, T. Maekawa, and S. Sawa,“Characterization of triplate strip resonators with a loadingcapacitor,” IEICE Transactions on Electronics, vol. 81, no. 12, pp.1793–1798, 1998.

[5] H. Egami, T. Kitamura, and M. Geshiro, “Study on meander-shaped microstrip comb-line filter,” The Institute of ElectricalEngineers of Japan, vol. 125, no. 10, pp. 1596–1601, 2005.

[6] A. M. E. Safwat, F. Podevin, P. Ferrari, and A. Vilcot,“Tunable bandstop defected ground structure resonator usingreconfigurable dumbbell-shaped coplanar waveguide,” IEEETransactions on Microwave Theory and Techniques, vol. 54, no.9, Article ID 1684152, pp. 3559–3564, 2006.

[7] M. Wang, Y. Chang, H. Wu, C. Huang, and Y. Su, “An inverses-shaped slotted ground structure applied to miniature widestopband lowpass filters,” IEICE Transactions on Electronics, vol.90, no. 12, pp. 2285–2288, 2007.

[8] J. Yang, C. Gu, and W. Wu, “Design of novel compact coupledmicrostrip power divider with harmonic suppression,” IEEEMicrowave and Wireless Components Letters, vol. 18, no. 9, pp.572–574, 2008.