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Order this document by AN1308/D MOTOROLA SEMICONDUCTOR APPLICATION NOTE MOTOROLA INC., 1992 AN1308 Prepared by: Andrew Hefley Audio Engineering Consultant INTRODUCTION Over the past two decades many types of solid state, high fidelity audio amplifier design approaches have been tried. Many of these designs have used large amounts of negative feedback to ensure low closed-loop harmonic distortion. The main contributors to this type of distortion are the output de- vices. Other contributors to this harmonic distortion are the drivers, or devices preceding the output devices and the high voltage gain stage, also referred to as the transconductance stage. A side effect of a high open-loop gain is that wide bandwidth is very difficult to achieve with two gain stages, therefore most of these high gain designs have an open-loop bandwidth of less than a few kilohertz. This means there is a need for even more than necessary gain mid-band (refer- ring to the audio band) to get a low distortion number at 20 kHz in the closed-loop condition. The audio band is de- fined as 2 Hz to 20 kHz with the mid-point from 1 kHz to 2 kHz. It was found that large amounts of negative feedback increased TIM (transient intermodulation distortion). The approach to dealing with this problem in amplifiers is to use lower open-loop voltage gain sections and increase the open-loop bandwidth to 20 kHz. The loss of feedback (and increased closed-loop distortion) has inspired different solutions for the output section as well as the voltage gain section. One approach is to change the distortion specification; however, this may not produce optimal results. A better solution is to use multiple output devices to keep the current excursion low for each device, so as to stay within a linear range of gain. This has been a popular approach with some manufacturers; however, it is a more costly solution. Other approaches use high feedback, linearized, unity gain output stages. This is done with either bipolar devices or with power MOSFETs. This note focuses on the use of a new pair of complementary Motorola bipolar power output transistors, the 2SC3281 and 2SA1302. These devices have better linearity than devices previously targeted for this type of application. The amplifier circuits presented use a topology that is fully complementary in design with a dual differential input. Other parameters sought after are a wide open-loop bandwidth, (greater than the audio band) and a minimal amount of negative feedback, (25 dB or less). BASIC AMPLIFIER DESIGN PHILOSOPHY A conservative design approach was taken with effort made to keep the circuitry design simple. The purpose of this note is to show that a low feedback design with wide bandwidth can yield quite low distortion without any special distortion cancelling circuitry or localized feedback loops. Design Description of the 100 Watt Amplifier A block diagram of the 100 watt amplifier is shown in Figure 3. This design begins with the MPS8099 and MPS8599 complementary pair for the input stage. These devices are rated at 80 volts, giving them adequate margin on a nominal 52 volt supply. These devices are arranged as dual differentials. For simplicity, no current sources are used. A ± 24 volt zener with 10K ohm resistors supplies approximately 2.25 mA of current to each pair. This scheme supplies enough current to achieve the bandwidth necessary in the first stage while keeping the bias currents low enough for an acceptable amount of input voltage offset error. In an attempt to eliminate a coupling capacitor in the feedback loop, low offset is important when trying to DC couple the feedback. The input impedance of the amplifier is primarily determined by the 33.2K ohm resistor used as the input return path to ground. That, coupled with the 2.2 μF coupling capacitor, sets the low corner frequency at approximately 2 Hz.

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Page 1: Amplificator Motorola

Order this documentby AN1308/DMOTOROLA

SEMICONDUCTORAPPLICATION NOTE

MOTOROLA INC., 1992 AN1308

# ! $ %"

Prepared by: Andrew HefleyAudio Engineering Consultant

INTRODUCTION

Over the past two decades many types of solid state, highfidelity audio amplifier design approaches have been tried.Many of these designs have used large amounts of negativefeedback to ensure low closed-loop harmonic distortion. Themain contributors to this type of distortion are the output de-vices. Other contributors to this harmonic distortion are thedrivers, or devices preceding the output devices and the highvoltage gain stage, also referred to as the transconductancestage. A side effect of a high open-loop gain is that widebandwidth is very difficult to achieve with two gain stages,therefore most of these high gain designs have an open-loopbandwidth of less than a few kilohertz. This means there isa need for even more than necessary gain mid-band (refer-ring to the audio band) to get a low distortion number at20 kHz in the closed-loop condition. The audio band is de-fined as 2 Hz to 20 kHz with the mid-point from 1 kHz to2 kHz. It was found that large amounts of negative feedbackincreased TIM (transient intermodulation distortion).

The approach to dealing with this problem in amplifiersis to use lower open-loop voltage gain sections and increasethe open-loop bandwidth to 20 kHz. The loss of feedback(and increased closed-loop distortion) has inspired differentsolutions for the output section as well as the voltage gainsection. One approach is to change the distortionspecification; however, this may not produce optimal results.A better solution is to use multiple output devices to keepthe current excursion low for each device, so as to stay withina linear range of gain. This has been a popular approachwith some manufacturers; however, it is a more costlysolution. Other approaches use high feedback, linearized,unity gain output stages. This is done with either bipolardevices or with power MOSFETs.

This note focuses on the use of a new pair ofcomplementary Motorola bipolar power output transistors,

the 2SC3281 and 2SA1302. These devices have betterlinearity than devices previously targeted for this type ofapplication. The amplifier circuits presented use a topologythat is fully complementary in design with a dual differentialinput. Other parameters sought after are a wide open-loopbandwidth, (greater than the audio band) and a minimalamount of negative feedback, (25 dB or less).

BASIC AMPLIFIER DESIGN PHILOSOPHY

A conservative design approach was taken with effortmade to keep the circuitry design simple. The purpose ofthis note is to show that a low feedback design with widebandwidth can yield quite low distortion without any specialdistortion cancelling circuitry or localized feedback loops.

Design Description of the 100 Watt Amplifier

A block diagram of the 100 watt amplifier is shown inFigure 3. This design begins with the MPS8099 andMPS8599 complementary pair for the input stage. Thesedevices are rated at 80 volts, giving them adequate marginon a nominal 52 volt supply. These devices are arrangedas dual differentials. For simplicity, no current sources areused. A ±24 volt zener with 10K ohm resistors suppliesapproximately 2.25 mA of current to each pair. This schemesupplies enough current to achieve the bandwidth necessaryin the first stage while keeping the bias currents low enoughfor an acceptable amount of input voltage offset error. In anattempt to eliminate a coupling capacitor in the feedbackloop, low offset is important when trying to DC couple thefeedback. The input impedance of the amplifier is primarilydetermined by the 33.2K ohm resistor used as the inputreturn path to ground. That, coupled with the 2.2 µF couplingcapacitor, sets the low corner frequency at approximately2 Hz.

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Figure 1. 100 Watt High Fidelity Audio Amplifier

Figure 2. 200 Watt High Fidelity Audio Amplifier

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Figure 3. Block Diagram of 100 Watt Amplifier

INPUT FEEDBACKINPUT STAGE

TRANSCONDUCTANCE STAGE

OUTPUT

OUTPUTSTAGE

PROTECTION

PROTECTION

TRANSCONDUCTANCE STAGE

OUTPUTSTAGE

+ VCC

–VCC

The second stage of voltage gain, sometimes referredto as the transconductance stage, is made up of a darlingtonpair. The input devices of the darlington pairs are theMPSW06 and MPSW56 respectively. Both of these devicesare also rated at 80 volts. These devices are operated ina common collector mode with their collectors grounded tominimize the Miller effect. Idling current is approximately3.5 mA with 511 ohm emitter resistors which in turn set theidling currents for the 2SC3298B and 2SA1306B. Becausethe idling currents are set at approximately 50 mA or2.4 watts in each of these devices, a small heat sink isrequired to keep the case temperatures down. The2SC3298B and 2SA1306B devices are rated at 200 voltswhich is more than adequate to handle the nominal voltageswings of 100 volts for this stage of the amplifier. Thetransconductance stage is loaded by both the output stageand a pair of 2.7K ohm resistors. These resistors set thevoltage gain at this stage. Looking at the input stage andthe second stage respectively, their gains are approximately

18.5 dB for the input stage and 34 dB for the second stagethus giving an overall gain of about 52.5 dB. Compensationnetworks are used on the outputs of both stages to providegood gain and phase margin for the closed-loop condition.The closed-loop gain is set for approximately 1 volt sensitivitygiving the amplifier a closed-loop gain of 27.8 dB.

The output stage is a complementary darlingtonconfiguration. This stage utilizes three 2SC3281 NPNdevices and three 2SA1302 PNP devices connected inparallel. These are driven by the complementary pairconsisting of an MJF15030 and MJF15031. The outputdevices are rated at 15 amps and 200 volts with powerdissipation ratings of 150 watts. The drivers are 8 amp,150 volt transistors with power dissipation ratings of36 watts. The voltage ratings are adequate to handle the100 volt nominal supply voltage. The drivers and outputdevices have excellent gain linearity which helps to minimizethe amount of feedback needed to achieve a low distortionnumber.

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The MPS650 and the MPS750 devices are used forcurrent limit protection. Both devices are rated at 2 ampsgiving them excellent saturation and gain characteristics at100 mA. A discussion on how the current limiters are setand the choice of the number of output devices used isdiscussed in the Output Transistors section.

Design Description of the 200 Watt AmplifierA block diagram of the 200 watt amplifier is shown in

Figure 4. The design of the 200 watt gain stages is verysimilar to the 100 watt amplifier with a few minor exceptions.Due to higher power supply voltages, a cascodeconfiguration was used for the input stage. The level shifterportion of the cascode is tied to the 33 volt zener supplieswhich are used for the input current resistors. An additionalchange is the use of paralleled pre-drivers, or transconduc-tance stage transistors. This accommodates the increased

current needed to bring the dual 2.7K ohm load andincreased base current requirement of the output stage tothe higher supply voltage. The benefit is a small increasein the voltage slew rate. The extra device also increases theopen-loop gain by approximately 4 dB. This in turn helpsthe closed-loop distortion that was increased because of a3 dB increase in closed-loop gain. This was done to keepthe sensitivity of the amplifier for full power at 1 volt. Anotherchange is the use of a cascode (series output stage). Byeffectively doubling the number of output devices withoutincreasing the voltage seen in operation, second breakdownwill not be a concern. The outside, or slave devices, aredriven by a series resistive divider network tied to the outputof the amplifier. This divider network forces the string ofoutput devices to share the voltage and power delivered tothe load.

Figure 4. Block Diagram of 200 Watt Amplifier

INPUT FEEDBACKINPUT STAGE

TRANSCONDUCTANCE STAGE

OUTPUT

OUTPUTSTAGE

PROTECTION

PROTECTION

TRANSCONDUCTANCE STAGE

OUTPUTSTAGE

+ VCC

–VCC

–33 V

+33 V

SLAVE

SLAVE

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POWER SUPPLY DESIGN

There are a number of formulas and philosophiespertaining to the selection of the size of a power transformerand the supply filters for use in an audio amplifier. Theselection of these components affects the supply regulationand the amount of ripple the amplifier will see under varyingload conditions. One specification that has come intowidespread use over the past decade is a concept calleddynamic headroom. Dynamic headroom is a measure of thedifference in the RMS power an amplifier can delivercontinuously and the power it can deliver dynamically asdefined by the Federal Trade Commission. It is measuredin dB, and can range from 0 dB for an amplifier operatingin a full class A mode to 3–4 dB for a class AB type amplifierwith poor supply regulation. Over the years designers havefound that they prefer a lower dynamic headroom, whichmake certain “rules of thumb” work well and providesacceptable performance. For transformer selection, take theexpected 4 ohm full power performance in watts, forexample, 180 watts for the 100 watt amplifier, and doubleit. This could be considered a worst case load for the powersupply even though under typical operating conditions thislevel of output will never be seen. By doubling the powerrating, a VA (Volt-Ampere) rating of 360 becomes the ratingfor the transformer to be selected. VA ratings typically arebased on a temperature rise of +55° to +65°C, whichequates to a regulation of approximately 5% for a resistiveload equal to the VA rating of the transformer. This does notequate to the regulation performance in the amplifierapplication. DC supply voltages are derived by the rectifieddiodes that charge the supply filters and these only occurduring the voltage peaks of the sine wave input. 360 VA and600 VA transformers were chosen for the 100 watt and 200watt amplifiers respectively. To avoid the cost of customtransformers, standard off-the-shelf toroidal transformerswere chosen. These transformers were then modified. Toachieve the exact voltage required for the designs, severalextra turns were added to the secondary winding (seeFigures 25 and 34).

Cost becomes an issue in choosing the amount of filteringneeded. In practice, the amount of filtering can be lookedat in terms of stored energy, or joules. Typically 1 to 2 joulesof stored energy per 10 watts of output power is sufficient.In the 100 watt amplifier, the maximum output power is 180watts at 4 ohms. This equates to 18 to 36 joules of storedenergy. The power supply in the 100 watt amplifier has apair of 10,000 µF filter capacitors with 52 volts across eachcapacitor. Using the formula 1/2CV2, this equates to 26

joules of stored energy. The 200 watt amplifier has amaximum output power at 4 ohms. Based on the previousdiscussion, 30 to 60 joules of stored energy is required. Thepower supply for the 200 watt amplifier contains four4,700 µF filter capacitors with 80 volts across eachcapacitor. This equates to 60 joules of stored energy.

OUTPUT TRANSISTORS

There are a number of considerations to be addressedwhen selecting output transistors for use in a high fidelityaudio amplifier design. Two key considerations are the useof a complementary pair and the type of device packaging,i.e., plastic or metal. Newer designs are using plasticpackages due to their simplicity in mounting and thermalperformance equalling the older metal packages such as theTO-204 (TO-3). Other areas of design importance includebreakdown voltage, power dissipation, safe operating area(SOA), current gain linearity, and fT. At present, there area limited number of complementary devices rated at 100volts in plastic packages. Additionally, very few of thesedevices have good current gain linearity beyond one amp.Most of these devices have second breakdown points thatusually fall between 20 and 40 volts. They may be classifiedas 150 watt transistors, but operate efficiently only up to 40volts; then their power handling capability drops off rapidly.

The output devices that were selected are the 2SC3281and 2SA1302 NPN and PNP transistors. These complemen-tary devices are rated at 200 volts, 15 amps, and have powerdissipation ratings of 150 watts. They are packaged in theTO-3PBL package, a high power plastic package with anisolated mounting hole and excellent thermal characteristics.The thermal resistance junction-to-case of this package isless than 0.83°C/watt with a maximum junction temperature,TJ equal to 150°C. The current gain is adequate beyond 3amps of collector current, the fT is greater than 20 MHz andthe second breakdown point is greater than 70 volts. Withthese specifications these devices are ideal for use underthe type of load conditions and voltages expected to be seenin low distortion wideband linear amplifier designs. Operatingcharacteristics for these devices are shown in Figures 5through 10.

Setting the Current Limits

Setting the current limits of an output stage in an audioamplifier is not an easy task. Calculation can be used to finda starting point, however the actual results must bedetermined through experimentation. There are twolimitations to consider when dealing with the power handling

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Figure 5. DC Current Gain Figure 6. DC Current Gain

1000

500

300

100

50

30

10

5– 0.03 – 0.1 – 0.3 – 1 – 3 – 10 – 30

IC, COLLECTOR CURRENT (AMPS)

hFE

, DC

CU

RR

ENT

GAI

N

25°C

– 25°C

COMMON EMITTERVCE = – 5 V

TC = 100°C

1000

500

300

100

50

30

10

5

hFE

, DC

CU

RR

ENT

GAI

N

0.03 0.1 0.3 1 3 10 30

IC, COLLECTOR CURRENT (AMPS)

200

100

50

30

10

5

3

1

Figure 7. Current-Gain Bandwidth Product

– 0.01

IC, COLLECTOR CURRENT (AMPS)

– 0.03 – 0.1 – 0.3 – 1 – 3 – 10

COMMON EMITTERVCE = 5 V

TC = 100°C

25°C

– 25°C

COMMON EMITTERVCE = – 5 VTC = 25°C

Figure 8. Current-Gain Bandwidth Product

0.01

IC, COLLECTOR CURRENT (AMPS)

0.03 0.1 0.3 1 3 10

200f T

, TR

ANSI

TIO

N F

REQ

UEN

CY

(MH

z) 100

50

30

10

5

3

1

COMMON EMITTERVCE = 5 VTC = 25°C

f T, T

RAN

SITI

ON

FR

EQU

ENC

Y (M

Hz)

– 50

I C, C

OLL

ECTO

R C

UR

REN

T (A

MPS

) – 30

– 10

– 5– 3

– 1

– 0.5

– 0.3

– 0.1

Figure 9. Active Region Safe Operating Area

– 0.3

VCE, COLLECTOR-EMITTER VOLTAGE (VOLTS)

– 1 – 3 – 10 – 30 – 100 – 300 – 1000

IC MAX (CONTINUOUS)

DC OPERATIONTC = 25°C

*SINGLE NONREPETITIVE*PULSE TC = 25°C

CURVES MUST BE DERATEDLINEARLY WITH INCREASEIN TEMPERATURE

IC MAX (PULSED)*

1 ms*

10 ms*

100 ms*

VCEO MAX

50

I C, C

OLL

ECTO

R C

UR

REN

T (A

MPS

) 30

10

53

1

0.5

0.3

0.1

Figure 10. Active Region Safe Operating Area

0.3

VCE, COLLECTOR-EMITTER VOLTAGE (VOLTS)

1 3 10 30 100 300 1000

IC MAX (CONTINUOUS)

DC OPERATIONTC = 25°C

*SINGLE NONREPETITIVE*PULSE TC = 25°C

CURVES MUST BE DERATEDLINEARLY WITH INCREASEIN TEMPERATURE

IC MAX (PULSED)*

1 ms*

VCEO MAX

10 ms*

100 ms*

OPERATING CHARACTERISTICS FOR OUTPUT DEVICES

2SA1302 — PNP 2SC3281 — NPN

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Figure 11. Thermal Response of Output Devices

0.01

0.020.03

0.050.07

0.1

0.2

0.3

0.50.7

1

0.01 0.02 0.03 0.05 0.1 0.2 0.3 0.5 1 2 3 5 10 20 30 50 100 200 300 500 1000

t, TIME (ms)

r(t),

EFFE

CTI

VE T

RAN

SIEN

T TH

ERM

ALR

ESIS

TAN

CE

(NO

RM

ALIZ

ED)

D = 0.5

0.2

0.1

0.05

0.02

0.01

SINGLE PULSE

P(pk)

t1t2

DUTY CYCLE, D = t1 / t2

RθJC(t) = r(t) RθJCRθJC = 0.83°C / W MAXD CURVES APPLY FOR POWERPULSE TRAIN SHOWNREAD TIME AT t1TJ(pk) – TC = P(pk) RθJC(t)

ability of a transistor: the average junction temperature andsecond breakdown. In a class AB output stage, the outputdevices are not really in a 50% duty cycle situation. The biascurrent needs to be added to the calculated current that theload may present to the device. At higher frequencies thepeak power can be considerably higher than the averagepower. At low frequencies the duration that one side of theoutput stage may endure during a load condition may beseveral hundred milliseconds, which nearly constitutes a DCcondition. By examining the thermal response curve of theoutput devices in Figure 11, the actual value of RθJC for thelow frequency condition can be determined by multiplyingthe specified value of RθJC by r(t) = 0.9. With this information,there are three more parameters that connect the transistorjunction to its surroundings — the thermal resistance fromthe transistor junction to the case of the transistor, thethermal resistance of the mounting interface, and the thermalresistance of the heat sink to air.

Figure 12 shows the load conditions for the entire outputstage of the 100 watt amplifier. The load line for an 8 ohm,45 degree load indicates the need for 6 amps of collectorcurrent dropping to 4 amps of collector current when thecollector voltage drops to 50 volts. Figure 12 also shows thepeak currents at lower resistive load conditions. Mostloudspeakers present a load impedance of less than 8 ohms;however, most don’t present a reactive load of less than8 ohms. As shown in Figure 12, a 90° reactive 8 ohm loadis handled easily and resistive loads as low as 2 ohms donot present a problem. The constant power curve shown at375 watts is drawn along the line where the current limitswere set for this amplifier. If this power is shared between3 devices, they will each be required to dissipate 125 wattsat the point where the power will be limited by the currentlimiters.

By limiting the power linearly from 150 watts at 25°C to0 at 150°C, the case temperature can rise to approximately60°C. This is determined from Figure 14, the power deratingcurve. By allowing the case temperature to rise to this limit,the power output will have to be limited to 75% of 150 watts,

or 112.5 watts. With this number and the information knownabout the thermal response of the device, one can allow thepeak power dissipated by each transistor to reach 125 watts.This means that under operating conditions where thetemperature is less than 60°C, the output devices will beoperating safely. Examining the safe operating area curvesin Figures 9 and 10, it can be seen that above 70 volts, thepower the output devices can dissipate, begins to drop.Using 100 volts as a reference, the power dissipation hasdropped from 150 watts at 70 volts to about 800 mA at 100volts, or 80 watts. The power derating curve (Figure 14)shows a reduction in power dissipation as temperature isincreased with the derating factor dropping even lower forthe second breakdown derating. The 60°C point on this curveis 85%, or 68 watts. This implies that the output devices areonly safe for a total power dissipation of 204 watts whenoperating the three devices at the full supply voltage of the100 watt amplifier. This is where the current limiters do notfully protect the output devices. As shown in the load lineplots in Figures 12 and 13, the protection provided by thecurrent limiters allows the power to reach 375 watts at100 volts. A fully reactive load of 90 degrees will never beseen in a loudspeaker and the current limits need to be setby experimentation. Figures 15 and 16 show photos of thecurrent limit, or protection circuit tested into a capacitor.Although Figures 15 and 16 show that the output devicesare capable of more than their published specifications, itis recommended that one not exceed published limits asshown on the manufacturer’s data sheets in the interestsof achieving long term reliability.

Measurements of the protection circuit shown in Figures15 and 16 show the actual current limit points of theamplifiers when driving large capacitive loads at a frequencyof approximately 100 Hz. As measured, the limits are setabout 25% higher than needed for the load conditionsoutlined. The type of current limiters used begins to limit thecurrent slightly lower than their final value causing substantialdistortion in the form of compression before the final limitvalue is reached.

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Figure 12. Load Lines of 100 Watt Amplifier

Figure 13. Load Lines of 200 Watt Amplifier

15

12

9

6

3

0

– 3

– 6

– 9

– 12

– 15100806040200

VOLTAGE ACROSS OUTPUT TRANSISTORS (VOLTS)

I C, C

UR

REN

T IN

OU

TPU

T TR

ANSI

STO

RS

(AM

PS)

375 WATT CONSTANT POWER CURVE

8Ω 45 DEG LOAD LINE

4Ω RESISTIVE2Ω RESISTIVE

POWER SUPPLY LIMITATIONCURRENT LIMITATION

8Ω RESISTIVE

8Ω RESISTIVE

25

20

15

10

5

0

– 5

– 10

– 15

– 20

– 25

I C, C

UR

REN

T IN

OU

TPU

T TR

ANSI

STO

RS

(AM

PS)

1601289664320VOLTAGE ACROSS OUTPUT TRANSISTORS (VOLTS)

750 WATT CONSTANT POWER CURVE

8Ω 45 DEG LOAD LINE

2Ω RESISTIVE4Ω RESISTIVE

CURRENT LIMITATION

CURRENT LIMIT

CURRENTLIMIT

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Figure 14. Power Derating Factor for Output Devices

0.2

0.4

0.6

0.8

1

020 40 60 80 100 120 140 160

TC, CASE TEMPERATURE (°C)

POW

ER D

ERAT

ING

FAC

TOR

SECOND BREAKDOWNDERATING

THERMALDERATING

Series Connected Output Devices

As discussed previously, the second breakdown of atransistor can severely limit the power dissipation capabilityof that device. When the supply voltages of an amplifier aregreater than 100 volts, the output devices are pushing theirlimits. By configuring the output devices in a series-parallelconfiguration one can obtain an increase in output powerfrom a set of devices rather than configuring them all inparallel. The load lines of the 200 watt amplifier indicate aconstant power curve of 750 watts, which is twice that ofthe 100 watt amplifier. There are twice the number of outputdevices connected in a series-parallel configuration resultingin the same 125 watt criteria for each as in the 100 wattamplifier. Since these devices do not see more than 80 voltseach in operation, second breakdown is of no concern.Although the devices can be operated in a series connectedoutput stage, its operation is similar to a bridge configurationbut the low impedance performance is diminished. Thisamplifier does current limit when driving a 2 ohm load wherethe 100 watt amplifier does not.

HEATSINK REQUIREMENTS

Choosing the right heatsink is very important. Regardlessof the requirements of size, shape, form factor andcosmetics, the bottom line is heat transfer in terms of degreescentigrade per watt (°C/W). The heatsink chosen for thesetwo amplifiers is a standard aluminum extrusion made byAHAM TOR INC., model #6071 as shown in Figure 17. Itis a tree shaped extrusion weighing 2.2 pounds per foot. Ithas a surface area of 31.7 square inches per linear inch anda convection heat transfer rating of 1.8°C/W per 3 inch piece.The design of this heatsink lends itself well to theconfiguration used for mounting the output and driverdevices. By using the PC board to clamp the devices to theheatsink, mounting hardware is eliminated. Anotheradvantage to having the heatsinks board mounted is thatthe output transistors need not be electrically isolated fromtheir mounting. This allows for nothing but thermal compound

Figure 15. Output Current Limit of 100 Watt Amplifier

Figure 16. Output Current Limit of 200 Watt Amplifier

between the devices and the heatsink. This allows foroptimum heat transfer (less than 0.1°C/W). The use of two7 inch extrusions in the 100 watt amplifier gives the outputstage the ability to dissipate about 90 watts while keepinga temperature rise of less than 35° above room temperature(TA = 25°C). This is based on 1.8°C of rise for a 3 inch piecemaking a 7 inch piece about 0.77°C/W. Tests run haveshown that with the 100 watt amplifier running at full power,the heatsinks stayed below 60°C. Operation with air flowingacross the heatsinks will give improved performance overthe convection numbers published by the heatsinkmanufacturer. The addition of a fan blowing across theheatsinks will allow the amplifier to operate at a 4 ohm loadcontinuously.

EVALUATION BOARD DESCRIPTIONS

100 Watt Amplifier

The 100 watt amplifier was constructed on a double-sided0.062″ G10 glass epoxy printed circuit board measuring 7inches x 12 inches. All components are included on the PCboard with the exception of the power transformer and bridgerectifier. Grounding is done with the star method; each pointin the circuit that is connected to ground has its own tracerunning to the input connection. This helps to reduce groundloops and give optimum noise performance. The powersupply center tap is connected as close to the centerbetween the supply filter capacitors as is physically possible.

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This keeps the power supply ripple balanced and as low aspossible. The input devices and transconductance stagesare grouped as close as physically possible on the board.This helps to reduce parasitics. Table 1 shows the actualmeasured performance of the 100 watt amplifier. Plots of themeasurements are shown in Figures 18 through 23. The DCstability of this amplifier measured at the output of theamplifier is ±100 mV at room temperature. All measurementswere made at room temperature. If this parameter is critical,a DC servo loop could be added to reduce the DC outputoffset. The schematics of the power supply and amplifier areshown in Figures 25 and 26. The components used in theconstruction of this amplifier are listed in Table 3.

200 Watt Amplifier

The 200 watt amplifier was also constructed on a doublesided 0.062″ G10 glass epoxy printed circuit boardmeasuring 13.5 inches x 12 inches. Other than the additionof the additional output devices and the two additionalheatsinks, this board is identical to the 100 watt amplifier.Table 2 shows the actual measured performance of the 200watt amplifier. Plots of the measurements are shown inFigures 27 through 33. The schematics of the power supplyand amplifier are shown in Figures 34 and 35. Thecomponents used in the construction of this amplifier arelisted in Table 4.

2.49063.246

R

2.40060.960

.1303.302

.1002.540

.2556.477

DIA.

1.25031.750

2.36059.944

2.62566.675

Figure 17. Heatsink Profile

6071WT. 2.22 LB/1′T.R. 1.8°C/WH.D.S. 31.7 SQ.″ / 1″

INCHESmm

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Table 1. Specifications — 100 Watt Amplifier

Parameter Performance Notes

Output Power into an 8 ohm load 100 watts 20 Hz to 20 kHz

Output Power into a 4 ohm load 175 watts 20 Hz to 20 kHz

Power bandwidth @ 8 ohms +0 dB, –3 dB 2 Hz–210 kHz* Source impedance = 600 ohms

Total Harmonic Distortion + Noise @ 8 ohms < .03% 20 Hz to 20 kHz, 1 watt to 100 watts

Total Harmonic Distortion + Noise @ 4 ohms < .05% 20 Hz to 20 kHz, 1 watt to 175 watts

Voltage Slew Rate 125 V /µs

DC output offset voltage < ±50 mV @ 25°CRise time < 1.8 µs 50% of full scale output voltage with 8 ohm load

Input Impedance > 32 k ohm 20 Hz to 20 kHz

Input Sensitivity for rated output 1.18 volts RMS 100 watts @ 8 ohms or 28.3 volts RMS

Input Sensitivity for 1 watt out @ 8 ohms 118 mV RMS

Output Impedance < 80 milliohms 20 Hz to 20 kHz

Damping Factor 100 Referred to 8 ohms

Output Noise A weighted < 75 µV Input shunted with 600 ohms

Output Noise C weighted < 150 µV Input shunted with 600 ohms

Signal to Noise A weighted 90 dB Referred to 1 watt out @ 8 ohms or 2.83 volts RMS

Signal to Noise C weighted 105 dB Referred to 100 watts @ 8 ohms or 28.3 volts RMS

Short Circuit current 7.5 A Peak

Peak output current 13 A Resistive load

Open-loop bandwidth 5 Hz–21 kHz No load, source impedance = 600 ohms,(small signal, inside output termination network)

Open-loop voltage gain 52.5 dB No load (small signal)

Closed-loop voltage gain 27.8 dB No load

*Output power above 10 kHz must be limited to 10% duty cycle to prevent over stressing the output termination network.

Table 2. Specifications — 200 Watt Amplifier

Parameter Performance Notes

Output Power into an 8 ohm load 200 watts 20 Hz to 20 kHz

Output Power into a 4 ohm load 300 watts 20 Hz to 20 kHz

Power bandwidth @ 8 ohms +0 dB, –3 dB 2 Hz–190 kHz* Source impedance = 600 ohms

Total Harmonic Distortion + Noise @ 8 ohms < .04% 20 Hz to 20 kHz, 1 watt to 200 watts

Total Harmonic Distortion + Noise @ 4 ohms < .09% 20 Hz to 20 kHz, 1 watt to 300 watts

Voltage Slew Rate 125 V /µs

DC output offset voltage < ±50 mV @ 25°CRise time < 1.9 µs 50% of full scale output voltage with 8 ohm load

Input Impedance > 32 k ohm 20 Hz to 20 kHz

Input Sensitivity for rated output 1.07 volts RMS 200 watts @ 8 ohms or 40 volts RMS

Input Sensitivity for 1 watt out @ 8 ohms 0.076 mV RMS

Output Impedance < 80 milliohms 20 Hz to 20 kHz

Damping Factor 100 Referred to 8 ohms

Output Noise A weighted < 100 µV Input shunted with 600 ohms

Output Noise C weighted < 200 µV Input shunted with 600 ohms

Signal to Noise A weighted 89 dB Referred to 1 watt out @ 8 ohms or 2.83 volts RMS

Signal to Noise C weighted 106 dB Referred to 200 watts @ 8 ohms or 40 volts RMS

Short Circuit current 9 A Peak

Peak output current 14 A Resistive load

Open-loop bandwidth 5 Hz–18 kHz No load, source impedance = 600 ohms,(small signal, inside output termination network)

Open-loop voltage gain 56.7 dB No load (small signal)

Closed-loop voltage gain 31.3 dB No load

*Output power above 10 kHz must be limited to 10% duty cycle to prevent over stressing the output termination network.

Page 12: Amplificator Motorola

AN1308MOTOROLA12

100 WATT AMPLIFIER MEASUREMENTS

∆ t 1.78 µs

Figure 18. Square Wave Rise Time Full Scale — 8 Ω

∆ t 1.6 µs

Figure 19. 20 kHz Square Wave Full Power — 8 Ω

∆ t 50 ms f 20 Hz

Figure 20. 20 Hz Square Wave Full Power — 8 ΩT/div 0.2 ms

Figure 21. 1 kHz, Full Power, 8 Ω HarmonicDistortion = 0.018%

T/div 10 µs

Figure 22. 20 kHz, Full Power, 8 Ω HarmonicDistortion = 0.028%

T/div 10 ms

Figure 23. 20 Hz, Full Power, 8 Ω HarmonicDistortion= 0.016%

10 V/div

10 V/div

Page 13: Amplificator Motorola

MOTOROLA13

AN1308

+ 52 V

– 52 V

GND

BR1

T1

THERMALBREAKER

60°C

5 AMP

120 Vac60 Hz

Notes: 8 Bifilar turns of #15 magnet wire are added to the secondary of the transformerNotes: to increase the voltage needed to obtain the required dc voltage.

Figure 25. 100 Watt Amplifier Power Supply Schematic

TERMINALSTRIP

Figure 24. 20 kHz, 1 Watt, 8 Ω HarmonicDistortion = 0.014%

T/div 10 µs

Page 14: Amplificator Motorola

AN

1308M

OT

OR

OLA

14

C190.1 µF

C11000 µF

+ C1010,000 µF

E1

E2

R221 MEG

C200.1 µF

+ C21000 µF

INPUT

R1851 k

C5

2.2 µF

R21

4.7 k

R31100 k

R20220

R1933.2 k3

12

R1751 k

+ C3470 µF Z1

1N5252B

R152.2 k

R52.21 k

R710 k

R95300

R9511

R1122.1

C15

C9122P

+ C4470 µF Z2

1N5252B

R162.2 k

R62.21 k

R94300

R10511

R1222.1

R810.0 k

C18

330P

C9022P

Q6MPSW06

Q82SC3298B

Q3MPS8599

R3

221

R4

221Q4

MPS8599

R241.0 k

C6

10P

R1

221

R2

221

Q1MPS8099

Q2MPS8099

R23

27.4 k

Q5MPSW56

Q72SA1306B

R93100

C16330P

C17330P

R92100

R34

15

D4

1N4148

D21N4148

R32100

3

12

R252.7 k

R273.3 k

R29200

R262.7 k

R73500

3

12

R33

15

D1

1N4148

C7

0.1 µF

R72500

3

12

Q19MJF15030

Q202SC3281

Q212SC3281

Q222SC3281

Q23MJF15031

Q242SA1302

Q252SA1302

Q262SA1302

E6

Q14MPS650

–52 V

+52 V

R9122 k

Q31MPSA06

R283.3 k

R30100

D31N4148

R38

221

R37

221

R39

221

+ C13100 µF

R40

221R41

221

R42

221

C8

0.1 µF

Q13MPS750

R9022 k

R50

1.2

R7122.1

R51

1.2R52

1.2

R630.47

R600.47

R49

1.2

–52V

E5+52 V

R48

1.2

R610.47

R620.47

E4

+

E3

SPEAKER2 µH

L1

R69

4.7R7010

C140.1 µF

R53

1.2

R640.47

R650.47

+ C1210,000 µF

Figure 26. 100 Watt Amplifier Schematic

+

330P

Page 15: Amplificator Motorola

AN1308 MOTOROLA15

200 WATT AMPLIFIER MEASUREMENTS

∆ t 1.90 µs T/div 1 µs

Figure 27. Square Wave Rise Time Full Scale — 8 Ω

∆ t 2 µs

Figure 28. 200 kHz Square Wave Full Power — 8 Ω

∆ t 50 ms 20 V/div

Figure 29. 20 Hz Square Wave Full Power — 8 Ω Figure 30. 1 kHz Full Power — 8 Ω HarmonicDistortion = 0.019%

T/div 10 µs

Figure 31. 20 kHz Full Power — 8 Ω HarmonicDistortion = 0.037%

T/div 10 ms

Figure 32. 20 Hz Full Power — 8 Ω HarmonicDistortion = 0.023%

20 V/div

T/div 0.2 ms

Page 16: Amplificator Motorola

16MOTOROLA AN1308

Notes: 13 Bifilar turns of #15 magnet wire are added to the secondary of the transformerNotes: to increase the voltage needed to obtain the required dc voltage.

Figure 34. 200 Watt Amplifier Power Supply Schematic

+ 81 V

– 81 V

GND

BR1

T1

THERMALBREAKER

60°C

8 AMP

120 Vac60 Hz

TERMINALSTRIP

Figure 33. 20 kHz, 1 Watt, 8 Ω HarmonicDistortion = 0.026%

T/div 10 µs

Page 17: Amplificator Motorola

MO

TO

RO

LA17

AN

1308

C15

330 P

R46

1.2

C190.1 µF

+ C1470 µF

R1751 k Z1

1N4752A

+ C3470 µF

R711.5 k

R154.3 k

R95300

R52.21 k

R35750

R9122 k

+C94700 µF

+ C104700 µF

R7451

R47

1.2

R570.47

R580.47

R590.47

E2

E5

+81 V

R45

1.2

Q162SC3281

Q15MJF15030

Q19MJF15030

Q72SA1306B

Q92SA1306B

Q5MPSW56 Q14

MPS650

Q11 MPS8099

R48

1.2R49

1.2

R600.47

+C114700 µF

C14 0.1 µF

R69

4.7

R61 0.47

R620.47

R630.47

R640.47

R650.47

R51

1.2 R52

1.2 R53

1.2

R50

1.2

R7010.0

R54

1.2R55

1.2

R7551

R660.47

R670.47

R680.47

+ C124700 µF

C8

0.1 µF

Q13MPS750

R9022 k R44

750

R43750

R42

221

Q102SC3298B

R41

221

R40

221

R39

221

R38

221

R37

221

Q31MPSA06

+ C13100 µF

R73500

R30100

R29200

R32100

R273.9 k

R252.7 k

R72500

C7

0.1 µF

R93100

C16330 P

C17330 P

R23

39.2 k

C6

10 P

R241.0 k

Q2 MPS8099

Q1MPS8099

R1

221

R2

221

R3

221

R4

221

C9022 P

C9122 P

C18

330 P

R811.5 k

Q82SC3298B

Q6MPSW06

R1222.1

R1422.1

R10511

R94300

R62.21 k

Z21N4752A

C200.1 µF

+ C4470 µF

R1851 k

R164.3 k

+ C2470 µF

R31100

R1933.2 k

E1

R20220

R221 MEG

R21

4.7 k

C5

2.2 µF

R92100

R7122.1

R56

1.2

INPUT

Q12 MPS8599

R9511

R1122.1

R1322.1

R33

15

D1

1N4148

R283.9 k

R262.7 k

D31N4148

D4

1N4148

R34

15

Q202SC3281

Q212SC3281

Q222SC3281

Q172SC3281

Q182SC3281

R36750

L1

2 µH

Q23MJF15031

Q242SA1302

Q252SA1302

Q262SA1302

Q27MJF15031

Q282SA1302

Q292SA1302

Q302SA1302

E6

–81 V

12

3

12

3

12

3

Q4 MPS8599

Q3MPS8599

D21N4148

12

3

–81V

+81V

E4

+

E3

SPEAKER

Figure 35. 200 Watt Amplifier Schematic

Page 18: Amplificator Motorola

18MOTOROLA AN1308

Table 3. Electrical Parts List for 100 Watt High Fidelity Amplifier Board

Designators Qty Description Rating Manufacturer Part Number

BR1 1 General Purpose Bridge Rectifier General Instruments KBPC35-02

C1, 2 2 1000 µF, ±20%, Electrolytic Capacitor 63 volt Panasonic ECE-A1JGE102

C3, 4 2 470 µF, ±20%, Electrolytic CapacitorLow Leakage, Less than 50 µA

50 volt Panasonic ECE-A50Z470

C5 1 2.2 µF, Polypropolyne Capacitor

C6 1 10 pF, Mica Capacitor 500 volt

C7, 8, 14, 19, 20 5 0.1 µF, X7R Ceramic Capacitor

C10, 12 2 10,000 µF, ±20%, ElectrolyticCapacitor (35.5 mm x 51 mm x 10 mm)

63 volt Panasonic ECE S1JU103Z

C13 1 100 µF, ±20%, Electrolytic Capacitor 10 volt Panasonic ECE-A1AGE101

C15, 16, 17, 18 4 330 pF, Mica Capacitor 500 volt

C90, 91 2 22 pF, Mica Capacitor 500 volt

D1, 2, 3, 4 4 Signal Diode 1N4148

L1 1 2 µH — 6 turns airwound#16 wire, 1.625″ diameter

Q1, 2 2 NPN Transistor 80 volt Motorola MPS8099

Q3, 4 2 PNP Transistor 80 volt Motorola MPS8599

Q5 1 NPN Transistor 80 volt Motorola MPSW56

Q6 1 NPN Transistor 80 volt Motorola MPSW06

Q7 1 PNP Transistor 200 volt Motorola 2SA1306B

Q8 1 NPN Transistor 200 volt Motorola 2SC3298B

Q13 1 PNP Transistor 40 volt Motorola MPS750

Q14 1 NPN Transistor 40 volt Motorola MPS650

Q19 1 NPN Transistor 150 volt Motorola MJF15030

Q20, 21, 22 3 NPN Transistor 200 volt Motorola 2SC3281

Q23 1 PNP Transistor 150 volt Motorola MJF15031

Q24, 25, 26 3 PNP Transistor 200 volt Motorola 2SA1302

Q31 1 NPN Transistor 80 volt Motorola MPSA06

R1, 2, 3, 4, 37, 38,39, 40, 41, 42

10 221 Ω, Resistor

R5, 6 2 2.21 kΩ, Resistor

R7, 8 2 10 kΩ, Resistor

R9, 10 2 511 Ω, Resistor

R11, 12, 71 3 22.1 Ω, Resistor

R15, 16 2 2.2 kΩ, 5%, metal oxide Resistor 3 watt Panasonic ERG-3SJ222

R17, 18 2 51 kΩ, 5%, carbon film Resistor

R19 1 33.2 kΩ, Resistor

R20 1 220 Ω, 5%, Resistor

R21 1 4.7 kΩ, 5%, carbon film Resistor

R22 1 1 MΩ, 5%, carbon film Resistor

R23 1 27.4 kΩ, Resistor

R24 1 1 kΩ, Resistor

R25, 26 2 2.7 kΩ, 5%, metal oxide Resistor 3 watt Panasonic ERG-3SJ272

R27, 28 2 3.3 kΩ, 5%, metal oxide Resistor 2 watt Panasonic ERG-2SJ332

R29 1 200 Ω, 5%, Resistor

R30, 92, 93 3 100 Ω, 5%, Resistor

R31 1 100 kΩ, 20%, Trim Potentiometer 1 /2 watt Panasonic EVM-QOGA01B15

R32 1 100 Ω, 20%, Trim Potentiometer 1 /2 watt Panasonic EVM-QOGA01B12

R33, 34 2 15 Ω, 5%, metal oxide Resistor 1 watt Panasonic ERG-3SJ150

R48, 49, 50, 51, 52, 53 6 1.2 Ω, 5%, carbon film Resistor 1 /2 watt

Note: All resistors are 1/4 watt with a tolerance of 1% unless otherwise noted.Note: All capacitors are 100 volt with a tolerance of 10% unless otherwise noted.

Page 19: Amplificator Motorola

MOTOROLA19

AN1308

Table 3. Electrical Parts List for 100 Watt High Fidelity Amplifier Board (continued)

Designators Qty Description Rating Manufacturer Part Number

R60, 61, 62, 63, 64, 65 6 0.47 Ω, 5%, non-inductive Resistor 5 watt RG Allen Co.

R69 1 4.7 Ω, 5%, metal oxide Resistor 3 watt Panasonic ERG-3SJ47R

R70 1 10 Ω, 5%, metal oxide Resistor 3 watt Panasonic ERG-3SJ100

R72, 73 2 500 Ω, 20%, Trim Potentiometer 1 /2 watt Panasonic EVM-QOGA01B52

R90, 91 2 22 kΩ, 5%, carbon film Resistor 1 /2 watt

R94, 95 2 300 Ω, 5%, Resistor

T1 1 120 V toroidal power transformer 360 VA Toroid Corp. of MD 636.332

Z1, 2 2 24 volt, Zener Diode 1 /2 watt Motorola 1N5252B

Fuse 1 5 amp slo-blo 250 volt

Breaker 1 60°C thermal breaker 15A Elmwood Sensors

Heatsinks, Output 2 Aluminum Extrusion, 7 inches long Aham Tor Inc. #6071

Heatsinks, pre-drivers 2 Aluminum Extrusion, 2.5 inches long Aham Tor Inc. #4405

— 1 14 pin IC socket tin plated

Note: All resistors are 1/4 watt with a tolerance of 1% unless otherwise noted.Note: All capacitors are 100 volt with a tolerance of 10% unless otherwise noted.

Page 20: Amplificator Motorola

20MOTOROLA AN1308

Table 4. Electrical Parts List for 200 Watt High Fidelity Amplifier Board

Designators Qty Description Rating Manufacturer Part Number

BR1 1 General Purpose Bridge Rectifier General Instruments KBPC35-02

C1, 2 2 470 µF, ±20%, Electrolytic Capacitor 100 volt Panasonic ECE-A1JGE470

C3, 4 2 470 µF, ±20%, Electrolytic CapacitorLo Ω Leakage, Less than 50 µA

50 volt Panasonic ECE-A50Z470

C5 1 2.2 µF, Polypropolyne Capacitor 100 volt

C6 1 10 pF, Mica Capacitor 500 volt

C7, 8, 14, 19, 20 5 0.1 µF, X7R Ceramic Capacitor 100 volt

C9, 10, 11, 12 4 4,700 µF, ±20%, Electrolytic Capacitor35.5 mm x 51 mm x 10 mm)

100 volt Panasonic ECE S1JU472Z

C13 1 100 µF, ±20%, Electrolytic Capacitor 10 volt Panasonic ECE-A1AGE101

C15, 16, 17, 18 4 330 pF, Mica Capacitor 500 volt

C90, 91 2 22 pF, Mica Capacitor 500 volt

D1, 2, 3, 4 4 Signal Diode 1N4148

L1 1 2 µH — 6 turns airwound#16 wire, 1.625″ diameter

Q1, 2, 11 3 NPN Transistor 80 volt Motorola MPS8099

Q3, 4, 12 3 PNP Transistor 80 volt Motorola MPS8599

Q5 1 NPN Transistor 80 volt Motorola MPSW56

Q6 1 NPN Transistor 80 volt Motorola MPSW06

Q7, 9 2 PNP Transistor 200 volt Motorola 2SA1306B

Q8, 10 2 NPN Transistor 200 volt Motorola 2SC3298B

Q13 1 PNP Transistor 40 volt Motorola MPS750

Q14 1 NPN Transistor 40 volt Motorola MPS650

Q15, 19 2 NPN Transistor 150 volt Motorola MJF15030

Q16, 17, 18, 20, 21, 22 3 NPN Transistor 200 volt Motorola 2SC3281

Q23, 27 1 PNP Transistor 150 volt Motorola MJF15031

Q24, 25, 26, 28, 29, 30 3 PNP Transistor 200 volt Motorola 2SA1302

Q31 1 NPN Transistor 80 volt Motorola MPSA06

R1, 2, 3, 4, 37, 38, 39, 40,41, 42

10 221 Ω, 5%, Resistor

R5, 6 2 2.21 kΩ, Resistor

R7, 8 2 11.5 kΩ, Resistor

R9, 10 2 511 Ω, Resistor

R11, 12, 13, 14, 71 5 22.1 Ω, Resistor

R15, 16 2 4.3 kΩ, 5%, metal oxide Resistor 3 watt Panasonic ERG-3SJ432

R17, 18 2 51 kΩ, 5%, carbon film Resistor

R19 1 33.2 kΩ, Resistor

R20 1 220 Ω, 5%, Resistor

R21 1 4.7 kΩ, 5%, carbon film Resistor

R22 1 1 MΩ, 5%, carbon film Resistor

R23 1 39.2 kΩ, Resistor

R24 1 1 kΩ, Resistor

R25, 26 2 2.7 kΩ, 5%, metal oxide Resistor 3 watt Panasonic ERG-3SJ272

R27, 28 2 3.9 kΩ, 5%, metal oxide Resistor 2 watt Panasonic ERG-2SJ392

R29 1 200 Ω, 5%, Resistor

R30, 92, 93 3 100 Ω, 5%, Resistor

R31 1 100 Ω, 20%, Trim Potentiometer 1 /2 watt Panasonic EVM-QOGA01B15

R32 1 100 Ω, 20%, Trim Potentiometer 1 /2 watt Panasonic EVM-QOGA01B12

R33, 34 2 15 Ω, 5%, metal oxide Resistor 1 watt Panasonic ERG-3SJ150

R35, 36, 43, 44 4 750 Ω, non-inductive Resistor 5 watt

Note: All resistors are 1/4 watt with a tolerance of 1% unless otherwise noted.Note: All capacitors are 100 volt with a tolerance of 10% unless otherwise noted.

Page 21: Amplificator Motorola

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AN1308

Table 4. Electrical Parts List for 200 Watt High Fidelity Amplifier Board (continued)

Designators Qty Description Rating Manufacturer Part Number

R37, 38, 39, 40, 41, 42 6 221 Ω, Resistor

R45, 46, 47, 48, 49, 50,51, 52, 53, 54, 55, 56

12 1.2 Ω, 5%, carbon film Resistor 1 /2 watt

R57, 58, 59, 60, 61, 62,63, 64, 65, 66, 67, 68

12 0.47 Ω, 5%, non-inductive Resistor 5 watt RG Allen Co

R69 1 4.7 Ω, 5%, metal oxide Resistor 3 watt Panasonic ERG-3SJ47R

R70 1 10 Ω, 5%, metal oxide Resistor 3 watt Panasonic ERG-3SJ100

R72, 73 2 500 Ω, 20%, Trim Potentiometer 1 /2 watt Panasonic EVM-QOGA01B52

R74, 75 2 51 Ω, Resistor

R90, 91 2 22 kΩ, ,5%, carbon film Resistor 1 /2 watt

R94, 95 2 300 Ω, 5%, Resistor

T1 1 120 V toroidal power transformer 600 VA Toroid Corp. of MD 660.502

Z1, 2 2 33 volt, Zener Diode 1 watt Motorola 1N4752A

Fuse 1 5 amp slo-blo 250 volt

Breaker 1 60°C thermal breaker 15A Elmwood Sensors

Heatsinks, Output 4 Aluminum Extrusion, 7 inches long Aham Tor Inc. #6071

Heatsinks, pre-drivers 4 Aluminum Extrusion, 2.5 inches long Aham Tor Inc. #4405

— 1 14 pin IC socket tin plated

Note: All resistors are 1/4 watt with a tolerance of 1% unless otherwise noted.Note: All capacitors are 100 volt with a tolerance of 10% unless otherwise noted.

Page 22: Amplificator Motorola

22MOTOROLA AN1308

Motorola reserves the right to make changes without further notice to any products herein. Motorola makes no warranty, representation or guarantee regardingthe suitability of its products for any particular purpose, nor does Motorola assume any liability arising out of the application or use of any product or circuit,and specifically disclaims any and all liability, including without limitation consequential or incidental damages. “Typical” parameters can and do vary in differentapplications. All operating parameters, including “Typicals” must be validated for each customer application by customer’s technical experts. Motorola doesnot convey any license under its patent rights nor the rights of others. Motorola products are not designed, intended, or authorized for use as components insystems intended for surgical implant into the body, or other applications intended to support or sustain life, or for any other application in which the failure ofthe Motorola product could create a situation where personal injury or death may occur. Should Buyer purchase or use Motorola products for any suchunintended or unauthorized application, Buyer shall indemnify and hold Motorola and its officers, employees, subsidiaries, affiliates, and distributors harmlessagainst all claims, costs, damages, and expenses, and reasonable attorney fees arising out of, directly or indirectly, any claim of personal injury or deathassociated with such unintended or unauthorized use, even if such claim alleges that Motorola was negligent regarding the design or manufacture of the part.Motorola and are registered trademarks of Motorola, Inc. Motorola, Inc. is an Equal Opportunity/Affirmative Action Employer.

Page 23: Amplificator Motorola

MOTOROLA23

AN1308

Page 24: Amplificator Motorola

24MOTOROLA AN1308

Literature Distribution Centers:USA: Motorola Literature Distribution; P.O. Box 20912; Phoenix, Arizona 85036.EUROPE: Motorola Ltd.; European Literature Centre; 88 Tanners Drive, Blakelands, Milton Keynes, MK14 5BP, England.JAPAN: Nippon Motorola Ltd.; 4-32-1, Nishi-Gotanda, Shinagawa-ku, Tokyo 141 Japan.ASIA-PACIFIC: Motorola Semiconductors H.K. Ltd.; Silicon Harbour Center, No. 2 Dai King Street, Tai Po Industrial Estate,ASIA-PACIFIC: Tai Po, N.T., Hong Kong.

AN1308/D◊