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8/10/2019 5. Kha-DC03-Equalization.pdf
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EqualizationHa Hoang Kha, Ph.D
Ho Chi Minh City University of Technology
Email: [email protected]
Chapter 3
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Content
1. Introduction
Multipath fading
ISI
2. Discrete-time channel model
3. Linear equalizer
Zero-forcing equalizer
MMSE equalizer
4. Nonlinear equalizer ZF decision feedback equalizer
MMSE decision feedback equalizer
Equalization 2 H. H. Kha, Ph.D.
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ISI due to Multi-Path Fading
Transmitted signal:
Received Signals:
Line-of-sight:
Reflected:
The symbols add up onthe channel
Distortion!
Delays
Equalization 4 H. H. Kha, Ph.D.
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Performance of multipath channels
First, compare 1-tap (i.e. f lat) Rayleigh-fading channelvs AWGN.
i.e. y = hx + w vs y = x + w
Note: all multipaths with random attenuation/phasesare aggregated into 1-tap
Next consider f requency select iv i ty, i.e. multi-tap,
broadband channel, with multi-paths Effect: ISI
Equalization techniques for ISI & complexities
Equalization 6 H. H. Kha, Ph.D.
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Flat Fading (Rayleigh) vs AWGN
Equalization 7 H. H. Kha, Ph.D.
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ISI/Freq. Selective Channel
Typical BER vs. S/N curves
S/N
BER
Frequency-selective channel(no equalization)
Flat fading channel
Gaussianchannel
(no fading)
Frequency selective fading
irreducible BERfloor!!!
Equalization 8 H. H. Kha, Ph.D.
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Freq. selective channel with equalization
Typical BER vs. S/N curves
S/N
BER
Flat fading channel
Gaussianchannel
(no fading)
Diversity (e.g. multipath diversity)
Frequency-selective channel
(with equalization)
improvedperformance
Equalization 9 H. H. Kha, Ph.D.
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Rayleigh Flat Fading Channel
BPSK: Coherent detection.
Conditional on h,
Averaged over h,
at high SNR.
Looks like
AWGN, butpeneeds to be
unconditioned
To get a much
poorer scaling
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2. Discrete time model for ISI channels
{bm} is a data sequence.
Equalization 11 H. H. Kha, Ph.D.
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Examples of waveforms
The output of the impulse modulator
L: the length of the data sequence
T: symbol duration
Equalization 12 H. H. Kha, Ph.D.
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Noise is assumed to be zero-mean white Gaussian with
double-sided spectral density
and autocorrelation function
The received signal after receiver filtering
a modified transmitter filter includes the transmitter filter
and channel
Channel output without noise
Equalization 13 H. H. Kha, Ph.D.
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Discrete-time CIR:
The sampled signal of y(t) at every T seconds is
given by
Sampled noise
Discrete channel
Equalization 14 H. H. Kha, Ph.D.
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Given
Example of discrete channel
Find the discrete-time CIR
Find the sampled noise nl
Equalization 15 H. H. Kha, Ph.D.
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With the discrete-time CIR channel
can be found
Solution
Equalization 16 H. H. Kha, Ph.D.
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The sampled noise is given by
Solution
The variance of noise is given by
It can be shown that
Equalization 17 H. H. Kha, Ph.D.
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The discrete-time CIR is a causal impulse response
of length P, i.e.,
The discrete-time channel
The received signal in the discrete-time domain is
given by
Equalization 18 H. H. Kha, Ph.D.
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Suppose that is the desired signal
ISI problem
The received signal can be re written as
is a delay.
Due to the ISI terms, the desired signal cannot
be clearly observed from the received signals.
It is necessary to eliminate the ISI terms to extract
the desired signal.
Equalization 19 H. H. Kha, Ph.D.
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It is desired when the noise is ignored.
Zero-forcing equalizer
In z-domain, it is required
The linear equalizer is called the zero-forcing (ZF) equalizer because the ISI is forced to be
zero.
Equalization 21 H. H. Kha, Ph.D.
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The noise after ZF equalization is
Zero-forcing equalizer
If has nulls (in frequency response), the
variance of noise can be infinity.
Another disadvantage of the ZF equalization is thatZF equalizer has an infinite impulse response for
finite-length channels.
Equalization 22 H. H. Kha, Ph.D.
m t ph c ng su tca nhiu
c p ng xung d i t i v c ng cho k nhc p ng xung hu hn
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Assume that is independent identically distributed
(iid) and , .
Minimum mean square error linear equalizer
The error is defined as
The desired signal
The MSE is given by
Rewritten in matrix form as
M: the length of the LE.
Equalization 23 H. H. Kha, Ph.D.
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MMSE equalizer: Orthogonal principle
The error should be uncorrected with at
the optimality, i.e.,
where and
At the optimality, the MMSE is given by
Equalization 24 H. H. Kha, Ph.D.
sai s kh ng tng quanvi tn hiu ng vo
-->
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MMSE equalizer: derivative
Note that the MSE cost function is a quadratic
function of g.
The minimum is obtained from
This is the same condition found from the
orthogonality principle.
Equalization 25 H. H. Kha, Ph.D.
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4. Nonlinear equalizer: DFE
Structure of the decision feedback equalizer
DFE
Equalization 27 H. H. Kha, Ph.D.
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Zero-forcing DFE
The convolution of gm and hp is given by
M: length of FFF and P is the length of channel
Equalization 28 H. H. Kha, Ph.D.
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Zero-forcing DFE
Output of the FFF is given by
To estimate at time l
Assume are available.
Equalization 29 H. H. Kha, Ph.D.
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Zero-forcing DFE
The ISI due to these past symbols can be
eliminated
is the impulse response of the FBF
denotes the detected symbols of
If the decision are correct , we chose
Equalization 30 H. H. Kha, Ph.D.
oi t n p n isi c a c c i u tr c
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Zero-forcing DFE
In matrix form
When
Equalization 32 H. H. Kha, Ph.D.
mong mu n ch thuc cm
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Solution
Equalization 34 H. H. Kha, Ph.D.
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S
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MMSE DFE
ZE DFE only attempts to remove the ISI, the noise
can be enhanced
The MSE is given by
Equalization 36 H. H. Kha, Ph.D.
E l MMSE DFE
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Example: MMSE DFE
Equalization 37 H. H. Kha, Ph.D.
S l ti
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Solution
Equalization 38 H. H. Kha, Ph.D.
S l ti
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Solution
Equalization 39 H. H. Kha, Ph.D.
S l ti
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Solution
Equalization 40 H. H. Kha, Ph.D.
E l Ch l ith f ll
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Example: Channels with frequency nulls
Equalization 41 H. H. Kha, Ph.D.
E l BER f MMSE LE
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Example: BER for MMSE LE
Equalization 42 H. H. Kha, Ph.D.
khng dngb linear
E l BER f MMSE DFE
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Example: BER for MMSE DFE
Equalization 43 H. H. Kha, Ph.D.
dng b linear
5 Adapti e linear eq ali er
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5. Adaptive linear equalizer
Adaptive equalizers can be considered as practical
approaches.
They do not require second-order statistics of
signals.
A training sequence is used to find the equalization
for the LE or DFE.
Equalization 44 H. H. Kha, Ph.D.
Iterative approaches: steepest descent algorithm
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Iterative approaches: steepest descent algorithm
The output of the LE is given by
Assume that an LE is causal and has a finite length of M.
The MSE as a function of g is defined by
Equalization 45 H. H. Kha, Ph.D.
p ng xung
Steepest descent algorithmphng php lp
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Steepest descent algorithm
Suppose that is an initial vector
The gradient is defined as
The next vector which may yield a smaller than
can given by
Steepest descent Constant step size
Equalization 46 H. H. Kha, Ph.D.
phng php lp
nu o hm g0 >0 --> ang tng--> tr xung
Steepest descent algorithm
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Steepest descent algorithm
A recursion toward the minimum is given by
k: iteration index
The iteration is terminated when the SD direction
becomes zeros, i.e.,
Equalization 47 H. H. Kha, Ph.D.
Steepest descent algorithm
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Steepest descent algorithm
If is large, the recursive
diverges and never findsthe minimum.
If is too small, it would
take too many iterations
to converge.
It is important to
determine the value of
such that the recursioncan converge at a fast
rate.
Equalization 48 H. H. Kha, Ph.D.
u ln
Convergence analysis of the SD algorithm
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Convergence analysis of the SD algorithm
Note that
We consider the different vector
Egeindecomposition of is given by
Equalization 49 H. H. Kha, Ph.D.
sai s bclp th k
Convergence analysis of the SD algorithm
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Convergence analysis of the SD algorithm
Let . Then it follows that
represents the mth element of
We can fined the following property
Equalization 50 H. H. Kha, Ph.D.
Convergence analysis of the SD algorithm
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Convergence analysis of the SD algorithm
The sufficient condition for convergence is
is the maximum eigenvalue
Since
another practical sufficient condition is
Equalization 51 H. H. Kha, Ph.D.
Ry = E(yyT)
Least mean square (LMS) approach
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Least mean square (LMS) approach
The SD algorithm can overcome the matrix
inversion of the MMSE approach.However, it has not been overcome the need for
second-order statistics.
The least mean square (LMS) algorithm is anapproximation of the SD algorithm.
Recall the MSE
Equalization 52 H. H. Kha, Ph.D.
Least mean square (LMS) approach
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Least mean square (LMS) approach
The SD algorithm can be represented by
where
If is replaced by without the expectationwe can obtain the LMS approach
Equalization 53 H. H. Kha, Ph.D.
t Ul = Sl - gT.Yl
yl: tn hiu u vo b cnbng thi im l
ko cn quan tmknh truyn
Adaptive decision feedback equalizers
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Adaptive decision feedback equalizers
We the training sequence is available, we can use
the correct symbols in the DFE.The MSE of the DFE is written as
where
The adaptive DFE finds the equalization vector
from the and
The LMS algorithm can used for the adaptive DFE.
Equalization 54 H. H. Kha, Ph.D.
g: b cn b ng thunf: feedback
`tn hiu ng vo
Home work 1
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Home work 1
Given the discrete-time channel model as fig
bn={-1, +1} denotes the information bits
channel impulse response h=[0.227, 0.460, 0.688, 0.460, 0.227]
Wn is AWGN with zero mean and variance N0/2
Design and simulation ZF and ZF-DFE, and for
SNR=[5, 10, 15, 20, 25, 30]
Plot BER vs SNR
Channel hn
dnnb n
Equalizer filter
oise wn
Equalization 55 H. H. Kha, Ph.D.
dB
3. contemporary communication system usingmatlab
adaptive equalizer for isi ch
t chn m, chn chi u d i
Home work 2
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Home work 2
Given the discrete-time channel model as fig
bn={-1, +1} denotes the information bits
channel impulse response h=[0.227, 0.460, 0.688, 0.460, 0.227]
Wn is AWGN with zero mean and variance N0/2
Design and simulation MMSE and MMSE-DFE,
and for SNR=[5, 10, 15, 20, 25, 30]
Plot BER vs SNR
Channel hn
dnnb n
Equalizer filter
oise wn
Equalization 56 H. H. Kha, Ph.D.
Home work 3
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Home work 3
Given the discrete-time channel model as fig
bn={-1, +1} denotes the information bits channel impulse response h=[0.227, 0.460, 0.688, 0.460, 0.227]
Wn is AWGN with zero mean and variance N0/2
Design and simulation: LMS algorithm for MMSE
and MMSE-DFE for SNR=[5, 10, 15, 20, 25, 30]
Plot BER vs SNR, and the convergence for
different
Channel hn
dnnb n
Equalizer filter
oise wn